Sensors and Actuators B 98 (2004) 18–27
Wheatstone-Bridge readout interface for ISFET/REFET applications
Arkadiy Morgenshtein a,∗ , Liby Sudakov-Boreysha a , Uri Dinnar a ,
Claudio G. Jakobson a , Yael Nemirovsky b
a
Biomedical Engineering Department, Technion, Israel Institute of Technology, Haifa, Israel
Electrical Engineering Department, Technion, Israel Institute of Technology, Haifa, Israel
b
Received 10 April 2003; received in revised form 20 July 2003; accepted 29 July 2003
Abstract
The paper presents a novel readout configuration for ISFET sensors based on Wheatstone-Bridge connection. This design technique
allows on-chip integration, temperature compensation and measurements from ISFET/REFET pairs. The circuit is capable of operating
in differential mode, and can also perform common mode and combined measurements, while improving the immunity to noise and
interferences. The Wheatstone-Bridge interface benefits from enhanced operational flexibility, due to the ability of relative sensitivity
control of the output signal. Direct and indirect feedback configurations are presented with operational analysis, simulations and application
options. Simulation results showing 9 ␮V accuracy are presented. A 4 mm × 4 mm test chip in 1.6 ␮m CMOS technology was used for
laboratory experiments using MOSFETs for emulation of ISFET/REFET sensors.
© 2003 Elsevier B.V. All rights reserved.
Keywords: ISFET; Readout; Wheatstone-Bridge
1. Introduction
The ISFET sensor’s integration in clinical applications
for pH measurements requires features such as temperature compensation, body-effect elimination, REFET operation, noise inhibition and sensitivity control [1,2]. In
CMOS-based integrations n-channel ISFETs are mostly
used due to low drift and high mobility properties [7,8] and
p-type substrate is globally and constantly grounded. The
body effect in n-channel sensors is limiting the possibilities
of source biasing in ISFET, which is a fundamental component in currently presented interfaces for monolithic ISFET
integration [9–11]. Thus, the applicability of the existing
interfaces in standard CMOS technology is problematic and
development of new design techniques for ISFET readout
is essential in order to provide a combination of all the
mentioned above advantages.
The Wheatstone-Bridge technique is widely used in numerous measurement applications [6], as resistance measurements, strain gauges, etc. due to its exclusive structure
that allows reduced temperature sensitivity.
The novel ISFET readout interface based on WheatstoneBridge configuration is presented in this study. Feedback
∗ Corresponding author.
E-mail address: arkadiy@tx.technion.ac.il (A. Morgenshtein).
0925-4005/$ – see front matter © 2003 Elsevier B.V. All rights reserved.
doi:10.1016/j.snb.2003.07.017
implementation and combination of ISFET and MOSFET
devices in on-chip integrated structure allow high system
accuracy, low temperature sensitivity and compatibility for
CMOS-based applications, including REFET function. Operational analysis, simulation results and measurements of
the 4 mm × 4 mm test chip are presented.
2. Wheatstone-Bridge readout interface
2.1. Basic structure and operation
The ISFET sensor’s operation [3,4] is based on the
conversion of pH changes into a corresponding channel resistance. Thus, detection of fluctuations in channel conductivity can lead directly to pH level sensing.
Changes of channel resistance are caused by the threshold
voltage VT , which is correlated with pH with a certain
sensitivity factor (about 58 mV/pH in high-performance
sensors). Wheatstone-Bridge configuration is a good candidate for implementation in this type of system, where
temperature-compensated resistance detection is requested.
Fig. 1 shows the structure of Wheatstone-Bridge readout
interface. An ISFET sensor and three MOSFET devices are
applied in place of standard resistors. In order to maintain a
balanced bridge, the diagonal is connected to the operational
amplifier with feedback to the reference electrode of ISFET
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19
Note, that by replacing the M3 device (connected to V3) in
the presented structure with a low-sensitive ISFET, a REFET
measurement can be obtained. In this case the result will be
proportional to the pH response ratio of ISFET and REFET.
A practical implementation of ISFET/REFET configuration
is stipulated by development of robust REFET device with
electrical and thermal characteristics close to ISFET and
MOSFET [2].
2.2. Sensitivity analysis
In the case of direct and indirect feedback, the fact that the
voltages at the inputs to the operational amplifier must be
equal ensures that Vout tracks changes in VT , regardless of
whether the transistors are in the linear or saturation region.
The sensitivity analysis of the circuit can be best performed by examination of the influence of the feedback voltage to the gate terminal and of the change in VT value of
ISFET, on the channel current according to Shockley model
[5]:
Id = 21 β(Vgs − VT )2 (1 + λVds )
Fig. 1. Direct (a) and indirect (b) Wheatstone-Bridge.
(2)
where λ is channel length modulation parameter and β a
parameter determined by physical properties of device:
W
L
(direct feedback) or the gate of a corresponding MOSFET
(indirect feedback).
In a standard bridge (Fig. 2) with gage connected in diagonal, the following expression describes the relationship
between the changes of four resistances and the diagonal
voltage Vg :
R1
r
R2
R3
R4
Vg =
VS (1)
−
+
−
R1
R2
R3
R4
(1 + r)2
β = µCox
where r is the ratio between the corresponding resistors.
Two important properties can be derived from (1): (a) the
changes in resistance due to temperature fluctuations result
in zero total contribution to Vg , assuming similar influence
of temperature on devices; and (b) changes in channel resistance of ISFET will contribute to Vg change. The operational
amplifier detects this change and feedback is applied to the
reference electrode or the gate of MOSFET, to maintain the
balance of the bridge, by adjusting the transconductance of
the corresponding device.
The best way to perform the sensitivity analysis of indirect
feedback is to inspect the behavior of the channel conductivity of the FET in saturation and linear regions:
Fig. 2. A standard Wheatstone-Bridge circuit.
(3)
where µ is the mobility, Cox the gate insulator capacitance,
W and L the width and length of the channel.
This equal influence of the Vgs and VT voltages on the
channel current allows a simple expression for Vout in case
of direct feedback configuration:
Vout = Vg = VT (pH)
(direct)
(4)
1
∂Ids,sat
1
=
=
Rsat
∂Vds
Kλ(Vgs − VT )2
(5)
∂Ids,linear ∼
1
1
=
=
Rlinear
∂Vds
K(Vgs − VT )
(6)
where K = β/2 is device-specific parameter.In the indirect
feedback configuration, the operation is based on the ratio
of conductivities of the corresponding FETs. The change of
VT due to pH fluctuation in one FET causes a change in gate
voltage of the correspondent one. Thus, the sensitivity factors derived from the ratio between two channel resistances
R1 and R2 are:
K 1 λ1
S=
(saturation)
(7)
K 2 λ2
S=
K1
K2
(linear)
(8)
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while the expression of the dependence of output voltage on
threshold voltage fluctuations due to pH in indirect feedback
is given by:
Vout = S VT (pH)
(9)
Here, the amplification factor can be controlled by proper
sizing the corresponding FETs.
3. Elimination of body effect in Wheatstone-Bridge
circuit
Fig. 3. A body-effect-free Wheatstone-Bridge circuit.
The threshold voltage of field-effect transistor in CMOS
technology is expressed as:
VT = VFB −
QB
+ 2φF
Cox
(10)
where VFB is the flat-band voltage, QB the depletion charge
in the silicon and φF the Fermi potential [4]. The regular
assumption for an ISFET is that VFB also contains terms,
which reflect the interfaces between the liquid and the gate
oxide, and the liquid and the reference electrode; which
makes VFB sensitive to the changes of pH. The terms QB ,
φF and Cox are assumed to be constant and uninfluenced by
pH or operation point changes.
However, even if not influenced by pH, the threshold voltage VT is not constant with respect to the voltage difference
VBS between the substrate and the source of the MOS transistor.
When an on-chip implementation of ISFET together with
related readout interfaces is considered, it is important to
remember, that all devices comprising an MOS device are
made on a common substrate. In a standard CMOS technology, it is a p-type substrate which is connected to a lowest
circuit potential.
In most of the existing readout techniques, the source of
ISFET is not constantly biased, and is used as an internal
node of the circuit, or a point of feedback application. When
VBS is not 0, the expression for the threshold voltage is
modified to incorporate VBS as follows:
√
2εSi qNA (2φB + |VBS |)
VT = VFB + 2φB +
(11)
Cox
where φB is the bulk potential.
This expression is critical, because of the influence of VBS
on the value of VT in integrated implementations of ISFET.
The term of VBS , if getting a non-zero value (which will
happen in most of the on-chip realizations of known readout
interfaces) causes a parasitic change in VT that is not due
to the change of pH level. The error that occurs in the case
of body effect is significant, and depending on technology
and operation point, the threshold shift can reach more than
a half of the initial VT .
Wheatstone-Bridge technique in configurations that were
presented in Fig. 1 has an advantage common to differential
techniques, which allows elimination of body-effect influence on measurement results. The body effect occurs in M1
(ISFET) and M2 when the bulks are connected to ground
(as all the substrate of the chip) and not to sources, as shown
in Fig. 1. However, this does not change the final result of
readout, due to an equal influence of body effect on M1 and
M2 (due to equal dimensions and VBS biases in the balanced
bridge), causing equal changes in transistors conductivity.
So, according to (1) same relationships will be obtained between the changes of four resistances and the diagonal voltage Vg , resulting in the same output voltage as without the
body effect. Thus, the body-effect influence is rejected here
as a common mode signal.
However, if needed, one can prevent the appearance of
body effect in the ISFET by using p-type MOSFETs as components of the bridge, as shown in Fig. 3. These transistors
have to be properly sized to match the resistance demands
that were applied to n-type MOSFETs in a regular configuration. This ratio is between 2.3 and 4 and is dependant
on technology parameters. An appropriate voltage has to be
applied to the gates of the devices in order to maintain the
same operation regime in all four transistors. This allows
placing the n-type ISFET, so that its source and substrate
will be constantly and equally biased. This configuration can
be used for p-type ISFET as well, to obtain operation with
identical devices.
4. Test results and implementations
4.1. Simulation results
Test circuit was implemented in Cadence, using transistor models from the MOSIS fabrication process. Test
simulations were performed using the SpectreS simulator,
using 1.6 ␮m technology models. An n-channel 300/30 ␮m
n-type ISFET sensor was emulated by MOSFET device
of the same dimensions and was used in simulation with
400 mV sinusoidal voltage applied to its gate, to represent
various pH levels (considering a typical ISFET sensitivity
of 53 mV). The simulations were performed at 1–500 Hz
frequencies, in order to ensure operation in different conditions of pH fluctuations. Three 300/30 ␮m n-MOSFETs and
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21
Fig. 4. Transient simulation results of Wheatstone-Bridge interface.
a 5 Vp–p amplifier were connected to obtain the required
configuration. A constant 2.3 V bias was applied to the
reference electrode (in indirect configuration), or to the correspondent MOSFET gate (in direct feedback) to maintain
operation in saturation region. A 1 V voltage was given to
the opposite MOSFET pair; while an additional bias voltage
was applied to the gate of M4 to compensate the built-in
offset voltage of the designed operational amplifier. In the
case of discrete applications, adjusting of the commercial
amplifier can perform the same compensation.
The results of the simulation can be seen in Fig. 4. The input signal is represented by fluctuations in threshold voltage
and plotted together with the resulting output signal and the
difference between two signals. A high accuracy with less
Fig. 5. Response of Wheatstone-Bridge to temperature fluctuations.
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Fig. 6. A 1800 ␮m × 860 ␮m layout of Wheatstone-Bridge circuit.
then 9 ␮V error was observed for the maximal simulated pH
levels.
One of the most prominent advantages of WheatstoneBridge readout is the temperature compensation. The influence of temperature is demonstrated in simulation in Fig. 5.
The temperature was changed in range of 20–40 ◦ C and the
resulting changes in output voltage were measured. As can
be seen, the Wheatstone-Bridge readout is practically insensitive to temperature fluctuations.
4.2. Layout implementation
In order to estimate the feasibility of CIMP implementation in miniaturized measurement equipment, a realistic
layout of both feedback configurations was carried out. The
layout implementations of Wheatstone-Bridge readout circuit in 1.6 ␮m CMOS technology are presented in Fig. 6.
The layout area of 1800 ␮m × 860 ␮m makes the circuit
suitable for implementation in a common catheter with 1mm
diameter for clinical applications, as well as in any kind of
miniaturized system. In this circuit, an operational amplifier
was implemented on-chip together with ISFET and MOSFET devices.
4.3. Test results
The verification of the Wheatstone-Bridge readout interface was performed using four MOSFET devices from the
fabricated test chip. The circuit was implemented in the indirect configuration and an OP77 operational amplifier was
utilized for feedback application. The interface was tested
under various operational conditions and signal forms, while
the n-channel 300/30 ␮m ISFET sensor was emulated by
MOSFET device with similar properties.
Some of the measurements of the output response to
sinusoidal and triangle waveforms that were obtained, are
presented in Figs. 7 and 8. A certain settling time can be
observed in the response to high-slope transition in Fig. 8.,
after which the output is precisely following the changes
in the threshold voltage. As can be seen from the measurements the response of the Wheatstone-Bridge readout interface is accurate for various signal forms and frequencies.
Fig. 7. Measured response of indirect feedback Wheatstone-Bridge to 400 mVp–p sinus fluctuations at 10 Hz.
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Fig. 8. Measured response of indirect feedback Wheatstone-Bridge to 500 mVp–p triangle fluctuations at 100 Hz.
5. Wheatstone-Bridge operation in REFET and
common mode
5.1. REFET operation
The Wheatstone-Bridge interface is very suitable for applications of REFET measurements. The participation of
four FETs in the bridge allows operation in differential mode
without the need for any change in the circuit configuration.
The concept of the differential operation can be seen from
the following representation of (1):


differential differential r
R3
R2
R4 
 R1
Vg =
+
−
−

 VS
R1
R2
R3
R4
(1 + r)2
(12)
According to this expression, there are two options
for obtaining differential ISFET/REFET operation in
Wheatstone-Bridge: (1) using M1 and M2 transistors as sensors; and (2) using M3 and M4 transistors as sensors. Each
of the pairs will produce the required differential response
in the output. The measurements of readout operation in
differential REFET mode are presented in Fig. 9.
An additional feature of Wheatstone-Bridge interface can
be observed from (12): in the case when two ISFET/REFET
pairs are operating simultaneously, the resulting output response is a sum of responses of each differential pair. This
feature can become an important advantage, if two identical
pairs are considered: the summation of the output signals after in-pair differentiation might contribute to enhanced noise
immunity of the output signal. Generally, the differentiation can supply certain immunity degree to changes in light
Fig. 9. Measurements of differential REFET operation in Wheatstone-Bridge.
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level and to drift, if the operational characteristics of the
fabricated ISFETs and MOSFETs in the circuit are similar.
Nevertheless, because of the non-uniformity of the devices
in recent fabrication processes [2], additional efficient techniques should be considered: application of light-insensitive
metal layers (as Pt) at the gate area of ISFET, or drift elimination by surface discharging [7].
5.2. Common mode operation
The summing operation of the bridge can be regarded as
an operation in common mode. The expression in (12) can
be rewritten to emphasize the common mode operation of
the interface:


common common r
R
R
R
R
1
3
2
4

 VS
Vg =
−
+
+
R1
R3
R2
R4
(1+r)2
(13)
As can be seen from (13), the summation of signals can be
obtained in two cases: (1) using M1 and M3 transistors as
sensors; and (2) using M2 and M4 transistors as sensors.
Each of the pairs will produce the required common mode
response in the output. This operational concept was verified by measurements, resulting in waveforms presented in
Fig. 10. The output response is a weighted summation of the
two ISFET responses, while the weights are function of the
absolute transconductances of the sensors (i.e. R1 and R3 ).
If two pairs of sensors are considered: M1 and M3 transistors as ISFET sensors, and M2 and M4 transistors as REFET
sensors; a combined operation will be obtained, resulting
in separate summation of ISFET and REFET responses and
differentiation of both pairs signals. As shown in Fig. 11, the
common reference electrode is connected to the amplifier
output as in direct feedback and the ISFETs and REFETs
Fig. 11. Combined common and differential mode operation of two ISFET
and REFET pairs.
experience a common mode solution bias. The differential
input signal comes from the VT difference between ISFET
and REFET pairs, which depends on pH. This double function produces enhanced noise immunity together with interference immunity obtained by REFET.
6. Controlled sensitivity of Wheatstone-Bridge
Close inspection of the expression in (1) reveals an additional feature of the Wheatstone-Bridge readout: the ability
to control the sensitivity of the sensors. It results from the
fact that the fluctuations in channel resistance of each FET
are divided by the absolute values of channel resistance.
Thus, the set-point resistances define the gain factor for each
FET in the bridge. The final relative sensitivity of the output
signal to the responses of each sensor can be controlled by
adjusting the gate voltage of the FET, which is identical to
adjusting its set-point channel resistance.
The ability of relative sensitivity control is an important
feature due to the operational flexibility obtained in the
Fig. 10. Measurements of common mode operation in Wheatstone-Bridge.
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25
Fig. 12. Demonstration of relative sensitivity control in differential ISFET/REFET pair: (a) simulations; (b) measurements.
circuit, without need in any configuration change or hardware addition.
Several sets of simulations and measurements were carried out in order to verify and demonstrate the concept of
sensitivity control. One of the experiments is presented in
Fig. 12, showing the responses of differential ISFET/REFET
pair for various set-point voltages applied to the gates. In
this case, the device representing the ISFET was responding to sinus fluctuations in the in the gate, while the REFET
was activated by square signals. The relative sensitivity was
controlled by adjusting the DC levels of the gate voltage
of the FETs. Both devices in the experiment were n-type,
thus increasing of the gate voltage of the device caused decreased resistance, and increased sensitivity of the output
signal to the response of this device. Left waveforms in simulations and measurements demonstrate the case in which
the ISFET was biased by higher gate voltage than REFET,
which made its response dominant with slight fluctuations
caused by REFET. Right waveforms present the opposite
case, where the REFET was dominant in the output sig-
Fig. 13. Demonstration of relative sensitivity control in common mode ISFET operation.
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Acknowledgements
This work was supported by the Women Division/ATS
MEP XXV Project.
References
Fig. 14. Microphotograph of the 4 mm × 4 mm test chip. ISFETs are
designated for further post-processing in future experiments.
nal, while slightly influenced by sinusoidal fluctuations from
ISFET response.
Another experiment was performed for pair of ISFETs operating in common mode. One of the ISFETs was responding to triangle fluctuations, while the other was responding
to sinusoidal signals. The results of sensitivity control of the
output signal can be seen in the measurements in Figs. 13
and 14.
Here the resulting sensitivity of the output signal was
also controlled by relative adjustment of gate voltages of
the devices. In each case, another device has a dominant
influence on the output, while the total response is the sum
of the response of the dominant and the inhibited devices.
7. Conclusions
The novel readout technique for ISFET-based applications based on Wheatstone-Bridge circuit was presented,
allowing temperature-compensated pH measurement without body effect, by determination of the channel resistance
changes in ISFET sensor. Simulation results were presented,
showing 2.4 ␮V/pH accuracy for 58 mV/pH sensitivity. A
4 mm × 4 mm test chip in 1.6 ␮m CMOS technology was
used for laboratory experiments. The measurements of the
interface showed an accurate response for a wide range of
forms and frequencies of pH fluctuations.
The circuit is capable of operating in REFET mode, and
can also perform common mode and combined measurements, while improving the immunity to noise and interferences.
The Wheatstone-Bridge interface benefits from enhanced
operational flexibility, due to the ability of relative sensitivity control of the output signal. This feature is determined
from the operational concept of the circuit, and does not
demand any configuration changes or hardware additions.
Wheatstone-Bridge readout interface proves to be a robust
alternative for ISFET integration in system-on-chip implementations in CMOS process.
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Biographies
Arkadiy Morgenshtein was born in Cishinev, Moldova in 1977. He
received his BSc degree in electrical engineering from Technion, Israel
Institute of Technology, Haifa, Israel, in 1999. He is currently working
on his MSc degree in biomedical engineering at Technion. He has been
a teaching and research assistant at Electrical Engineering Department,
Technion since 1999. His research interests include low-power design
techniques for digital circuits, biosensor microsystems for biotelemetry,
and digital cameras design in CMOS technology.
Liby Sudakov-Boreysha received her BSc degree in electrical engineering
from Technion, Israel Institute of Technology, Haifa, Israel, in 2001.
She is currently working on her MSc degree in biomedical engineering
at Technion. She is a teaching and research assistant at Electrical Engineering Department, Technion. She works at IBM Haifa Research Labs
as a staff member since 2000. Her research interests include analog and
mixed signals circuits, biosensor microsystems for brain monitoring, and
wide-band linear amplifiers.
Claudio G. Jakobson was born in Buenos Aires, Argentina in 1966. He
received his PhD degree from the Technion, Israel Institute of Technology
in 2001, MSc degree in electrical engineering from the Technion, Israel
Institute of Technology in 1995, and Electronic Engineer degree from
the University of Buenos Aires, Argentina, in 1992. His PhD research
focused on CMOS compatible ISFET microsystems, noise and drift in
ISFETs, as well as the application of ISFETs for brain monitoring at
the cerebro-spinal fluid. The research was granted the Eshkol scholarship
from the Israeli Ministry of Science and the support of the Women
A. Morgenshtein et al. / Sensors and Actuators B 98 (2004) 18–27
Division/ATS MEP XXV Project. His MSc thesis was on low noise
CMOS analog channels for X-ray detection. His research contributed to
the space X-ray detection experiment at the Technion satellite TECHSAT, including VLSI electronics that successfully operated on space.
In 2001, he joined Bluebird Optical MEMS Ltd. and is now working
on the development of micro-electro-mechanical systems (MEMS) and
microsystems. Other fields of research and expertise are VLSI analog
electronics, MEMS, readout interfaces for CMOS compatible sensors,
and noise phenomena in MOSFETs.
Yael Nemirovsky (IEEE Fellow, IEE Fellow ’99) received her BSc degree
in 1966 and DSc degree in 1971 from the Technion, Israel Institute of
Technology, Haifa. She joined the Department of Electrical Engineering
in Technion in 1980. Prior to that she was a research scientist specializing in microelectronics in Rafael, a National R&D Organization. She
graduated from Technion in chemistry and her DSc thesis was in electrochemistry. For over 20 years, she has been active in electro-optical
devices in II–VI compound semiconductors and additional advanced
semiconductor materials as well as infrared focal plane arrays. She has
been involved in growth, processing, device design and modeling of
detectors as well as VLSI circuits. She has a well-equipped MOCVD
laboratory for growth of heterostructures, extensive facilities for device
and interfaces processing and characterization. She has been a principal
investigator in large funded research programs that ended in prototype
infrared detectors and systems that were transferred to industry. Twice
she was the Head of the Microelectronics Research Center of the Department of Electrical Engineering at Technion. Currently, her research
focuses on micro-opto-electro-mechanical systems (MOEMS), CMOS
compatible micromachining and microsystems implemented in CMOS
technology and integrated with silicon devices. She has published over
27
130 papers in the open literature, has filed several patents and a large
number of classified reports. She has collaborated with the microelectronics industry as a consultant in sensors and VLSI technology and
has been quite active in national and international conferences. She
has supervised over 40 graduate students for MSc and DSc. She is an
IEEE Fellow, an IEE Fellow and has been the Chairperson of the Israeli
Association for Crystal Growth. Currently, she is the Chairperson of the
Microelectronics and Photonics Section of URSI. In the past, she received
awards as a “Best Teacher” at Technion, a national award of high esteem
“The Award for the Security of Israel” and a Technion award for “Novel
Applied Research”. She has received The Kidron Foundation award for
“Innovative Applied Research” (a US$ 100,000 grant for research program). She is a distinguished lecturer of the electron device society of
IEEE.
Uri Dinnar was born in Israel in 1939. He received his BSc degree in medical engineering from the Technion, Israel Institute of Technology, Haifa,
in 1964 and MSc and PhD degrees in engineering and applied physics
from Harvard University, Cambridge, MA, in 1967 and 1969, respectively.
He is currently Head of the Department of Biomedical Engineering at the
Technion, where he is also the Director of the Laboratory of Biological
Fluid Dynamics and holds the Henry Goldberg Chair of Biomedical Engineering. He joined the Technion in 1969 and was appointed full professor
with the Department of Biomedical Engineering in 1990. He held visiting appointments at the College of Medicine, Michigan State University,
East Lansing, from 1976 to 1978, Drexel University, Philadelphia, PA, in
1983, University of Houston, Houston, TX, University of Texas Medical
Branch, Galveston, in 1991, and the City College of New Your in 1999.
His research interests are in cardiovascular fluid dynamics, blood flow in
bones, and micro-devices for physiological monitoring.