TR41.9/00-05-031
DRAF SPECTRUM MANAGEMENT STANDARD
From T1E1.4
(as of May 1, 2000)
COMMITTEE T1 – TELECOMMUNICATIONS
Working Group T1E1.4
Lisle, IL; May 1, 2000
T1E1.4/2000-002R2
CONTRIBUTION
TITLE:
Draft proposed American National Standard, Spectrum Management for Loop
Transmission Systems
SOURCE*: Editor
PROJECT: T1E1-38, Spectral Compatibility Aspects for Facilities between a Central Office and the
Network-to-Customer Interface (Twisted Pair Transmission System)
ABSTRACT
Attached is year 2000 R2 version of the draft standard for Spectrum Compatibility. It reflects the
changes resulting from agreements (or provisional agreements) made during the February 2000
meeting in Lahaina, HI.
NOTICE
This is a draft document and thus, is dynamic in nature. It does not reflect a consensus of Committee T1Telecommunications and it may be changed or modified. Neither ATIS nor Committee T1 makes any representation
or warranty, express or implied, with respect to the sufficiency, accuracy or utility of the information or opinion
contained or reflected in the material utilized. ATIS and Committee T1 further expressly advise that any use of or
reliance upon the material in question is at your risk and neither ATIS nor Committee T1 shall be liable for any
damage or injury, of whatever nature, incurred by any person arising out of any utilization of the material. It is
possible that this material will at some future date be included in a copyrighted work by ATIS.
* CONTACT: Craig F. Valenti; email: cvalenti@telcordia.com; Tel: 973-829-4203; Fax: 973-829-5962
American National Standard
For Telecommunications
Spectrum Management
For Loop Transmission Systems
Secretariat
Alliance for Telecommunications Industry Solutions
Approved <Date to be determined>
American National Standards Institute, Inc.
Abstract
This standard provides spectrum management requirements and recommendations for the
administration of services and technologies that use metallic subscriber loop cables. Spectrum
management is the administration of the loop plant in a way that provides spectral compatibility for
services and technologies that use pairs in the same cable. In order to achieve spectral compatibility,
the ingress energy that transfers into a loop pair, from services and transmission system technologies
on other pairs in the same cable, must not cause an unacceptable degradation of performance. In
addition, the egress energy from a particular loop pair must not transfer into other pairs in a manner
that causes an unacceptable degradation in the performance of services and technologies on those
pairs. This standard includes signal power limits and technology deployment guidelines for digital
subscriber line spectrum management classes. It also provides a generic analytical method to
determine spectral compatibility.
This is a draft document and thus, is still dynamic in nature.
American
National
Standard
Approval of an American National Standard requires verification by ANSI that the
requirements for due process, consensus, and other criteria for approval have
been met by the standards developer.
Consensus is established when, in the judgment of the ANSI Board of Standards
Review, substantial agreement has been reached by directly and materially
affected interests. Substantial agreement means much more than a simple
majority, but not necessarily unanimity. Consensus requires that all views and
objections be considered, and that a concerted effort be made toward their
resolution.
The use of American National Standards is completely voluntary; their existence
does not in any respect preclude anyone, whether he has approved the standards
or not, from manufacturing, marketing, purchasing, or using products, processes,
or procedures not conforming to the standards.
The American National Standards Institute does not develop standards and will in
no circumstances give an interpretation of any American National Standard.
Moreover, no person shall have the right or authority to issue an interpretation of
an American National Standard in the name of the American National Standards
Institute. Requests for interpretations should be addressed to the secretariat or
sponsor whose name appears on the title page of this standard.
CAUTION NOTICE: This American National Standard may be revised or
withdrawn at any time. The procedures of the American National Standards
Institute require that action be taken periodically to reaffirm, revise, or withdraw
this standard. Purchasers of American National Standards may receive current
information on all standards by calling or writing the American National Standards
Institute.
Published by
American National Standards Institute
11 West 42nd Street, New York, New York 10036
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Table of Contents
Foreword ........................................................................................................................................ viii
1.
Scope, purpose, and application............................................................................................... 2
1.1
Scope................................................................................................................................. 2
1.2
Purpose ............................................................................................................................. 3
1.3
Application ......................................................................................................................... 3
2.
Normative references ............................................................................................................... 3
3.
Definitions, abbreviations, acronyms, and symbols .................................................................. 4
4.
5.
3.1
Definitions .......................................................................................................................... 4
3.2
Abbreviations, acronyms, and symbols ............................................................................. 6
General Information .................................................................................................................. 8
4.1
Crosstalk............................................................................................................................ 8
4.2
Spectral compatibility......................................................................................................... 9
4.3
Spectrum management ..................................................................................................... 9
4.3.1
Basis loop systems..................................................................................................... 9
4.3.2
Legacy systems........................................................................................................ 12
4.3.3
Signal power limitations (method A) ......................................................................... 12
4.3.4
Technology deployment guidelines .......................................................................... 13
4.3.5
Analytical method of determining spectral compatibility (method B) ........................ 14
Signal power limits and other criteria ...................................................................................... 15
5.1
Short-term stationary systems ......................................................................................... 15
5.2
Spectrum management classes ...................................................................................... 15
5.2.1
Spectrum management class 1................................................................................ 15
5.2.2
Spectrum management class 2................................................................................ 16
5.2.3
Spectrum management class 3................................................................................ 17
5.2.4
Spectrum management class 4................................................................................ 17
5.2.5
Spectrum management class 5................................................................................ 18
5.2.6
Spectrum management class 6................................................................................ 19
5.2.7
Spectrum management class 7................................................................................ 20
5.2.8
Spectrum management class 8................................................................................ 20
5.2.9
Spectrum management class 9................................................................................ 21
5.3
6.
Spectral Compatibility Limitations for Repeatered Systems ............................................ 22
Conformance testing methodology ......................................................................................... 22
6.1
General conformance criteria .......................................................................................... 22
6.2
PSD conformance criteria unique to spectrum management classes............................. 23
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6.2.1
Specific conformance criteria for spectrum management class 1 ........................... 23
6.2.2
Specific conformance criteria for spectrum management class 2 .......................... 23
6.2.3
Specific conformance criteria for spectrum management class 3 ........................... 24
6.2.4
Specific conformance criteria for spectrum management class 4 .......................... 24
6.2.5
Specific conformance criteria for spectrum management class 5 ........................... 24
6.2.6
Specific conformance criteria for spectrum management class 6 ........................... 24
6.2.7
Specific conformance criteria for spectrum management class 7 .......................... 24
6.2.8
Specific conformance criteria for spectrum management class 8 .......................... 24
6.2.9
Specific conformance criteria for spectrum management class 9 .......................... 24
6.3
PSD and total average power measurement procedure ................................................. 24
6.3.1
Test circuit for PSD and total average power measurement.................................... 24
6.3.2
Calibration of the test circuit and termination impedance......................................... 25
6.3.3
Operation of the DUT ............................................................................................... 25
6.3.4
Total average power measurement procedure ........................................................ 25
6.3.5
Power spectral density (PSD) measurement procedure .......................................... 25
6.4
Short-term stationary conformance criteria ..................................................................... 26
6.4.1
Determination of whether to apply short-term stationary conformance criteria ........ 26
6.4.2
Continuous mode for conformance testing .............................................................. 26
6.4.3
Frequency domain requirements.............................................................................. 26
6.4.4
Time domain requirements ...................................................................................... 27
6.5
Transverse balance testing methodology........................................................................ 27
6.6
Longitudinal output voltage testing methodology............................................................. 28
Annex A: Evaluation of interference from new technologies into existing technologies................. 43
A.1
Goals and framework for evaluation ................................................................................... 43
A.2
Analytical Method: Detailed crosstalk margin evaluations .................................................. 44
A.2.1
General Methodology................................................................................................... 44
A.2.2
DFE-based PAM signals (e.g., 2B1Q ISDN and HDSL) .............................................. 46
A.2.3
DFE-based QAM/CAP signals ..................................................................................... 46
A.2.4
DMT margin computations........................................................................................... 46
A.2.4
A.2.5 ........................................................................................................................... 47
A.3
Compatibility with voicegrade services and technologies.................................................... 47
A.3.1
Description of voicegrade services and technologies .................................................. 47
A.3.1.1 Speech signals ......................................................................................................... 48
A.3.1.2 Single and dual tone signals..................................................................................... 48
A.3.1.3 Low frequency (< 100 Hz) signals ............................................................................ 48
A.3.1.4 Digital data................................................................................................................ 48
A.3.1.5 Analog data .............................................................................................................. 48
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A.3.2
Voicegrade evaluation.................................................................................................. 48
A.3.2.1 Evaluation loop ......................................................................................................... 49
A.3.2.2 Reference crosstalk environment............................................................................. 49
A.3.2.3 Crosstalk noise and peak power levels computation ............................................... 49
A.3.3
A.4
Spectral compatibility of voicegrade systems with basis systems ............................... 50
Compatibility with Enhanced Business Services ................................................................. 50
A.4.1
Description of Enhanced Business Services ............................................................... 50
A.4.1.1 Speech signals ......................................................................................................... 50
A.4.1.2 Signalling functions................................................................................................... 50
A.4.2
Enhanced Business Service Evaluation....................................................................... 51
A.4.2.1 Reference crosstalk environment............................................................................. 51
A.4.2.2 Crosstalk noise and peak power levels computation ............................................... 51
A.4.2.3 Spectral compatibility of Enhanced Business Services with basis systems ............. 51
A.5
Compatibility with T1.410 .................................................................................................... 52
A.5.1
Computation of DDS Performance – Margin Computation for AMI Transceivers ....... 52
A.5.2
Evaluation loops ........................................................................................................... 53
A.5.3
Reference crosstalk environment ................................................................................ 53
A.5.4
Margin computation...................................................................................................... 54
A.6
Compatibility with ISDN DSL ............................................................................................... 54
A.6.1
Evaluation loops ........................................................................................................... 54
A.6.2
Reference Crosstalk environment................................................................................ 54
A.6.3
Margin Computation..................................................................................................... 54
A.7
Compatibility with HDSL ...................................................................................................... 55
A.7.1
Evaluation loops ........................................................................................................... 55
A.7.2
Reference crosstalk environment ................................................................................ 55
A.7.3
Margin computation...................................................................................................... 55
A.8
Compatibility with ADSL and RADSL technologies ............................................................. 55
A.8.1
Evaluation loops and performance levels .................................................................... 55
A.8.2
Reference crosstalk environments .............................................................................. 56
A.8.3
Margin computation...................................................................................................... 56
A.8.4
Compatibility with RADSL ............................................................................................ 57
A.9
Compatibility with HDSL2 .................................................................................................... 57
A.9.1
Evaluation loops ........................................................................................................... 57
A.9.2
Reference crosstalk environment ................................................................................ 58
A.9.3
Margin computation...................................................................................................... 58
A.10 Compatibility with 2B1Q SDSL ............................................................................................ 59
A.10.1
Evaluation loops and performance levels .................................................................... 59
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A.10.2
Reference crosstalk environment ................................................................................ 59
A.10.3
Margin computation...................................................................................................... 59
A.10.4
2B1Q SDSL Technology Specification......................................................................... 59
A.10.4.1
Power Spectrum Density ...................................................................................... 59
A.10.4.2
Performance ......................................................................................................... 60
A.10.4.3
Return loss............................................................................................................ 60
A.10.4.4
Longitudinal Balance............................................................................................. 61
A.11 Combination of crosstalk sources: composite crosstalk model .......................................... 61
A.12 Customer end-point separation ........................................................................................... 61
Annex B: Loop Information............................................................................................................. 74
B.1
General................................................................................................................................ 74
B.1.1
The loop environment .................................................................................................. 74
B.1.1.1 Background noise..................................................................................................... 75
B.1.1.2 Impulse noise ........................................................................................................... 75
B.1.1.3 Radio frequency interference (RFI) .......................................................................... 75
B.1.1.4 Structural cable faults ............................................................................................... 75
B.1.1.5 The loop environment............................................................................................... 75
B.1.1.6 Telephone cable and subscriber loop structures...................................................... 76
B.1.2
Loop plant design rules: resistance design .................................................................. 77
B.1.3
Loop plant design rules: carrier serving area (CSA) .................................................... 78
B.1.4
Distribution area (DA)................................................................................................... 79
B.1.5
Loop statistics .............................................................................................................. 79
B.2
AWG and metric cable: diameters and DC resistance and capacitance ............................ 79
B.3
Cable primary constants (RLGC) characterization.............................................................. 80
B.3.1
Transmission-Line Characterization ............................................................................ 80
B.3.1.1 “ABCD” modeling...................................................................................................... 80
B.3.1.2 Transmission-line RLCG characterization................................................................ 82
B.3.1.3 Power for transmission lines .................................................................................... 84
B.3.1.4 Reflection coefficients .............................................................................................. 85
B.3.1.5 Characterization of a bridge-tap section – a three-port ............................................ 86
B.3.1.6 Computation of transfer function .............................................................................. 86
B.3.1.7 Relationship of transfer function and “insertion loss” ............................................... 87
B.3.2
TP1............................................................................................................................... 89
B.3.3
TP2............................................................................................................................... 89
B.3.4
22-Gauge Phone-Line Twisted Pair ............................................................................. 89
B.3.5
TP3............................................................................................................................... 89
B.3.6
FP................................................................................................................................. 90
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B.3.7
Category-5 Twisted Pair............................................................................................... 90
B.3.8
Two-Pair Twisted Drop................................................................................................. 90
B.3.9
Two-Pair Quaded Drop ................................................................................................ 90
B.3.10
Flat-Pair Drop............................................................................................................... 90
B.3.11
Additional Models ......................................................................................................... 91
B.4
Cable crosstalk models ....................................................................................................... 91
B.4.1
Near end crosstalk, NEXT ........................................................................................... 91
B.4.2
Far end crosstalk, FEXT .............................................................................................. 93
B.4.3
Method for combining crosstalk contributions from unlike types of disturber .............. 93
B.4.3.1 Base models for NEXT and FEXT ........................................................................... 93
B.4.3.2 Combining crosstalk from mixed disturber types ..................................................... 94
B.4.3.3 Application to two NEXT terms................................................................................. 94
B.4.3.4 Application to FEXT terms........................................................................................ 95
B.4.3.5 Crosstalk is non-decreasing ..................................................................................... 96
B.4.3.6 All disturbers are treated equally .............................................................................. 96
B.4.3.7 Adding NEXT and FEXT .......................................................................................... 96
Annex C: Probability of error estimation....................................................................................... 120
C.1
Effect of input bit sequence ............................................................................................... 121
C.2
Period of injected “Gaussian” noise .................................................................................. 121
C.3
dB margin and importance sampling................................................................................. 122
Annex D: Additional spectrum management topics currently under study by the formulating
committee of this standard........................................................................................................... 123
Annex E: Time varying, user data-dependent crosstalk from T1 and DDS services ................... 124
Annex F: Non-continuous CO signaling events ........................................................................... 127
F.1
Ringing .............................................................................................................................. 127
F.2
Supervision (hook flash).................................................................................................... 127
F.3
Dial Pulse .......................................................................................................................... 128
Annex G: ADSL Calculated Capacities ........................................................................................ 131
Annex H: Technology Effects Of and On Legacy Systems.......................................................... 133
H.1
T1 Carrier .......................................................................................................................... 133
H.1.1
Margin computations for linear equalization systems (e.g., T1)................................. 133
H.1.2
Compatibility with AMI T1........................................................................................... 133
H.1.2.1 Evaluation Loops .................................................................................................... 134
H.1.2.2 Reference crosstalk environments ......................................................................... 134
H.1.2.3 Margin computation ................................................................................................ 134
Annex I: C-code ........................................................................................................................... 135
Annex J: Informative references .................................................................................................. 136
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TABLES
Table 1 - Spectrum management class 1 PSD template definition ............................................... 29
Table 2 - Minimum transverse balance requirements.................................................................... 29
Table 3 - Spectrum management class 2 PSD template definition ............................................... 30
Table 4- Spectrum management class 3 PSD template definition ................................................ 30
Table 5 - PSD mask definition for downstream transmission from a spectrum management class
4 TU-C............................................................................................................................................ 31
Table 6 - PSD mask definition for upstream transmission from a spectrum management class 4
TU-C............................................................................................................................................... 31
Table 7 - PSD template definition for downstream transmission from a spectrum management
class 5 TU-C .................................................................................................................................. 31
Table 8 - Spectrum management class 7 PSD template definition ............................................... 31
Table 9 - Spectrum management class 8 PSD template definition. .............................................. 32
Table 10 – PSD template definition for downstream transmission from a spectrum management
class 9 TU-C .................................................................................................................................. 32
Table 11 - PSD template definition for upstream transmission from a spectrum management class
9 or spectrum management class 5 TU-R ..................................................................................... 33
Table 12 - Temination impedances................................................................................................ 33
Table 13 - Resolution bandwidth for measuring a DUT PSD for conformance with spectrum
management classes 1, 2, 3, and 4. .............................................................................................. 33
Table 14 – Resolution bandwidth for measuring a DUT PSD for conformance with spectrum
management class 5. ..................................................................................................................... 33
Table 15 – Resolution bandwidth for measuring a DUT PSD for conformance with spectrum
management class 6. ..................................................................................................................... 34
Table 16 – Summary of transverse balance testing criteria........................................................... 34
Table 17 - Maximum longitudinal output voltage limit .................................................................... 34
Table A.1 - Code for DFE PAM SNR computation......................................................................... 62
Table A. 2 - Code for DFE QAM/CAP computation ....................................................................... 63
Table A.3 - Matlab code to set-up ADSL margin computation....................................................... 64
Table A. 4 -- Matlab Code to compute a DMT margin ................................................................... 66
Table A. 5 – Data Points for Unger NEXT Model (see Figure A. 1) ............................................... 67
Table A. 6 – Spectral Compatibility into downstream single-carrier RADSL.................................. 67
Table A. 7 – Spectral Compatibility into upstream single-carrier RADSL ...................................... 67
Table A. 8 - HDSL2_delta (in dB) for various test crosstalk combinations .................................... 67
Table A. 9 - 2B1Q SDSL data rate and associated spectrum management classes .................... 67
Table A. 10 – 2B1Q SDSL reach target at sample data rates ....................................................... 67
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Table B.1 – American wire gauge (AWG) and metric wire ............................................................ 97
Table B. 2 - Cable model parameters for TP1 (0.4 mm or 26-gauge twisted pair) ........................ 97
Table B.3 - Primary constants for TP1 (0.4 mm or 26-gauge twisted pair).................................... 98
Table B.4 - Cable parameters for 26-AWG PIC air core................................................................ 99
Table B.5– Cable parameters for 26-AWG filled PIC .................................................................. 100
Table B.6 – Cable model parameters for TP2 (0.5 mm or 24-gauge twisted pair) ...................... 102
Table B.7 – Primary constants for TP2 (0.5 mm or 24-gauge twisted pair) ................................. 102
Table B.8 – Cable parameters for 24-AWG PIC air core............................................................. 103
Table B.9 – Cable parameters for 22-AWG PIC air core............................................................. 105
Table B.10 – Cable model parameters for TP3 (DW10 reinforced .5 mm copper PVC-insulated
steel strength member, polyethelene sheath) .............................................................................. 106
Table B.11 – Primary constants for TP3 (DW10 reinforced .5 mm copper PVC-insulated steel
strength member, polyethelene sheath)....................................................................................... 107
Table B.12 – Cable model parameters for FP (1.14 mm flat cable) ............................................ 107
Table B.13 – Primary constants for FP (1.14 mm flat cable) ....................................................... 107
Table B.14 – Cable model parameters for category 5 twisted pair .............................................. 108
Table B.15 – Primary constants for category 5 twisted pair ......................................................... 108
Table B.16 – Cable parameters, two-pair twisted drop ................................................................ 109
Table B.17 – Cable parameters, two-pair quad drop ................................................................... 111
Table B.18 – Cable parameters, flat-pair Drop ............................................................................ 113
Table G. 1 – ADSL Evaluation Results ........................................................................................ 132
Table H. 1: Coefficients for T1 repeater input filtering gain equation........................................... 134
Table I. 1: C-code for DMT margin computation.......................................................................... 135
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FIGURES
Figure 1 – Configuration for evaluation of effect of NEXT and FEXT into downstream................. 37
Figure 2 – Configuration for evaluation of effect of NEXT and FEXT into upstream ..................... 37
Figure 3 - Spectrum management class 1 PSD template.............................................................. 38
Figure 4 - Spectrum management class 2 class PSD Template ................................................... 38
Figure 5 - Spectrum management class 3 PSD template.............................................................. 39
Figure 6 - PSD mask for downstream transmission from a spectrum management class 4 TU-C
....................................................................................................................................................... 39
Figure 7 - PSD mask for upstream transmission from a spectrum management class 4 TU-R.... 40
Figure 8 - Spectrum management class 7 PSD template.............................................................. 40
Figure 9 - Spectrum management class 8 PSD template............................................................. 41
Figure 10 - PSD and total average power measurement setup ..................................................... 41
Figure 11 – Example PSD and total average power measurement setup ..................................... 42
Figure 12 - Illustrative test configuration for transverse balance conformance testing .................. 42
Figure A. 1 – Unger NEXT model and simplified NEXT model of 1% NEXT for 18kft of 22GA PIC
....................................................................................................................................................... 68
Figure A. 2 – Crosstalk into a Basis System: NEXT and FEXT ..................................................... 69
Figure A. 3 – Simulation Model for Reference and New Crosstalk into Downstream Receiver..... 69
Figure A. 4 – Crosstalk into Basis System: NEXT & FEXT with reduced new loop length ............ 69
Figure A. 5 – Simulation Model for Self- and New Crosstalk into Downstream Receiver with
reduced new loop length ................................................................................................................ 70
Figure A. 6 - Process flow for spectral compatibility calculations................................................... 71
Figure A. 7 – 2B1Q SDSL PSD at several data rates .................................................................... 72
Figure A. 8 - 2B1Q SDSL PSD at several data rates ..................................................................... 72
Figure A. 9 - Minimum return loss for 784kbps system.................................................................. 73
Figure A. 10 - Longitudinal balance for 784kbps system ............................................................... 73
Figure B.1 – Loop ABCD parameters, impedance and voltages ................................................. 115
Figure B. 2 – Two-port network model. ........................................................................................ 116
Figure B.3 – Incremental section of twisted-pair transmission line. ............................................ 116
Figure B.4 – Simple load circuit for power analysis...................................................................... 116
Figure B.5 – Examples of two-port cascades for twisted-pair transmission line configurations .. 117
Figure B.6 – Near end crosstalk (NEXT) ..................................................................................... 117
Figure B.7 – NEXT power sum losses for 25 pairs of PIC cable binder group ............................ 118
Figure B.8 – Comparison of ANSI NEXT with Measured NEXT .................................................. 118
Figure B.9 – Far end crosstalk (FEXT) ........................................................................................ 119
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Figure B.10 – Comparison of ANSI FEXT with Measured FEXT................................................. 119
Figure E. 1 – Examples of T1 power spectral density variations ................................................. 125
Figure E. 2 – Examples of DDS power spectral density variations .............................................. 125
Figure E. 3 – Data dependent power changes in a wide band due to T1 data patterns .............. 126
Figure E. 4 – Data dependent power changes in a narrow band due to T1 data patterns........... 126
Figure F. 1 – Standard ringing potential with best case start/end ................................................ 129
Figure F. 2 – Standard ringing potential worst case start/end...................................................... 129
Figure F. 3 – Ringing waveforms (worst case generalization) ..................................................... 130
Figure F. 4 – Triple ringing interval............................................................................................... 130
Figure F. 5 – Simple battery feed arrangement ........................................................................... 130
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This is a draft document and thus, is still dynamic in nature.
Foreword (This foreword is not part of American National Standard T1.XXX-2000)
Accredited Standards Committee T1, Telecommunications serves the public through improved
understanding between carriers, customers, and manufacturers. Technical Subcommittee T1E1 of
Committee T1 develops telecommunications standards and technical reports related to various
digital subscriber line technologies. This standard is intended to be a living document, subject to
revision and updating as warranted by advances in network and equipment technology.
This standard provides spectrum management requirements and recommendations for the
administration of services and technologies that use metallic subscriber loop cables. Spectrum
management is the administration of the loop plant in a way that provides spectral compatibility for
services and technologies that use pairs in the same cable. In order to achieve spectral
compatibility, the ingress energy that transfers into a loop pair, from services and transmission
system technologies on other pairs in the same cable, must not cause an unacceptable
degradation of performance. In addition, the egress energy from a particular loop pair must not
transfer into other pairs in a manner that causes an unacceptable degradation in the performance
of services and technologies on those pairs. This standard includes signal power limits and
technology deployment guidelines for the digital subscriber line spectrum management classes
defined herein. It also provides a generic analytical method to determine spectral compatibility.
Because of the wide range of network switching systems, network transport systems, subscriber
loop plant, and customer installations in North America, conformance with this standard does not
guarantee spectral compatibility or acceptable performance under all possible operating
conditions.
ANSI guidelines specify two categories of requirements: mandatory and recommendation. The
mandatory requirements are designated by the word shall and recommendations by the word
should. Where both a mandatory requirement and a recommendation are specified for the same
criterion, the recommendation represents a goal currently identifiable as having distinct
compatibility or performance advantages.
There are 7 annexes in this standard. Annex A is normative and considered to be part of this
standard; Annexes B-I are informative and are not considered part of this standard, that is, they do
not include requirements but provide information that may be helpful to users of this standard.
Suggestions for improvement of this standard are welcome. They should be sent to the Alliance
for Telecommunications Industry Solutions, T1 Secretariat, 1200 G Street NW, Suite 500,
Washington, DC 20005.
This standard was processed and approved for submittal to ANSI by Accredited Standards
Committee on Telecommunications, T1. Committee approval of the standard does not necessarily
imply that all members voted for its approval. At the time it approved this standard, the T1
Committee had the following members:
G. H. Peterson, Chair
E. R. Hapeman, Vice-Chair
S. D. Barclay, Secretary
Organization Represented
Name of Representative
EXCHANGE CARRIERS
Exchange Carrier Member.............................................................................. Name of Representative
Name of Alternate (Alt)
Organization Represented
Name of Representative
INTEREXCHANGE CARRIERS
Interexchange Carrier Member ....................................................................... Name of Representative
Name of Alternate (Alt.)
MANUFACTURERS
Manufacturer Member .................................................................................... Name of Representative
Name of Alternate (Alt.)
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GENERAL INTEREST
General Interest Member................................................................................ Name of Representative
Name of Alternate (Alt.)
Technical Subcommittee T1E1 on Interfaces, Power and Protection of Networks, which is
responsible for the development of this standard, had the following members:
Ed Eckert, Chair
Dick Brandt, Vice-Chair
John Roquet, Secretary
Organization Represented
Name of Representative
Member Name
Organization Represented
Name of Representative
Name of Representative
Member Name
....................................................................................... Name of Representative
Name of Alternate (Alt.)
Working Group T1E1.4 on DSL Access, which had the technical responsibility during the
development of this standard, had the following members:
Thomas J. J. Starr, Chairman
Massimo Sorbara, Vice-Chairman
Ron McConnell, Secretary
Editors:
Craig Valenti,
John E. Roquet,
Richard A. McDonald,
Behrooz Rezvani
Syed A. Abbas
Robyn Aber
Oscar Agazzi
Cajetan M. Akujuobi
Ron Allen
Subra Ambati
Tariq Amjed
Candare M. Anderson
Ephraim Arnon
James Aslanis
Keith Atwell
Hiromitsu Awai
Jein Baek
Scott J. Baer
Rupert Baines
H. Charles Baker
LeRoy Baker
John T. Balinski
Chuck Balogh
Art Barabell
Uri Baror
John Barselloti
Roy Batruni
Don Bellenger
Daniel Bengtssen
Rafi Ben-Michael
Ben Bennett
Bill Bergman
Dev Bhattacharya
Nigel Billington
Bora Biray
Larry Bishop
Richard Bishop
Trone Bishop
Ray Blackham
R. T. Bobilin
Gary Bolton
Jan Bostrom
Mark F. Bowen
Bruce Bowie
Peter Brackett
Richard Brandt
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Dave Brier
Les Brown
Randy Brown
Curtis Brownmiller
William Buck
William Buckley
Bill Burton
John Bush
Richard Cam
John Camagna
Patrick Cameron
Jim Carlo
Art Carlson
Paulus Carpelan
C. A. Carpenter
Ken Cavanaugh
Guy Cerulli
Paul Chang
Yen T. Chang
Trang Chan-virak
Joe Charboneau
Adam Chellali
S. John Chen
W. I. H. Chen
Wen S. Chen
Raymond Chen
Daniel Chen
Hoover Chen
Jacky Chow
Peter Chow
John Cioffi
Alan Cohen
Nigel Cole
Terry Cole
Marty Colombatto
Kevin Cone
Greg Copeland
Graham G. Copley
Lawrence H. Corbett
Mauro Costa
Ray Countermann
Bill Crane
Phil Crawby
David Cummings
Kim Currie
Aaron Dagen
Tom Daly
Tamar Danon
Michel Darveau
Jim Dell
Michael Demjanenko
Shuang Deng
Andre' P. des Rosiers
Philip DesJardins
Franz Dielacher
Curtis Dodd
Jean-Louis Dolmeta
Guojie dong
Bernard Dugerdil
Craig Edwards
George Eisler
Tsur Eitan
Earl Emerson
Dan Etz-Hadar
Dave Evans
Vedat Eyuboglu
Charles Fadel
Guy Fedorkow
Michael Firth
Rocky Flaminio
Kay Fleskes
Steve Follett
Al Forcucci
Klaus Fosmark
Kevin Foster
Vladimir Friedman
Hans-Joerg Frizlen
Robin Gangopadhya
Clete Gardenhour
Juan Garza
Amit Gattani
Lajos Gazsi
Tom Geary
Nabil Gebrael
Al Gharakhanian
Emil Ghelberg
Mike Gilbert
Jim Girardeau
Hugh Goldberg
Yuri Goldstein
David Goodman
Richard Goodson
Steven Gordon
Linda Gosselin
Peter T. Griffiths
Glen Grochowski
John Gruber
Sanjay Gupta
L. B. Gwinn
Cliff Hall
Rabah Hamdi
Rodney Hanneman
Chris Hansen
Gopal Harikumar
Roy Harvey
Roy Harvey
Josef Hausner
Tom Haycock
Shahin Hedayat
Chris Heegard
Peter Niels Heller
Brian Henrichs
Malcolm Herring
Hanan Herzberg
Curt Hicks
Amir Hindie
Minnie Ho
David Hoerl
David Holien
Mahbub Hoque
James C. Horng
Gary R. Hoyne
Gang Huang
Les Humphrey
Marlis Humphrey
Cannon Hwu
Ishai Ilani
Greg Ioffe
Mikael Isaksson
Tomokazu Ito
Krista S. Jacobsen
Ken Jacobson
Charlie Jenkins
Ralph Jensen
Scott Jezwinski
Jim Jollota
Albin Johansson
David C. Jones
Edward Jones
Ragnar Jonsson
Anjal Joshi
John Joyce
Vern Junkmann
Wen-Juh Kang
Satoru Kawanago
Ken Kerpez
Kamran Khadavi
Babak Khalaj
Sayfe kiaei
Avi Kliger
Ron Knipper
Robert Kniskern
Ken Ko
Yosef Kofman
Jouni Koljonen
Hajime Koto
Tetsu Koyama
James Kroll
Philip J. Kyees
Robert LaGrand
T. K. Lala
Chi-Ying Lan
John Langevin
Martin LaRose
Steven C. Larsen
Mike Lassandrello
George J. Lawrence
Dong Chul Lee
Howard Levin
Gabriel Li
Haixiang Liang
Ze'ev Lichtenstein
Simon Lin
Jari Lindholm
Stan Ling
James Liou
Dave Little
Fuling Frank Liu
Qing Li Liu
Valentino Liva
G. W. (Wayne) Lloyd
Bob Locklear
Guozhu Long
Pini Lozowick
Perry Lu
Ahmed Madani
Rabih Makarem
Marcus Maranhao
Dan Marchok
Ron Marquardt
Doug W. Marshall
Al Martin
Kazuya Matsumoto
Bo Matthys
Thomas Maudoux
Jack Maynard
Gary McAninch
Kent McCammon
John McCarter
Shawn McCaslin
Ronald C. McConnell
Keith McDonald
Richard A. McDonald
Peter Melsa
Denis J. G. Mestdagh
Harry Mildonian
Dave Milliron
Khashayar Mirfakhraei
Steve Milkan
Cory Modlin
Michael Moldoveanu
Steven Monti
David R. Moon
Lane Moss
Kevin Mullaney
Joe Muller
Babak Nabili
Donovan Nak
Randy Nash
Frank Navavi
Gil Naveh
Gunter F. Neumeier
Mai-Huang Nguyen
Ramin Nobakht
Andy Norrell
Rao Nuthalapati
Stephen Oh
Franz Ohen
Hans Öhman
Yusaku Okamura
Kazu Okazaki
Vladimir Oksman
Al Omran
Mike O'Neill
Aidan O'Rourk
Tom O'Shea
Eric Paneth
Panos Papamichalis
Yatendra K. Pathak
Shimon Peleg
Michael Pellegrini
Matt Pendleton
Larry Perron
Todd Pett
Willie Picken
Ashley Pickering
Thierry Pollet
Michael Polley
Bob Poniatowski
Boaz Porat
Ron Porat
Carl Posthuma
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Philip Potter
Amit Preuss
Aleksandar Purkovic
Gordon Purtell
Dan Queen
Jim Quilici
Jack Quinnell
Ariel Radsky
Selem Radu
Sreen Raghavan
Ali Rahjou
Jeffrey M. Rakos
Avi Rapaport
Janice Rathmann
Dennis J. Rauschmajer
Richard Rawson
Gord Reesor
John Reister
Behrooz Rezvani
Ron Riegert
Terry Riley
Boaz Rippin
Jorge Rivera
Richard Roberts
Silvana Rodrigues
John E. Roquet
John Rosenlof
Eric J. Rossin
Mike Rude
Mark Russell
Christopher J. Rust
Kimmo K. Saarela
Ken Sakanashi
Debbi Sallee
Henry Samueli
Hal Sanders
Wayne Sanderson
Jamal Sarma
Sabit Say
Denny Schart
Kevin Schneider
Gary Schultz
Bob Scott
Linda Seale
Reuven Segev
Radu Selea
Ahmed Shalash
Mark Shannon
Donald P. Shaver
Greg Sherrill
Tzvi Shukhman
Eli Shusterman
Rex Siefert
Kevin Sievert
Richard Silva
Doug Silveira
Peter Silverman
Mark Simkins
Kamran Sistanizadeh
Don Skinfill
Joe Smith
P. Norman Smith
R. K. Smith
Stephen Smith
Edwin J. Soltysiak
Massimo Sorbara
Andrew Sorowka
Walt Soto
J. Scott Spradley
Paul Spruyt
Tom Starr
Mark Steenstra
William Stewart
James Stiscia
Jeff Strait
Caleb Strittmatter
Richard Stuart
Ray Subbankar
Henri Suyderhoud
James Szeliga
Hiroshi Takatori
Daryl C. Tannis
Larry Taylor
Matthew Taylor
Steve Taylor
Gary Tennyson
Rainer Thoenes
Vernon Tice
Ed Tirakian
Chi-Lin Tom
Antti Tommiska
J. Alberto Torres
Richard L. Townsend
Bob Tracy
Dwen-Ren Tsai
Marcos Tzannes
Masami Ueda
John Ulanskas
Juan Ramon Uribe
Peter Vaclavik
Craig Valenti
Nick van Bavel
Harry van der Meer
Frank Van der Putten
Dick van Gelder
Jeff Van Horne
M. Vautier
Robert L. Veal
Dale Veeneman
Rami Verbin
Pieter Versavel
Raman Viswanathan
Jeff Waldhuter
Josef Waldinger
Qi Wang
Brian Waring
Dewight Warren
Curtis Waters
Alan Weissberger
J. J. Werner
Rick Wesel
Greg Whelan
Albert White
Song Wong
Bernard E. Worne
Cliff Yackle
Han Yeh
Soobin Yim
Kyung-Hyun Yoo
Gavin Young
Irvin Youngberg
Xiaolong Yu
Shaike Zalitzky
Xuming Zhang
George Zimmerman
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
American National Standard
for Telecommunications
Spectrum Management
for Loop Transmission Systems
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
1.
Scope, purpose, and application
1.1
Scope
This standard provides spectrum management requirements and recommendations for the administration
of services and technologies that use metallic subscriber loop cables. Spectrum management is the
administration of the loop plant in a way that provides spectral compatibility for services and technologies
that use pairs in the same cable. In order to achieve spectral compatibility, the ingress energy that
transfers into a loop pair, from services and transmission system technologies on other pairs in the same
cable, must not cause an unacceptable degradation of performance. In addition, the egress energy from
a particular loop pair must not transfer into other pairs in a manner that causes an unacceptable
degradation in the performance of services and technologies on those pairs.
This standard includes the following types of requirements and recommendations for defined digital
subscriber line spectrum management classes and legacy systems:
-
power spectral density (PSD)
-
total average power
-
transverse balance
-
deployment guidelines
The standard also specifies a generic analytical method (Annex A) to determine the spectral compatibility
of loop technologies that do not qualify for one of the spectrum management classes defined in this
standard.
Requirements in this standard are specified for insulated solid copper conductor twisted-pair cables used
in the subscriber loop environment.
A system that fits in a Spectrum Management class complies with the Spectrum Management and
Spectral Compatibility requirements of this standard. A system that complies with Annex A complies with
the Spectrum Compatibility requirements of this standard. Compliance with a Spectrum Management
class provides knowledge of the characteristics of the loop system to aid deployment practices that reduce
the adverse impact to the basis systems. Not all basis systems conform to this standard.
DSL transmission systems that meet all of the specifications associated with one of the DSL spectrum
management classes are assumed to be spectrally compatible in the same binder group with all of the
basis systems defined in this standard. Meeting the specifications associated with one of the spectrum
management classes in this standard does not assure spectral compatibility with non-basis loop
transmission systems.
The requirements in this issue of this standard assume that the DSL system is deployed between a
Central Office (CO) and a customer installation (CI). Applications that locate the TU-C at an intermediate
point or applications that use intermediate repeaters can, in some cases, cause crosstalk that is greater
than those that use only a TU-C at the CO and a TU-R at the CI. Applications that locate the TU-C at an
intermediate point are beyond the scope of the guidelines in this issue of this standard.
Electromagnetic Compatibility (EMC) is outside of the scope of this standard. In addition, the spectrum
management of privately owned twisted-pair cables or customer premises twisted-pair cabling are beyond
the scope of this standard although the information in this standard may be useful in such applications.
The guidelines in this standard are based strictly on spectrum management requirements. It is
understood that a technology may have performance capabilities that either exceed or fall short of the loop
lengths specified for spectrum management.
The signals that the network and customer installation (CI) apply to the loop are basically of two types:
normal telecommunications transmission system voltages and currents, and voltages and currents due to
maintenance activities. The normal network and CI signals are addressed in this standard. Voltages and
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
currents due to network maintenance activities and abnormal voltages and currents that are the result of
the environment (e.g., induced voltages and currents or lightning) are not covered in this standard.
1.2
Purpose
The purpose of this standard is to facilitate a reasonable spectral environment for the co-existence of
multiple technologies in the loop plant with an acceptable level of crosstalk between them. When a single
carrier deploys technologies in loop plant, it alone has the responsibility for spectral compatibility and may
select any combination of compatible loop technologies. In an unbundled loop environment however,
multiple carriers utilize pairs in the same loop cables. In such instances, if services and technologies are
deployed without regard to spectral compatibility, they may interfere with each other. This standard
assumes that loop cables are shared by multiple carriers and that all carriers share the responsibility for
spectral compatibility.
This standard provides information that will help to ensure that twisted-pair transmission systems can coexist without impaired operation due to crosstalk interference. The standard is intended for use by
carriers to manage the loop plant and by manufacturers in the design of loop transmission systems.
This standard was also developed to assist carriers, manufacturers, and users of products to be
connected to local loops, to understand the characteristics of twisted-pair loop cables. In addition, this
standard can be used to determine if new services and loop transmission system technologies are
spectrally compatible with certain basis systems and technologies that are defined in this standard.
This standard is intended to be consistent with Part 68, Subpart D, of the FCC Rules and Regulations that
contains requirements for the registration of customer installation terminal equipment to protect the
network from harm. Some of the digital subscriber line spectrum management classes defined in this
standard are not covered by Part 68. If Part 68 rules are subsequently established for technologies that
fall into those categories, the requirements in this standard can be referenced. Tariffs, contracts, or
regulatory acts in various jurisdictions may contain requirements different from those in this standard.
The provisions of this standard are also intended to be consistent with applicable requirements concerning
safety and environmental conditions.
1.3
Application
This standard is applicable to twisted-pair cables that are used by multiple carriers in the local loop
environment.
All of the loops described in this standard may not be universally available. For example, a loop that
supports Basic Rate ISDN can only be provided if the facilities serving the CI are qualified to support such
technology.
Because of the wide range of network switching systems, network transport systems, subscriber loop
plant, and CIs in North America, conformance with this standard does not guarantee acceptable
performance under all possible operating conditions. In some cases, additional measures will be needed.
2.
Normative references
The following standards contain provisions that, through reference in this text, constitute provisions of this
American National Standard. At the time of publication, the editions indicated were valid. All standards
are subject to revision, and parties to agreements based on this standard are encouraged to investigate
the possibility of applying the most recent editions of the standards indicated below.
ITU-T Recommendation G.991.1, High Speed Digital Subscriber Line (HDSL) Transmission System on
Metallic Local Lines.
ITU-T Recommendation G.992.1, Asymmetrical Digital Subscriber Line (ADSL) Transceivers.
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ITU-T Recommendation G.992.2, Splitterless Asymmetrical Digital Subscriber Line (ADSL) Transceivers.
ANSI T1.413-1998, American National Standard for Telecommunications – Network and Customer
Installation Interfaces – Asymmetrical Digital Subscriber Line (ADSL) Metallic Interface.
BSR T1.418, High Bit Rate Digital Subscriber Line - 2nd Generation (HDSL2).
BSR T1.419, Splitterless Asymmetric Digital Subscriber Line (ADSL) Transceivers
ANSI T1.601-1998, American National Standard for Telecommunications – Integrated Services Digital
Network (ISDN) – Basic Access Interface for Use on Metallic Loops for Application on the Network Side of
the NT (Layer 1 Specification).
ANSI T1.403-1999, American National Standard for Telecommunications – Network and Customer
Installation Interfaces - DS1 Electrical Interface.
ANSI T1.410-1992, Carrier-to-Customer Metallic Interface - Digital Data at 64kb/s and Subrates.
EIA/TIA TSB-31-B, February 1998; Part 68 Rationale and Measurement Guidelines; Telecommunications
Industry Association, 1998.
Committee T1 Technical Report No. 59, Single Carrier Rate Adapative Digital Subscriber Line (RADSL).
3.
Definitions, abbreviations, acronyms, and symbols
3.1
Definitions
3.1.1.
american wire gauge: A unit used to measure the diameter of round wire.
3.1.2.
balance: See longitudinal balance and transverse balance.
3.1.3. basis system: A term used in this standard to describe a loop transmission system with which
DSL systems and other new loop transmission systems are required to demonstrate spectral
compatibility.
3.1.4. binder group: In this standard, the smallest cable unit consisting of a group of twisted pairs that
are wrapped with colored binders for identification and separation from other units.
3.1.5. bit error ratio: A performance measure consisting of the ratio of bits in error to the total number
of bits transmitted.
3.1.6.
carrier: An organization that provides telecommunications services to customer installations.
3.1.7.
central office: In this standard, the telephone building that is the origin of the outside loop plant.
3.1.8.
conductor: A continuous solid copper or aluminum wire that has a circular cross-section.
3.1.9. crosstalk: Electromagnetic energy that couples into a metallic cable pair from signals on other
pairs in the same cable.
3.1.10. customer installation: All cabling and equipment on the customer side of the network interface.
3.1.11. customer premises equipment: Telecommunications equipment located at the customer
installation on the customer side of the network interface.
3.1.12. demarcation point: See network interface.
3.1.13. disturbed pair: A cable pair that has a service or technology that is experiencing crosstalk
interference from one or more other pairs in the same cable.
3.1.14. disturbing pair: A pair with a signal that is contributing to crosstalk interference into a service or
technology on another pair in the same cable.
3.1.15. downstream: The direction of transmission from the carrier Central Office to the Customer
Installation.
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3.1.16. drop wire: A type of loop cable, consisting or one or more pairs, that is used between the loop
cable terminal and the network interface device.
3.1.17. equivalent working length (EWL):
EWL = L 26 +
3( L 24)
4
where L26 is the total length of 26 gauge cable in the loop excluding any bridged tap and L24 is the
total length of 19, 22 or 24 gauge cable in the loop excluding any bridged tap. All lengths are in kilofeet
(kft).
3.1.18. far-end crosstalk: Crosstalk that occurs when the receiver on a disturbed pair is located at the
other (far) end of the cable as the transmitter of a disturbing pair.insulated conductor: A conductor
that has been surrounded with insulation that is often color-coded.
3.1.19. insulation: The dielectric material that surrounds a conductor and prevents it from contacting
other conductive material.
3.1.20. longitudinal balance: Describes the degree of symmetry with respect to ground of a twoconductor transmission line. Longitudinal balance may be expressed as 20 times the log10 of the
magnitude of the ratio of an applied longitudinal voltage (referenced to ground) to the resultant
metallic voltage.
3.1.21. loop: A communication path between the distributing frame in a carrier Central Office and the
network interface at a customer location.
3.1.22. near-end crosstalk: Crosstalk that occurs when the receiver on a disturbed pair is located at the
same (near) end of the cable as the transmitter of a disturbing pair.
3.1.23. network: All equipment and facilities, including loop plant, located on the carrier side of the
network interface.
3.1.24. network interface: The physical demarcation point between carrier network loop facilities and the
CI.
3.1.25. non-basis system: A term used in this standard to describe a loop transmission system with
which DSL systems and other new loop transmission systems are not required to demonstrate
spectral compatibility.
3.1.26. pair: Two insulated conductors.
3.1.27. power spectral density (PSD): The power level and frequency content of a transmitted signal.
3.1.28. short-term stationary: A term used in this standard to describe a loop transmission system in
which an “ON” condition (in which the transmitter generates a signal) alternates with an “OFF”
condition (in which the transmitter is silent or generates only a pilot tone).
3.1.29. spectral compatibility: The capability of two loop transmission system technologies to coexist in
the same cable and operate satisfactorily in the presence of crosstalk noise from each other.
3.1.30. spectrum management: In this standard, the term refers to processes that are intended to
minimize the potential for interference and maximize the utility of the frequency spectrum of metallic
loop cables.
3.1.31. spectrum management class: In this standard, the term refers to the classes defined in 5.2,
classifying the technologies in terms of their PSD. Abbreviated SM class.
3.1.32. transverse balance: A comparison of the voltage of a transmitted metallic or transverse signal to
the voltage of any resulting longitudinal signal. See 6.5.
3.1.33. type I PSDS: A legacy loop transmission system based on 56 kbps digital data service that uses
AMI operating at 56 kbps on two loop pairs to provide a 4-wire full-duplex digital channel. Network
signaling is accomplished using bipolar patterns that include bipolar violations. For more information,
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see TIA/EIA-596.
3.1.34. type II PSDS: A legacy loop transmission system that functions in two modes: analog and digital.
Analog signaling is used to perform network supervisory and address signaling. The system is
switched to the digital mode after a connection is established. Type II PSDS uses Time Compression
Multiplexing and AMI operating at 144 kbps to provide a full-duplex 56 kbps service on a 2-wire loop.
For more information, see TIA/EIA-596.
3.1.35. type III PSDS: A legacy loop transmission system that uses Time Compression Multiplexing and
AMI operating at 160 kbps to provide two full-duplex digital channels on a 2-wire loop. One digital
channel is an 8 kbps signaling channel for supervisory and address signaling and the other is a 64
kbps data channel. For more information, see TIA/EIA-596.
3.1.36. twisted pair: A balanced transmission line consisting of two insulated conductors that have been
twisted together during the manufacturing process to reduce coupling to and from external circuits.
See balanced.
3.1.37. upstream: The direction of transmission from the Customer Installation to the carrier Central
Office.
3.1.38. voicegrade: A term used to qualify a channel, facility, or service that is suitable for the
transmission of speech, data, or facsimile signals; generally with a frequency range of about 300 to
3000 Hz.
3.1.39. working length: The sum of all cable segment lengths from the central office to the network
interface at a customer location, excluding non-working bridged taps.
3.2
Abbreviations, acronyms, and symbols
The following acronyms are used throughout this document.
2B1Q
Two Binary, One Quatenary
ADSL
Asymmetric Digital Subscriber Line
ANS
American National Standard
ANSI
American National Standards Institute
AWG
American Wire Gauge; see definition
BER
bit error ratio; see definition
Bps
bits per second
CAP
Carrierless Amplitude and Phase Modulation
CI
customer installation; see definition
CO
central office; see definition
CPE
customer premises equipment; see definition
CSA
carrier serving area
DB
decibel
DBm
decibel referenced to 1 milliwatt
DBrn
decibel referenced to noise
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DBrnC
decibel referenced to noise with C-message weighting
DDS
digital data service
DMT
Discrete Multitone
DSL
digital subscriber line
DUT
device under test
FCC
Federal Communications Commission
FEXT
far-end crosstalk; see definition
HDSL
high-bit-rate digital subscriber line
HDSL2
high-bit-rate digital subscriber line over a single pair
Hz
hertz
ISDN
Integrated Services Digital Network
ITU-T
International Telecommunication Union – Telecom Sector
KHz
kilohertz
L26
The total working length of 26 AWG cable on a loop
MH
millihenry
Ms
millisecond
NEXT
near-end crosstalk; see definition
NI
network interface; see definition
PAM
Pulse Amplitude Modulation
PSD
power spectral density
PSDS
public switched digital service. See definitions of type 1, type II, and
type III.
QAM
Quadrature Amplitude Modulation
RADSL
rate adaptive digital subscriber line
RLCG
resistance, inductance, capacitance, and conductance
RRD
revised resistance design
SM
Spectrum management, e.g., SM class 1. See definition.
T1
type of 4-wire metallic 1.544 Mbps transmission system
TU-C
Transceiver Unit – Central office end. Sometimes combined with
another letter; e.g., ATU-C for a central office ADSL transceiver
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TU-R
Transceiver Unit – Remote terminal end. Sometimes combined with
another letter; e.g., ATU-R for a remote ADSL transceiver
VDSL
very-high-bit-rate digital subscriber line
4.
General Information
Most of the subscriber loop plant in North America consists of metallic cables that were designed primarily
for voicegrade services. Several other types of services and technologies use these loop cables however
including, but not limited to, digital data services, T1-carrier systems, and Digital Subscriber Line (DSL)
transmission systems.
Metallic loop cables generally contain several solid copper conductors that are circular in cross-section.
Each conductor is surrounded by insulation that is usually color-coded. During manufacturing, pairs of
insulated conductors are twisted together. Several twisted pairs are then assembled together into units
called binder groups that are bound with colored tape for identification.
The signals that are transmitted on a loop cable pair create an electromagnetic field that surrounds nearby
pairs and induces voltages into those pairs. The twisting of the insulated conductors into pairs minimizes
this coupling as does the bundling of pairs into binder groups. Despite these measures however, a
capacitive coupling still exists between the pairs of a multipair loop cable.
This clause provides general information about crosstalk interference in metallic loop cables, the spectral
compatibility of loop transmission systems, and various aspects of spectrum management. Clause 5
provides signal power limitations and deployment guidelines for the DSL spectrum management classes.
Conformance testing methodology is provided in clause 6.
4.1
Crosstalk
The electromagnetic energy that couples into a metallic cable pair from services and transmission system
technologies on other pairs in the same cable is unwanted energy and is called crosstalk noise or simply
“crosstalk”. Crosstalk may, or may not, be disturbing. When crosstalk causes an unacceptable
degradation in the performance of victim services or technologies in the same cable, it is called crosstalk
interference.
Preventing crosstalk interference requires the careful manufacturing, installation,
maintenance, and administration of loop cables.
Crosstalk is sensitive to frequency, signal strength, and exposure. High frequency energy couples into
other pairs more than low frequency energy because as the signal frequency increases, the crosstalk
coupling loss between the pairs of a cable decreases. Hence, for two signals of equal strength, the higher
the frequency, the greater the crosstalk noise.
A strong signal will transfer more power into other pairs than will a weaker signal. The amount of
crosstalk noise is directly proportional to the power of the disturbing signal. The stronger the disturbing
signal, the greater the crosstalk noise. Thus, one of the most effective means of controlling crosstalk
noise is to limit the signal energy that is applied to cable pairs. Signal power limitations for several DSL
classes are provided in clause 5.
Exposure is a measure of the proximity of metallic pairs at various points along a cable and the length
over which pairs are in close proximity. The greater the exposure, the greater the total crosstalk noise.
Since it is impossible to predict the exact amount of exposure between any two pairs in a cable, statistical
exposure models are used for the crosstalk margin evaluations described in Annex A. In this standard, it
is assumed that all loops in a binder are of same length; it is known that this could cause additional
degradation. For example, different loop lengths can result from feeder-distribution cross-connection.
Crosstalk noise that occurs when a receiver on a disturbed pair is located at the same end of the cable as
the transmitter of a disturbing pair is called Near-End-Crosstalk (NEXT). Crosstalk noise that occurs when
a receiver on a disturbed pair is located at the other end of the cable as the transmitter of the disturbing
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
pair is called Far-End-Crosstalk (FEXT). NEXT coupling is generally greater than FEXT coupling when
transmission takes place in both directions in a binder and there is an overlap between the upstream and
downstream signals.
4.2
Spectral compatibility
In general, spectral compatibility is the capability of two loop transmission system technologies to coexist
in the same cable and operate satisfactorily in the presence of crosstalk noise from each other.
A loop transmission system technology is considered to be spectrally compatible with other loop
transmission systems when:
a) It meets the signal power limits, the deployment guidelines and other criteria for one or more
of the Spectrum Management classes defined in clause 5 of this standard.
Or
b) It meets the criteria of the analytical method defined in Annex A of this standard.
This standard does not explicitly define significant service degradation.
4.3
Spectrum management
In this standard, the term spectrum management refers to processes that are intended to minimize the
potential for crosstalk interference and maximize the utility of the frequency spectrum in multipair metallic
loop cables.
The spectrum management requirements and recommendations in this standard include signal power
limitations, technology deployment guidelines, and a generic analytical method that can be used to define
new DSL spectrum management classifications or determine the spectral compatibility of different
technologies. The requirements and recommendations in this standard are intended to provide spectral
compatibility with certain defined basis loop transmission systems and thereby maximize the use of the
bandwidth provided by metallic loop cables.
4.3.1
1
Basis loop systems
Basis systems, defined as loop transmission systems with which the DSL spectrum management classes
2
defined in this standard and other new loop transmission systems , are required to demonstrate spectral
compatibility. The basis systems are systems that are currently deployed.
It is not necessary, nor sufficient, for a system to be on the list of basis systems for the system to be
compliant with this standard. The list of basis systems is a living list; new systems may be added to the
list, and eventually systems may be retired from the list when the need to guard a system has passed its
usefulness. To avoid an excessive impediment to potential new technologies and to simplify the
Spectrum Management Standard, it is highly desirable to include in the list of basis systems only those
systems that have the greatest total impact on the population of subscriber line users. To be included in
the list of basis systems, the following factors shall apply:
1) It is highly preferred that the system be standardized by the ITU or an ANSI accredited standards
organization or that a draft standard is expected to be approved by the time the forthcoming issue of
the Spectrum Management is expected to be published. If an effort has been made to standardize
the system and there is a clear reason why the system can not be standardized or completed in a
timely manner, then a physical-layer specification shall be publicly available.
–––––––
1
Basis systems are not defined or intended for the purpose of resolving interference disputes.
2
This includes high bandwidth CPE-based systems.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
2) The specification for a non-standard system shall be stable, widely accepted by most of the industry,
and shall specify all aspects necessary to determine spectral compatibility (e.g. transmitted signal
PSD, modulation method, coding, bit-rate, start-up process, and margin to be achieved for certain
reference loops and reference noise).
3) Preferably, a new basis system should not require changes to the existing Spectrum Management
Classes to maintain spectral compatibility with the new basis system.
4) Preferably, a new basis system should not be adequately addressed by the existing systems on the
basis system list.
5) New basis systems should demonstrate possible scenarios where the new system could be disturbed
while other basis systems are not.
In order to assure spectral compatibility with the anticipated mix of current and future technologies on loop
3
binder groups, this standard has defined a set of loop transmission basis systems as to which spectral
compatibility shall be demonstrated:
4
-
Voicegrade services .
-
Enhanced Business Services (P-Phone) based on xxx.
-
Digital Data Service (DDS) based on T1.410.
-
Basic Rate Integrated Services Digital Network (ISDN) based on T1.601. Note that this includes 2channel digital systems (UDC-2) based on ISDN technology.
-
High-Bit-Rate Digital Subscriber Line (HDSL) based on G.991.1, Annex A.
-
Asymmetrical Digital Subscriber Line (ADSL) based on T1.413-1998 with non-overlapped
upstream/downstream mode.
-
RADSL based on Committee T1 Technical Report No. 59.
-
Splitterless ADSL based on BSR T1.419. G.992.2 with non-overlapped upstream/downstream
mode
-
Repeatered T1 (1.544 Mbps) technology based on T1.403
-
HDSL2 (DS1 payload on single pair) based on the BSR T1.418.
-
2B1Q SDSL @ 400 kb/s, 1040 kb/s and 1552 kb/s.
This set is defined to take into account: 1) voluntary DSL standards based on industry consensus and
open specifications and 2) several legacy loop transmission systems. Spectral compatibility – generally in
the same binder group - with the basis systems listed above shall be demonstrated by meeting all of the
signal power limitations and other criteria for one of the DSL spectrum management classes defined in
clause 5 (Method A).
See Section 5.2 for further information regarding the spectral compatibility of basis systems.
4.3.1.1 Voicegrade services
Voicegrade services include speech, network signaling, data, and tone signals that use the frequency
spectrum from 0 to 4 kHz. (See Annex A.)
–––––––
3
Very-high-speed Digital Subscriber Line (VDSL) and G.SHDSL technology are currently in the standards
development process. VDSL and G.SHDSL are expected to be added to the basis system list. In anticipation of
this, some accommodation of VDSL and G.SHDSL has been made in the signal power limitations for the spectrum
management classes defined in clause 5.
4
Voicegrade services include speech, data, and tone signals that use the frequency spectrum from 0 to 4 kHz. For
more information, see Annex A.
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4.3.1.2 Enhanced Business Services (P-Phone)
Enhanced Business Services use the frequency spectrum from 0 to 10 kHz and are used to transport
speech signals in the same way as done by traditional voicegrade services. A digital signalling channel
centered around 8 kHz allows the performance of all functions associated with the setting up and tearing
down of voice calls without the use of high voltage signalling.
4.3.1.3 Digital Data Services (DDS)
Digital Data Services, based upon T1.410, operate at 64 kb/s and subrates of 2.4, 4.8, 9.6, 19.2 38.4, 56
kb/s. Secondary channel services are also available for all subrates. While all DDS subrates and
subrates with secondary channels are basis systems, the DDS analytical evaluation procedure in Annex A
focuses on 56 kb/s and 64 kb/s DDS in order to reduce the number of DDS evaluations that a new system
must undergo. Since Type 1 Public Switched Digital Service (PSDS) uses the same physical layer as 56
kb/s DDS, any new technology that demonstrates compatibility with 56 kb/s DDS will also be compatible
with Type 1 PSDS.
4.3.1.4 Basic Rate ISDN (BRI)
In the context of this standard, BRI represents a family of basis loop transmission systems that uses the
transceiver technology described in T1.601. The family includes traditional BRI that uses the ISDN data
link layer protocols described in T1.602 as well as other systems that have adapted the T1.601 layer 1
transceiver technology for use as:
-
a packet network access system (IDSL)
-
a point-to-point transport system sometimes referred to as a Universal Digital Channel (UDC).
BRI, IDSL, and UDC are defined in this standard as systems that use the 2B1Q line code, operate at 80
kbaud for transmission at 160 kbps, and may be transported via DLC by using BRI termination extension
(BRITE) devices. The entire BRI family is a basis system. The analytical method for demonstrating
compatibility with BRI in Annex A does not differentiate between the members of the BRI family and
adequately guards all members of the family.
4.3.1.5 High-Bit-Rate Digital Subscriber Line (HDSL)
HDSL systems are designed to transport 784 kbps over Carrier Serving Area (CSA) distances on a single
non-loaded twisted pair. The most common application transports a 1.544 Mbps payload on two nonloaded twisted pairs but some applications may use a single pair. Some HDSL applications extend the
reach by the use of intermediate repeaters. Basis HDSL systems are echo canceller hybrid systems that
use the 2B1Q line code and operate at 392 kbaud. The analytical method for demonstrating compatibility
with HDSL in Annex A does not differentiate between one pair and two pair applications.
4.3.1.6 HDSL2
HDSL2 is a second generation HDSL loop transmission system that is currently in the standards
development process. The system is designed to transport a 1.544 Mb/s payload on a single non-loaded
twisted pair at Carrier Serving Area (CSA) distances.
4.3.1.7 ADSL, RADSL, and Splitterless ADSL
The basis asymmetrical DSL systems operate using different frequency bands (non-overlapped) for
upstream and downstream operation. The analytical method for demonstrating compatibility with these
systems in Annex A is described in terms of the relevant line code (i.e., DMT, CAP, or QAM).
4.3.1.8 2B1Q SDSL
2B1Q SDSL uses 4-PAM modulation. Symbol rate, baud rate, and power spectrum density at both HTUC
and HTUR transceivers are the same. 2B1Q SDSL system may vary its data rate from 64 kb/s to 2320
kb/s, with granularity of data rate of greater than or equal to 8 kb/s. 2B1Q SDSL at 160kb/s and 784kb/s
are represented by basic rate ISDN and HDSL, respectively.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
4.3.2
Legacy systems
Newly deployed loop services and technologies may encounter a loop environment that includes one or
more legacy systems. A legacy system is a loop service or technology that was defined many years ago
and is nearing the end of its life cycle.
The following services and technologies are legacy systems. However, this list may not be all inclusive:
-
15 kHz Program Audio services
-
Type II PSDS
-
Type III PSDS
-
Local Area Data Channels
-
Data-Over-Voice services and technologies
-
Analog Carrier technologies
The legacy systems listed above were not addressed during the development of this standard.
4.3.3
Signal power limitations (method A)
Since strong signals transfer more power into other pairs than weaker signals, the most widely used and
most successful method of controlling crosstalk interference and achieving spectral compatibility is
through the use of signal power limitations. Signal power limitations specify the amplitude, frequency
distribution, and total power of electrical signals at the point where the signal enters the subscriber loop
cable.
For all DSL spectrum management classes addressed in this standard, Clause 5 defines signal power
limits. The requirements apply to signals transmitted by DSL transceiver units whether located in a
Central Office (TU-C) or a remote terminal location (TU-R). The remote terminal location is usually on or
near the customer premises.
The set of spectrum management classes is a living list; new classes may be added to the list and
eventually classes could be retired from the list when there is widespread agreement that a class is no
longer desirable or useful. To simplify the Spectrum Management process and this Standard, it is
desirable that the number of spectrum management classes be no larger than necessary.
A new spectrum management class may be added if the following five conditions are satisfied:
1) The new class is fully specified.
2) The new class is spectrally compatible with all basis systems, per Annex B.
3) The spectral compatibility with well know non-basis systems that are members of the existing
spectrum management classes has been investigated and the committee responsible for
development of the standard agrees that the impacts are acceptable.
4) The committee responsible for development of the standard agrees that the new spectrum
management class is needed.
5) Preferably, a new class should offer substantial benefits beyond the existing classes.
example:
For
a) Have a deployment guideline (in 500-foot steps) more than 10% different from existing similar
classes.
b) Enable members of the class to achieve the same bit rate and loop reach as members of the
existing classes while reducing the SNR margin impact of crosstalk to a guarded system by at
least 10%, and causing no more crosstalk impact to all other guarded systems.
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TU-C and TU-R equipment that meets the signal power limitations and other criteria for one of the DSL
spectrum management classes defined in clause 5 is expected to achieve spectral compatibility – in the
same binder group unless otherwise specified - with the basis transmission systems defined in this
standard.
The characterization of a transmitted signal by power level and frequency content is called the power
spectral density (PSD) of the signal. The primary signal power requirements in this standard are specified
through the use of PSD masks and templates. The PSD mask shows the maximum power boundary or
limit, in dBm per Hz, for the transmitted signal. The use of the PSD masks and templates is described
more fully in 6.1, 6.2, and 6.3.
4.3.3.1 Transceiver unit – remote terminal end (TU-R)
Part 68 of the FCC Rules and Regulations contain mandatory signal power limits for several types of
customer premises equipment (CPE) including voice, voiceband data, DDS subrates, public switched
digital services (PSDS), ISDN, local area data channel (LADC), and DS1. Clause 5 of this standard
defines signal power limits for several DSL spectrum management classifications that are not currently
covered by Part 68.
The TU-R equipment used with DSL systems is usually CPE. However, in some cases it may be network
equipment. The TU-R signal power limits in clause 5 shall be applicable regardless of whether or not the
TU-R is network equipment or CPE. Any TU-R that transmits a signal into a metallic loop cable shall meet
the relevant upstream signal power limitations and other criteria associated with one of the DSL spectrum
management classifications defined in section 5.
4.3.3.2 Transceiver unit – central office end (TU-C)
Historically, carriers have controlled the transmitted signal power of network elements through the
development and use of voluntary industry standards related to particular technologies. Clause 5 of this
standard defines signal power limits for several DSL spectrum management classifications. The DSL
classifications defined in clause 5 are based on the industry’s current view of requirements for spectrum
management.
The TU-C is network equipment. Any TU-C that transmits a signal into a metallic loop cable shall meet
the relevant downstream signal power limitations and other criteria associated with one of the DSL
spectrum management classifications defined in section 5.
4.3.4
Technology deployment guidelines
Some loop transmission system technologies can be deployed in a manner that substantially increases
the likelihood of crosstalk interference. To prevent interference in such instances, it is necessary to
adhere to certain technology deployment guidelines in addition to signal power limitations.
Technology deployment guidelines, if applicable, have been provided along with the signal power
limitations for each of the DSL spectrum management classes defined in clause 5. Any service or loop
transmission system that meets the signal power limitations for one of the DSL spectrum management
classes defined in this standard shall be deployed according to the relevant deployment guidelines that
are specified for that DSL spectrum management class in clause 5.
4.3.4.1 Deployment guidelines
Deployment guidelines constrain the way loop transmission systems are operated so that the assumptions
on which spectral compatibility was determined will remain valid. Deployment guidelines may include such
things as loop reach guidelines or prohibitions against spectrum management class 5 systems using
reverse upstream/downstream operation. If a DSL spectrum management class has applicable
deployment guidelines, they shall be specified in clause 5.
4.3.4.2 Binder Group Considerations
Binder group integrity is not always maintained in the loop plant. Technologies that demonstrate spectral
compatibility by using the analytical method in Annex A (Method B) shall not rely upon binder group
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
separation in order to achieve full compatibility with any basis transmission system. However, this
standard does not preclude the use of binder group separation.
4.3.5
Analytical method of determining spectral compatibility (method B)
It is recognized that future technologies may transmit signals that do not conform to the signal power
specifications for one of the spectrum management classes defined in clause 5, but which might still be
spectrally compatible with the basis systems listed in clause 4.3.1. In order to nurture innovation in the
development of new technologies which further maximize the utility of the copper loop plant, an analytical
method for evaluating new technologies is provided in Annex A.
This method (referred to as Method B) involves the computation of signal to noise margins for basis
systems, and provides an industry-approved method of determining the spectral compatibility of any loop
transmission system with the basis loop transmission systems defined in this standard. For each of these
basis systems, Annex A provides the specific NEXT margin formulas, evaluation loops, and defined
crosstalk environments required by Method B.
The analytical method in Annex A should be used to develop new signal power limits and deployment
guidelines for new DSL spectrum management classes. It is expected that this analytical method will also
be used to provide guidance during new system development. However, as noted in 4.3, such use could
lead to the introduction of several new technologies that would be compatible with basis systems but not
necessarily compatible with each other. Therefore, system developers are encouraged to bring new DSL
technologies that do not fit into existing spectrum management classes into the formulating group for this
standard, so that the creation of a new class and any associated deployment guidelines can be
considered. Other processes, such as the disclosure of verifiable methods to assess spectral compatibility
with the new technology, may also help avoid the uncoordinated introduction of new technologies that
could result in crosstalk interference.
The telephone loop plant consists of 12, 13, 25, 50, and 100 pair binder group cables. This standard
employs a 50 pair binder group model for the analysis of spectral compatibility.
4.3.5.1 Margin computations
Margin computations determine the crosstalk margin in decibels (dB). Each basis system should have the
–7
unless
margin specified for that system in Annex A. The margin shall be calculated with BER ≤ 10
otherwise specified. Margin is a function of many variables including:
a)
Crosstalk coupling loss,
b)
Loss characteristics of loop cables,
c)
Characteristics of the disturbed signal,
d)
Receiver technology of the disturbed system, and
e)
Characteristics of the disturber signal.
Annex A provides the margin formula and the information associated with item a) thorough d). The user
will have to supply the information for item e).
The configuration in Figure 1 shall be used when the effect of system B NEXT and FEXT interference into
a system A downstream receiver is evaluated and system A has the longer loop reach.
This simulation set-up assumes that all of the head-end transmitters (ATU-C, HTU-C, etc) of both systems
are co-located at a central location and that the distance-limited system B does not use range-extending
repeaters. It is also assumed that all of the system B upstream transmitters are co-located at the longest
supported loop length. This gives a worst case view of the effect of system B interference upon the
operation of system A.
The first cable section is adjusted to cover the maximum reach distance of system B, and the second
cable section is adjusted to cover the remaining length of any test loop under consideration. The system
B FEXT noise generator shall generate FEXT noise equivalent to a system B output signal passed
through the FEXT coupling loss, with coupling length equal to the first cable section, and through the
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whole cable from the system B transmitter location to the system A receiver location (sum of first and
second sections).
The configuration in Figure 2 shall be used to simulate the effect of system B interference onto the
upstream operation of system A.
This simulation set-up assumes that all of the head-end receivers (ATU-C, HTU-C, etc) of both systems
are co-located at a central location. In this case, the system B FEXT noise generator shall generate FEXT
noise equivalent to a system B output signal passed through the FEXT coupling loss and through the
cable section from the system B transmitter location to the system A receiver location (first cable section
only).
4.3.5.2 Evaluation loops
For each basis system, Annex A provides a set of loops that shall be used for analytical evaluations.
5.
Signal power limits and other criteria
Crosstalk noise is controlled primarily through the use of signal power limits that consist of Power Spectral
Density (PSD) limitations and total average power limitations. Additional criteria, such as transverse
balance requirements and deployment guidelines, are also important. This clause provides all of these
specifications for the DSL spectrum management classes. The conformance testing methodology in
clause 6 shall be used to determine compliance with the requirements in this clause.
DSL transmission systems that meet the PSD limitations, total average power limitations and transverse
balance requirements for one of the DSL spectrum management classes defined in this clause shall be
considered spectrally compatible with all basis systems if they are deployed according to the applicable
deployment guidelines that are specified in this clause. The deployment guidelines for some DSL classes
limit the distance that a system can operate at in order to ensure that crosstalk from systems in that class
will not impair the basis systems.
A multirate DSL system shall be considered spectrally compatible if it is deployed according to the
applicable deployment guidelines associated with the class for which it is configured.
5.1
Short-term stationary systems
Some types of DSL transmitters operate in transmission modes in which an “ON” condition (in which the
transmitter generates a signal) alternates with an “OFF” condition (in which the transmitter is silent or
generates only a pilot tone). Examples of such transmitters include burst transmission systems and
systems that use quiescent modes to reduce power consumption during idle data periods. Such
transmitters are referred to as “short-term stationary,” since during the ON condition the transmitted signal
has the same effect as a stationary (or cyclo-stationary) signal when observed over an appropriately short
time interval. Due to the relative frequency of ON/OFF and OFF/ON transitions in short-term stationary
transmitters, additional conformance criteria are applied to these transmitters. Additionally, there shall not
be any intentional synchronization of transmission bursts of short-term stationary systems.
Clause 6.4 defines a test to determine whether short-term stationary conformance criteria shall be applied
to a DUT and defines the short-term stationary conformance criteria.
5.2
Spectrum management classes
5.2.1
Spectrum management class 1
Spectrum management class 1 is intended for DSL transmission systems that operate in the frequency
spectrum up to about 115 kHz, including most, but not all, T1.601 compliant systems.
5.2.1.1 Spectrum management class 1 PSD and total average power limitation
Spectrum management class 1 TU-C and TU-R equipment shall meet the PSD conformance criteria in
6.1 using the PSD template described in Table 1 and Figure 3. The total average power into 135 Ohms
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
and below 115 kHz that is transmitted by the spectrum management class 1 TU-C and TU-R equipment
shall be 14.0 dBm or less.
5.2.1.2 Spectrum management class 1 transverse balance requirement
The transverse balance of spectrum management class 1 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 135-ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 1 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.1.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 1 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.1.4 Spectrum management class 1 deployment guidelines
Non-repeatered loop transmission systems that meet the signal power and transverse balance
requirements associated with Spectrum Management Class 1 may use any non-loaded loop facility and
may be assigned to pairs that are in the same binder group as any of the basis systems defined in this
standard.
5.2.2
Spectrum management class 2
Spectrum management class 2 is intended for DSL transmission systems that operate in the frequency
spectrum from 0 to about 238 kHz.
5.2.2.1 Spectrum management class 2 PSD and total average power limitation
Spectrum management class 2 TU-C and TU-R equipment shall meet the PSD conformance criteria in
section 6 using the PSD template described in Table 3 and Figure 4.
The total average power below 238 kHz that is transmitted by Spectrum Management Class 2 TU-C and
TU-R equipment shall be 14.0 dBm or less.
5.2.2.2 Spectrum management class 2 transverse balance requirement
The transverse balance of spectrum management class 2 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 135-ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 2 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.2.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 2 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.2.4 Spectrum management class 2 deployment guidelines
Spectrum management class 2 DSL transmission systems shall use non-loaded loop facilities. Class 2
systems are spectrally compatible with the basis systems in the same binder group for those loops with an
equivalent working length of less than TBD kilofeet. To assure acceptable performance of basis systems,
loop length guidelines may be needed for this spectrum management class. The specification of loop
length guidelines for this class is expected to be provided in a future version of this standard.
16
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
5.2.3
Spectrum management class 3
Spectrum management class 3 is intended for DSL transmission systems that operate in the frequency
spectrum up to about 370 kHz.
5.2.3.1 Spectrum management class 3 PSD and total average power limitation
Spectrum management class 3 TU-C and TU-R equipment shall meet the PSD conformance criteria in
section 6 using the PSD template described in Table 4. At frequencies at or below 1.05 MHz, linear
interpolation of the frequency and PSD entries of Table 4 is used to define the template. At frequencies
above 1.05 MHz, the template is –143 -10log10(f1.5/1.134x1013). The template is shown graphically in
Figure 5.
The total average power below 370 kHz that is transmitted by spectrum management class 3 TU-C and
TU-R equipment shall be 14.0 dBm or less.
5.2.3.2 Spectrum management class 3 transverse balance requirement
The transverse balance of spectrum management class 3 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 135-ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 3 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.3.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 3 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.3.4 Spectrum management class 3 deployment guidelines
Spectrum management class 3 DSL transmission systems shall use non-loaded loop facilities. Based
upon a conservative model using the 1% Unger near-end crosstalk coupling model (see Figure A. 1), with
24 disturbers, 6 dB margin, co-located CPE, and the assumed acceptable performance objectives for the
basis systems, Class 3 systems are spectrally compatible with the basis systems in the same binder
3
group for those loops with a working length of less than CSA reach. This is a provisional value and may
be modified in a future version of this standard.
5.2.4
Spectrum management class 4
Spectrum management class 4 class is intended to include standard compliant HDSL2 equipment and
other DSL transmission systems that have TU-C equipment that operates in the frequency spectrum up to
about 440 kHz and TU-R equipment that operates in the frequency spectrum up to about 300 kHz.
5.2.4.1 Spectrum management class 4 PSD and total average power limitation
Spectrum management class 4 TU-C equipment shall meet the PSD conformance criteria in section 6.2.4
–––––––
3
CSA-reach is defined as a loop distance that meets Carrier Serving Area (CSA) length guidelines but not the CSA
restrictions on bridged tap and the number of different gauges. Thus, the working length of a CSA-reach loop is
within CSA range (9 kft of 26 AWG or 12 kft of 24, 22, or 19 AWG) but the length of the bridged tap and the total
cable length including bridged tap may exceed CSA guidelines. The working length of a CSA-reach multi-gauge
cable that contains 26 AWG cable may not exceed 12 kft minus the length of the 26 AWG cable in kft divided by
three [12 kft – (L26 ÷ 3)]. Deployment is limited here on the basis of crosstalk impact. Bridged tap has very little
effect on the power of disturbing crosstalk. This is not the same as limiting the transmission range of a system
based on performance, which can be noticeably affected by bridged tap. Similarly, multiple gauge changes have
very little effect on crosstalk power.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
using the downstream PSD mask described in Table 5 and Figure 6. Spectrum management class 4 TUR equipment shall meet the PSD conformance criteria in section 6.2.4 using the upstream PSD mask
described in Table 6 and Figure 7.
The total average downstream power (into 135 Ohms) below 450 kHz that is transmitted by the spectrum
management class 4 TU-C shall not exceed 17.3 dBm. The total average upstream power (into 135
Ohms) below 350 kHz that is transmitted by the spectrum management class 4 TU-R shall not exceed
17.0 dBm.
5.2.4.2 Spectrum management class 4 transverse balance requirement
The transverse balance of spectrum management class 4 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 135-ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 4 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.4.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 4 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.4.4 Spectrum management class 4 deployment guidelines
Spectrum management class 4 DSL transmission systems shall use non-loaded loop facilities. Based
upon a conservative model using the 1% Unger near-end crosstalk coupling model (see Figure A. 1), with
24 disturbers, 6 dB margin, co-located CPE, and the assumed acceptable performance objectives for the
basis systems, Class 4 systems are spectrally compatible with the basis systems in the same binder
3
group for those loops with an equivalent working length of less than CSA reach. This is a provisional
value and may be modified in a future version of this standard.
5.2.5
Spectrum management class 5
Spectrum management class 5 is intended for DSL transmission systems that have TU-C equipment that
operates in the frequency spectrum from about 138 25 kHz to about 1104 kHz and TU-R equipment that
operates in the frequency spectrum from about 25 kHz to about 138 kHz.
5.2.5.1 Spectrum management class 5 PSD and total average power limitation
Spectrum management class 5 TU-C equipment shall meet the PSD conformance criteria in section 6
using the reduced-NEXT downstream downstream PSD mask template defined in Table 7in Annex F of
T1.413-1998 and TR-59.
Spectrum management class 5 TU-R equipment shall meet the PSD conformance criteria in section 6
using the upstream PSD mask template defined in Table 11T1.413-1998 and TR-59.
The total average downstream power between 138 25 kHz and 1104 kHz that is transmitted by the
spectrum management class 5 TU-C shall not exceed 19.920.9 dBm.
The total average upstream power below 138 kHz that is transmitted by the spectrum management class
5 TU-R shall not exceed 12.513 dBm.
5.2.5.2 Spectrum management class 5 transverse balance requirement
The transverse balance of spectrum management class 5 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 100 ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 5 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
5.2.5.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 5 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.5.4 Spectrum management class 5 deployment guidelines
Spectrum management class 5 DSL transmission systems shall use non-loaded loop facilities. Nonrepeatered spectrum management class 5 systems may be assigned to pairs that are in the same binder
group as any of the basis systems.
Spectrum management class 5 systems shall not be deployed in the following modes:
-
Overlapping downstream PSD mode defined in T1.413 that allows the TU-C to transmit significant
downstream power in the 25 kHz to 138 kHz frequency band.
-
Power boost mode described in the first version of the ADSL standard (ANSI T1.413-1995).
-
Transceivers located at the customer end of the loop transmitting in the downstream frequency
band (18825 - 1104 kHz). This does not preclude adjacent collocation configurations, but such
configurations should use a dedicated binder.
5.2.6
Spectrum management class 6
Spectrum management class 6 is intended for DSL transmission systems that operate in the frequency
spectrum up to about 10 - 20 MHz.
5.2.6.1 Spectrum management class 6 PSD and total average power limitation
Spectrum management class 6 TU-C and TU-R equipment shall meet the PSD conformance criteria in
section 6 using a PSD template (or templates) that are TBDshall address both CO and remote
deployments. The spectrum management class 6 PSD template should be based on emerging VDSL
standards, which were not completed in time for this issue of this standard. Spectrum management class
6 should be frequency-division duplex (FDD), with distinct PSD templates for upstream and downstream
transmission. There may also be distinct PSD templates for symmetric spectrum management class 6
systems and for asymmetric spectrum management class 6 systems.
The total average power that is transmitted by spectrum management class 6 TU-C and TU-R equipment
shall be 11.5 dBm or less.
5.2.6.2 Spectrum management class 6 transverse balance requirement
The transverse balance of spectrum management class 6 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 100 ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 6 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band. Above 3 MHz, the transverse balance requirements is
TBDshall address both CO and remote deployments.
5.2.6.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 6 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
5.2.6.4 Spectrum management class 6 deployment guidelines
Spectrum management class 6 DSL transmission systems shall use non-loaded loop facilities. Unlike
other DSLs, spectrum management class 6 systems were created to offer high bit rates over short ranges
when deployed from remote fiber-fed terminals, pedestals, or cases. Deployment guidelines for spectrum
management class 6 systems shall address both CO and remote deployments. Class 6 systems are
spectrally compatible with the basis systems in the same binder group for those loops with an equivalent
working length of less than TBD kilofeet.
5.2.7
Spectrum management class 7
Spectrum management class 7 is intended for DSL transmission systems that operate in the frequency
spectrum from 0 to about 776kHz.
5.2.7.1 Spectrum management class 7 PSD and total average power limitation
Spectrum management class 7 TU-C and TU-R equipment shall meet the PSD conformance criteria in
section 6 using the PSD template described in Table 8 and Figure 8.
The total average power below 776kHz that is transmitted by spectrum management class 7 TU-C and
TU-R equipment shall be 14.0dBm or less.
5.2.7.2 Spectrum management class 7 transverse balance requirement
The transverse balance of spectrum management class 7 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 100 ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 7 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.7.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 7 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.7.4 Spectrum management class 7 deployment guidelines
Spectrum management class 7 symmetric DSL transmission systems shall use non-loaded loop facilities.
Non-repeatered class 7 symmetric DSL transmission systems are spectrally compatible with the basis
systems in the same binder group for those loops with an equivalent working length of less than 7 kft
(provisional).
Note: The ITU-T currently has a project (G.shdsl) that addresses data rates similar to those intended for
this class. It is expected that this class will be superceded by a newer one class that reflects the outcome
of that effort. When the new class is defined in a future version of this standard, it is expected that any
new deployments using this class (Class 7) will not be compliant with the new version of this standard.
5.2.8
Spectrum management class 8
Spectrum management class 8 is intended for DSL transmission systems that operate in the frequency
spectrum from 0 to about 584kHz.
5.2.8.1 Spectrum management class 8 PSD and total average power limitation
Spectrum management class 8 TU-C and TU-R equipment shall meet the PSD conformance criteria in
section 6 using the PSD template described in Table 9 and Figure 9.
The total average power below 584kHz that is transmitted by the spectrum management class 8 TU-C
and TU-R equipment shall be 14.0dBm or less.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
5.2.8.2 Spectrum management class 8 transverse balance requirement.
The transverse balance of spectrum management class 8 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 100 ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 8 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.8.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 8 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
5.2.8.4 Spectrum management class 8 deployment guidelines
Spectrum management class 8 symmetric DSL transmission systems shall use non-loaded loop facilities.
Non-repeatered class 8 symmetric DSL transmission systems are spectrally compatible with basis
systems in the same binder group for those loops with an equivalent working length of less than TBD kft.
5.2.9
Spectrum management class 9
Spectrum management class 9 is intended for DSL transmission systems that have TU-C equipment that
operates in the frequency spectrum from about 25 kHz to about 1104 kHz and TU-R equipment that
operates in the frequency spectrum from about 25 kHz to about 138 kHz.
5.2.9.1 Spectrum management class 9 PSD and total average power limitation
Spectrum management class 9 TU-C equipment shall meet the PSD conformance criteria in section 6
using the downstream PSD template defined in Table 10.
Spectrum management class 9 TU-R equipment shall meet the PSD conformance criteria in section 6
using the upstream PSD template defined in.Table 11.
The total average downstream power between 25 kHz and 1104 kHz that is transmitted by the spectrum
management class 9 TU-C shall not exceed 20.420.9 dBm.
The total average upstream power below 138 kHz that is transmitted by the spectrum management class
9 TU-R shall not exceed 12.513 dBm.
5.2.9.2 Spectrum management class 9 transverse balance requirement
The transverse balance of spectrum management class 9 TU-C and TU-R equipment shall be measured
over the applicable frequency range using the procedures and 100 ohm measurement configuration
specified in clause 6. The transverse balance of spectrum management class 9 TU-C and TU-R
equipment shall exceed the values in Table 2 over the entire range of frequencies between the upper and
lower –20 dB points of the signal pass-band.
5.2.9.3 Longitudinal Output Voltage
The longitudinal output voltage of spectrum management class 9 TU-C and TU-R equipment shall be
measured over the applicable frequency range using the procedures and measurement configuration
specified in clause 6. The longitudinal output voltage in all 4 kHz frequency bands averaged over 1
second shall not exceed the values in Table 17 over the indicated range of frequencies. For this
requirement, the operating band is the range of frequencies between the upper and lower –30 dB points of
the signal pass-band. There is no requirement for frequencies below the operating band.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
5.2.9.4 Spectrum management class 9 deployment guidelines
Spectrum management class 9 DSL transmission systems shall use non-loaded loop facilities. Nonrepeatered spectrum management class 9 systems may be assigned to pairs that are in the same binder
group as any of the basis systems.
Spectrum management class 9 systems shall not be deployed in the following modes:
-
Power boost mode.
-
Transceivers located at the customer end of the loop transmitting in the downstream frequency band
(138-1104 kHz). This does not preclude adjacent colocation configurations, but such configurations
should use a dedicated binder.
-
Lines with equivalent working length greater than 13.5 kft.
5.3
Spectral Compatibility Limitations for Repeatered Systems
T1 is spectrally compatible with the basis systems in the adjacent binder group for loops with a working
length less than TBD kft. Repeatered HDSL is spectrally compatible with the basis systems in the same
binder group for loops with a working length less than TBD kft. It is expected that a future version of this
standard may provide additional specifications relating to repeatered systems.
6.
Conformance testing methodology
The conformance testing methodology in this clause shall be used to determine compliance with the
signal power limitations and transverse balance requirements in clause 5.
6.1
General conformance criteria
The conformance testing methodology is designed for the purpose of lab evaluation of the compliance of
equipment to the SM classes defined in Section 5. As explained in clause 6.3 and Table 13, Table 14 and
Table 15, PSDs are defined at a number of discrete points with resolution bandwidths as defined for each
SM class and each frequency. Let the PSD template of a SM class be denoted as PT(n) in units of
dBm/Hz, where 1 ≤ n ≤ N , let fr(n) denote the center frequency in kHz at which PT(n) is defined, and so
fr(N) is the highest frequency for which the PSD template PT(N) is defined. Unless otherwise stated, fr(N) =
30 MHz. The points of PT(n) are in order of increasing frequency so that fr(n) monotonically increases with
n. The resolution bandwidth is a function of the SM class and the frequency, and is denoted as BW (n)
kHz at frequency fr(n) kHz for point n as defined in clause 6.3 and Table 13, Table 14 and Table 15.
The PSD mask associated with a SM class is denoted as PM(n) dBm/Hz, and unless otherwise stated the
PSD mask is equal to the PSD template plus 3.5 dB, so that PM(n) = PT(n) + 3.5 dB.
The first step in the testing process is measurement of the transmitted PSD of the equipment under test,
which is done with the procedure described in clause 6.3. The result of the PSD measurement, in units of
dBm/Hz and at a center frequency of fr(n) kHz is denoted by Pa (n ) , and is recorded with resolution
bandwidth BW (n) kHz as defined for the appropriate SM class in clause 6.3 and Table 13, Table 14 and
Table 15.
PSD conformance is achieved by meeting all the following conditions:
a) For all n such that 1 ≤ n ≤ N , Pa (n ) ≤ PM (n ) , where PM (n ) is the PSD mask at frequency fr(n), and
PM (n ) = PT (n ) + 3.5dB .
b) For all integers m such that 1 ≤ m and such that M ≤ N (i.e., for all possible 100 kHz sliding
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
 M
10 Pa (n ) 10

BW (n) P (n ) 10

10 T
windows): 10 × log10  n = m M

BW ( n)

n=m

∑
∑



 ≤ 1 dB, where M is the maximum integer such that fr(M) <



fr(m) + 100 kHz (the inequality is strict, so if BW(m) = 100 kHz, then M = m). In other words, the PSD
power of each measured point in mw is divided by the PSD template in mw at that point; then summed
and averaged over a bandwidth that is as close as possible to 100 kHz, and must be less than or
equal to 1 dB in all 100 kHz sliding windows.
c) The total power of the transmitted PSD shall be no greater than the total power limit for that SM class,
as defined in clause 5.
The transverse balance of the associated TU-C and TU-R shall be greater than or equal to the
requirement for that SM class, as defined in clause 5.
The conformance testing methodology is designed for the purpose of lab evaluation of the compliance of
equipment to the SM classes defined in section 5. Its SM template defines a SM class, and the associated
SM mask is the SM template plus 3.5 dB.
The first step in the testing process is measurement of the transmitted PSD of the equipment under test.
The appropriate termination impedance and resolution bandwidth will a function of the SM class, and
therefore defined in sections 6.2 – 6.7. The result of the PSD measurement, in units of dBm/Hz and at a
center frequency of n × BW , is denoted by Pa (n ) , where BWr is the resolution bandwidth in kHz for the
SM class in question. The range of n is 1 ≤ n ≤ 30000 BWr .
Conditions for compliance:
Compliance is achieved by meeting the following conditions:
1 ≤ n ≤ 30000 BWr , Pa (n ) ≤ PM (n ) , where
frequency n × BWr , and PM (n ) = PT (n ) + 3.5dB .
a)For
P (n )
is the PSD of the SM mask at
 1 m + l −1 P (n ) 
a
 ≤ 1 dB for all 1 ≤ m ≤ 30000 BW − l + 1 and l = 100 BW . In other words,
b) 10 × log10  ×
r
r
l

(
)
P
n
T
n =m


the average over the measured PSD normalized by the template for the number of points equivalent
to 100 kHz must be less than the 1 dB.
∑
c)The total power of the transmitted PSD shall be no greater than the total power limit for that SM
class, as defined in sections 6.2-6.7.
d)The transverse balance of the associated TU-C and TU-R shall be greater than or equal to the
requirement for that SM class, as defined in section 5.
NOTE: numerical values 100 kHz and 3.5 dB are TBD.
6.2
PSD conformance criteria unique to spectrum management classes
6.2.1
Specific conformance criteria for spectrum management class 1
There are no specific PSD conformance criteria for Spectrum Management Class 1.
6.2.2
Specific conformance criteria for spectrum management class 2
There are no specific PSD conformance criteria for Spectrum Management Class 2.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
6.2.3
Specific conformance criteria for spectrum management class 3
There are no specific PSD conformance criteria for Spectrum Management Class 3.
6.2.4
Specific conformance criteria for spectrum management class 4
The general PSD conformance criteria in section 6.1 do not apply to spectrum management class 4. A
PSD mask is specified for the spectrum management class 4 instead of a PSD template. A member of
spectrum management class 4 shall have a measured PSD that shall not exceed the PSD mask that is
specified for spectrum management class 4 in Table 5 and Table 6 and Figure 6 and Figure 7 at any
frequency. A member of spectrum management class 4 shall also meet the total average power
limitations, transverse balance requirement, and deployment guidelines defined in 6.5 of this standard as
well as meeting all other applicable requirements in this standard.
6.2.5
Specific conformance criteria for spectrum management class 5
There are no specific PSD conformance criteria for spectrum management class 5.
6.2.6
Specific conformance criteria for spectrum management class 6
There are no specific PSD conformance criteria for spectrum management class 6.
6.2.7
Specific conformance criteria for spectrum management class 7
There are no specific PSD conformance criteria for spectrum management class 7.
6.2.8
Specific conformance criteria for spectrum management class 8
There are no specific PSD conformance criteria for spectrum management class 8.
6.2.9
Specific conformance criteria for spectrum management class 9
There are no specific PSD conformance criteria for spectrum management class 9.
6.3
PSD and total average power measurement procedure
The test methodology for measuring the PSD and the total average power of a device under test (DUT)
are defined in this subsection. For each spectrum management class, there are two different transmit
PSD test cases:
a) Downstream (CO to Remote) transmission: the measured output of a central office transmission
unit (TU-C).
b) Upstream (Remote to CO) transmission: the measured output of a remote transmission unit (TUR).
A DUT shall have total average power and PSD measured as described in this subsection in both the
upstream case and the downstream case in order to determine compliance with the total average power,
PSD conformance test, and other applicable conditions of a spectrum management class as defined in
this standard. Unless otherwise stated, all specifications apply to both the upstream case and the
downstream case. All measurements are performed directly at the transmitter output of the DUT with no
additional attenuation.
6.3.1
Test circuit for PSD and total average power measurement
A test setup as pictorially shown in Figure 10 shall be used for measuring total average power and PSD.
An example of a specific embodiment of this test setup is the circuit in Figure 11. VOUT is connected to a
high-impedance wideband rms voltmeter or spectrum analyzer. The PSD may be tested while line
powered or locally powered as required by the intended application of the DUT.
If the DUT is line powered then the test circuit shall contain provisions for DC power feed. If the DUT is not
line powered then the DC power-feed circuitry may be omitted from the test circuit. For line powered
applications, if the DUT is a TU-C the test shall be performed with the line power supply activated and an
appropriate DC current sink (with high AC impedance) attached to the test circuit. If the DUT is a TU-R the
test shall be performed with power (DC voltage) applied at the line interface (TIP/RING) by an external
voltage source feeding through an AC blocking impedance. Note that the DC current source/sink must
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
present high impedance (at signal frequencies) to common ground. The test circuit contains provisions for
transformer isolation for the measurement instrumentation. Transformer isolation of the instrumentation
input prevents measurement errors from unintentional circuit paths through the common ground of the
instrumentation and the DUT power feed circuitry. When the termination impedance of the test circuit
seen by the DUT output meets the calibration requirements defined in 6.3.2 the test circuit will not
introduce more than ± 0.25 dB error with respect to a perfect test load of exactly the specified resistance.
The DUT shall be measured by equipment that is not synchronous with the transmitted symbols of the
DUT, and there shall be no synchronization between the measurement equipment and the DUT. This is to
avoid making an inaccurate measurement because of the effects of cyclostationarity.
6.3.2
Calibration of the test circuit and termination impedance
The nominal termination impedance of the test circuit as seen by the DUT output shall be resistive with a
resistance of R Ohms as specified in Table 12 for the appropriate spectrum management class. The
minimum return loss with respect to the termination impedance R over the frequency band of 1 kHz to 5
MHz shall be 35 dB from 10 kHz to 2 MHz with a slope of 20 dB/decade below and above these corner
frequencies for measuring a DUT for conformance with Spectrum Management Classes 1, 2, 3, 4, and 5.
The minimum return loss with respect to the termination impedance R over the frequency band of 1 kHz to
30 MHz shall be 35 dB from 10 kHz to 20 MHz with a slope of 20 dB/decade below and above these
corner frequencies for measuring a DUT for conformance with spectrum management class 6.
Note: 35 dB return loss will allow ±0.20 dB measurement error with respect to the nominal termination
impedance value, R.
6.3.3
Operation of the DUT
The DUT shall be tested while it transmits the maximum power, and the maximum PSD levels at all
frequencies, at which it can transmit data when deployed. The DUT shall not have any power cutback
enabled. The DUT shall be tested under steady state conditions, after all start-up and initialization
procedures have been completed and while the DUT is transmitting data. To ensure that the DUT is in a
steady-state condition, while undergoing test the DUT shall not have measured total average powers in
any distinct 1.25 millisecond time intervals that differ by more than 8 dB. Although specific measurements
of average power and PSD during start-up and other non-data transmission phases are not provided, a
DUT that transmits inordinately high power or PSD levels during these phases may be considered to be in
non-compliance with this standard. The DUT input shall consist of a pseudo-random uniformly distributed
data sequence, and the DUT output shall be a fully modulated transmit signal with all overhead, framing,
coding, scrambling, modulation, filtering and all other operations performed on the data stream that the
modem would normally perform while transmitting data.
6.3.4
Total average power measurement procedure
The average power of a DUT shall meet the total average power requirements as specified in Section 5 of
this standard over the bandwidth specified in Section 5 of this standard for conformance with a spectrum
management class. The total average power may be tested while line powered or locally powered as
required by the intended application of the DUT. The total average power shall be measured and
averaged over a time span of at least 10 seconds.
6.3.5
Power spectral density (PSD) measurement procedure
6.3.5.1 PSD resolution bandwidth
PSDs are recorded by averaging the observed output power of the DUT on each of a number of
contiguous, regularly spaced, small frequency bands; with each frequency band having a defined
resolution bandwidth. The PSD of a DUT that is measured for conformance with Spectrum Management
Classes 1, 2, 3, or 4 shall be recorded with frequency spacing equal to the resolution bandwidths specified
in Table 13 at all frequencies from 1 kHz to 30 MHz.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
The PSD of a DUT that is measured for conformance with spectrum management class 5 shall be
recorded with frequency spacing equal to the resolution bandwidths specified in Table 14 at all
frequencies from 1 kHz to 30 MHz.
The PSD of a DUT that is measured for conformance with spectrum management class 6 shall be
recorded with frequency spacing equal to the resolution bandwidths specified in Table 15 at all
frequencies from 1 kHz to 30 MHz.
6.3.5.2 PSD Measurement time duration
Each frequency point (corresponding to a measurement in single resolution bandwidth) of a PSD shall be
measured by averaging the power in the resolution bandwidth of that frequency point for a time period of
at least 2.0 seconds. This requirement is equivalent to setting the sweep time for a single sweep of a
spectrum analyzer for duration equal to at least 2.0 seconds per frequency point.
Note: this requirement is based on a statistical derivation that showed that to measure the average power
in a given resolution bandwidth within 0.1 dB accuracy with 99% confidence required observation of about
9,000 transmitted symbols, and the slowest common signal is an ADSL tone which is at a 4 kHz rate.
Measuring an entire PSD for 2.0 seconds in all of each of the resolution bandwidths in Table 13, Table 14
and Table 15 requires a minimum observation time of 44 minutes.
6.4
Short-term stationary conformance criteria
6.4.1
Determination of whether to apply short-term stationary conformance criteria
The short-term stationary conformance criteria in clause 6.4.2 through 6.4.4 shall be applied to a DUT if
the total average power transmitted by the DUT in any two non-overlapping 1.25 millisecond time intervals
separated by less than 60 seconds can differ by more than 8 dB. This includes variation due to the
presence or absence of input data for transmission or the presence of specific input data sequences but
does not include variations due to external stimuli such as the application of externally controlled power
management, externally initiated retrain, or a change in crosstalk levels or loop conditions that causes
automatic retrain.
Equipment to which short-term stationary criteria are applied shall transmit at TBD dB below the SM
mask. In addition, the short-term stationary transmitter shall continuously transmit in the ON condition for
a minimum of 500 µsec.
6.4.2
Continuous mode for conformance testing
Equipment to which short-term stationary conformance criteria are applied shall provide a test
configuration in which the transmitter remains in the ON condition continuously. In the ON condition, the
DUT shall transmit the maximum power and the maximum PSD levels at all frequencies, at which it can
transmit data when deployed. The DUT shall not have any power cutback enabled. The DUT shall not
have measured total average powers in any distinct 1.25 millisecond time intervals that differ by more than
8 dB, including variation due to the presence or absence of input data for transmission or the presence of
specific input data sequences.
6.4.3
Frequency domain requirements
6.4.3.1 Continuous mode testing
Equipment to which short-term stationary conformance criteria are applied shall be tested in the
continuous ON condition specified in clause 6.4.2 using the conformance testing methodology defined in
clauses 6.1, 6.2, and 6.3. The PSD template used for testing conformance to a specific management
class shall be the template specified for that class, attenuated by a dB value provided in based on the
minimum percentage of time that the short-term stationary transmitter is on and transmitting full power in
any 4 second sliding window. The attenuation specified in applies only to the inband frequencies defined
for each management class in .
Automatic power adaptation between the 0 dB and 3 dB attenuation values in based on momentary
percentage of time that the short-term stationary transmitter is on and transmitting shall not be allowed.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
6.4.3.2 Short-term stationary mode testing
Equipment to which short-term stationary conformance criteria are applied shall be tested with input
conditions that generate the most frequent mode transitions permitted by the equipment. The
conformance testing methodology shall be as defined in clauses 6.1, 6.2, and 6.3 with the following
exceptions:
-
The requirement in clause 6.3.3 that the DUT shall not have measured total average powers in any
distinct 1.25 millisecond time intervals that differ by more than 8 dB shall be waived.
-
Each frequency point (corresponding to a measurement in single resolution bandwidth) of a PSD
shall be measured by averaging the power in the resolution bandwidth of that frequency point for a
time period of at least 4.0 seconds. This requirement is equivalent to setting the sweep time for a
single sweep of a spectrum analyzer for duration equal to at least 4.0 seconds per frequency point.
This requirement is used in place of the requirement in section 6.3.5.2.
-
The equipment vendor shall identify the input conditions necessary to generate the mode
transitions for this test.
6.4.4
Time domain requirements
Equipment to which short-term stationary conformance criteria are applied shall transmit in the ON
condition for a cumulative total of 10 40 milliseconds minimum in any 4 second period sliding window. In
addition, the short-term stationary transmitter shall continuously transmit in the ON condition for a
minimum of 500246 µsec. These requirements are. This requirement is intended to facilitate detection of
crosstalk from short-term stationary equipment by other receivers within a defined time interval.
6.5
Transverse balance testing methodology
Transverse balance is a comparison of the voltage of a transmitted metallic signal to the voltage of any
resulting longitudinal signal. It is the ratio of the metallic voltage VM at any frequency (f) to the transverse
voltage VL at frequency (f). The result in dB is expressed as:
Transverse Balance M −L = 20 Log10 [VM (f ) VL (f )]
where VM (f) = the metallic voltage applied across the tip and ring conductors of the port under test at any
frequency (f) between F1 and F2 is from a balanced source with a metallic impedance ZM, and VL (f) = the
resultant longitudinal voltage appearing across a longitudinal impedance ZL .
The greater the VM to VL ratio, the better the transverse balance of the transceiver unit and the less
likelihood that it will contribute to a crosstalk interference problem.
When calibrating the testing arrangement, the source metallic voltage should equal VM volts for each DSL
class when a metallic termination of ZM is substituted for the equipment under test. The metallic
impedance (ZM) shall be either 100 or 135 ohms as specified in clause 5.
The applicable ZL, ZM , F1, F2, and VM values for each DSL class are summarized in Table 16.
The minimum transverse balance requirements for the TU-C and TU-R equipment under test shall be
equaled or exceeded during all operating states and under all reasonable application of earth ground to
the equipment for the range of applicable frequencies (from F1 to F2) at all 2-wire loop ports with all
values of loop current that the port under test is capable of drawing when attached to the appropriate loop
simulator circuit.
The transverse balance testing methodology in TIA/EIA TSB31-B (or equivalent) shall be used to
determine conformance with the transverse balance requirements as specified in clause 5 for each
spectrum management DSL class. An illustrative test configuration for transverse balance conformance
testing is shown in Figure 12.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
The equipment under test, at the CO end, must meet the transverse balance requirements in Table 16.
The testing methods or equivalent are given in TIA/EIA TSB-31-B.
Table 2 provides a template to be used for the transverse balance requirements for various frequency
ranges;. tThe actual frequency range over which the requirements apply and to be included in testing is
dependent on the system under test.
Transverse balance testing shall only be performed over the range of frequencies included in the power
spectral density (PSD) applicable to the equipment under test and actually used in data transmission. For
that purpose, all of the signal pass-band shall be included, between the upper and lower –20 dB points.
Transverse balance may be measured while the DUT is line powered or locally powered. If the DUT is
line powered then the test circuit shall contain a dc voltage source. In such applications, if the DUT is a
TU-C the test shall be performed with TU-C line power activated and an appropriate dc current sink (with
high ac impedance) attached to the test circuit. If the DUT is a TU-R, the test shall be performed with the
appropriate dc voltage source applied between the tip and ring conductors through an ac blocking
impedance. The dc current source or sink must present high impedance (at signal frequencies) to
common ground. In line powered applications, the test circuit shall contain provisions for isolation of the
measurement instrumentation from unintentional circuit paths through the common ground of the
instrumentation and the DUT power feed circuitry.
6.6
Longitudinal output voltage testing methodology
Compliance with the limits as specified in clause 5 for each spectrum management DSL class is required
with a longitudinal termination having an impedance equal to or greater than a 100 ohm resistor in series
with a 0.15 uF capacitor. An illustrative test configuration for longitudinal output voltage limit conformance
testing is shown in Figure 20 of T1.601. For direct use of that test configuration, the near end transmitter
should be able to generate a signal in the absence of a signal from the far-end transceiver. The ground
reference for these measurements shall be the building or green-wire ground of the DUT.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table 1 - Spectrum management class 1 PSD template definition
Frequency Range, f (Hz)
0< f ≤ 25000 Hz
25000< f ≤ 76000 Hz
PSD Template (dBm/Hz)
-32.5

 f
− 32.5 − 10.35 × log10 

 25000 
76000< f ≤ 79000 Hz
 f − 76000 
− 37.5 − 0.5 × 

 3000 
79000 < f ≤ 85000 Hz
85000< f ≤ 100000 Hz
100000 < f ≤ 115000 Hz
115000 Hz < f ≤ 120000 Hz
120000 Hz < f ≤ 225000 Hz
225000 Hz < f ≤ 635000 Hz
635000 Hz < f
 f − 69000 
− 38 − 19.6 × log10 

 10000 
f − 85000
− 42 − 4 ×
15000
f − 100000
− 46 − 7 ×
15000
-53
f


-53 − 55 × log10 

 120,000 
f


- 68 − 70 × log10 

 225,000 
3




f 2
-143 - 10 log10 
13 
 1.134 × 10 


Table 2 - Minimum transverse balance template for the xTU-Crequirements
Frequency band
200 Hz -12 kHz
12 kHz - 1544 kHz
1544 kHz - 3000 kHz
Minimum transverse balance
40 dB
35 dB
30 dB
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table 3 - Spectrum management class 2 PSD template definition
Frequency, f (kHz)
PSD Template (dBm/Hz)
0
25
75
100
150
200
230
245
335
390
440
475
500
-36
-36
-36.5
-39
-45
-54
-64
-71
-72
-76
-83
-90
-98
3




f 2
-143 - 10 log10 
13 
 1.134 × 10 


500 < f
Table 4- Spectrum management class 3 PSD template definition
Frequency (khz)
0
50
125
210
310
370
550
670
750
980
1050
PSD template
(dBm/Hz)
-37
-37
-38
-41
-57
-73
-75
-85
-97
-98
-102.75
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table 5 - PSD mask definition for downstream transmission from a spectrum
management class 4 TU-C
Frequency
(kHz)
≤1
2
12
190
236
Maximum
Power
(dBm/Hz)
-54.2
-42.2
-39.2
-39.2
-46.2
Frequency
(kHz)
280
375
400
440
600
Maximum
Power
(dBm/Hz)
-35.7
-35.7
-40.2
-68.2
-76.2
Frequency
(kHz)
1000
2000
≥3000
Maximum
Power
(dBm/Hz)
-89.2
-99.7
-108
Table 6 - PSD mask definition for upstream transmission from a spectrum management
class 4 TU-C
Frequency
(kHz)
≤1
2
10
175
Maximum
Power
(dBm/Hz)
-54.2
-42.1
-37.8
-37.8
Frequency
(kHz)
220
255
276
300
Maximum
Power
(dBm/Hz)
-34.4
-34.4
-41.1
-77.6
Frequency
(kHz)
555
800
1400
≥2000
Maximum
Power
(dBm/Hz)
-102.6
-105.6
-108
-108
Table 7 - PSD template definition for downstream transmission from a spectrum
management class 5 TU-C
FREQUENCY BAND (Hz)
0<f<4
4000 < f < 25875
25875 <= f <= 81000
81000 < f <= 85000
85000 < f <=100000
100000 < f <=115000
115000 < f <= 120000
120000 < f < 138000
138000 <= f <= 1104000
1104000 < f <= 3093000
3093000 < f <= 4545000
4545000 < f <= 11040000
EQUATION FOR LINE (dBm/Hz)
-101, with max power in the in 0-4 kHz band of +15 dBrn
-96 + 21*log2(f/4000)
-40
-38-19.6*log10((f-6900)/10000)
-42-4*((f-85000)/15000)
-46-7*((f-100000)/15000)
-53
-72.5 +36*log2(f/80000)
-40
-40-36*log2(f/1104000)
–90 peak, with max power in the [f, f + 1 MHz] window of
(–36.5 –36 × log2(f/1104) + 60) dBm
-90 peak, with max power in the [f, f + 1 MHz] window of
-50 dBm
Table 8 - Spectrum management class 7 PSD template definition
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Frequency (Hz)
0
100,000
150,000
200,000
300,000
390,000
420,000
Power (dBm/Hz)
-40
-40
-40.5
-41.5
-42
-42
-43
Frequency (Hz)
775,000
1,000,000
1,100,000
1,300,000
1,500,000
1,900,000
2,000,000
Power (dBm/Hz)
-77
-77
-80
-86
-102
-104
-107
500,000
-51
> 2,000,000
3




f 2
-143 - 10 log10 
13 
.
1
134
10
×




Table 9 - Spectrum management class 8 PSD template definition.
Frequency
(kHz)
0
60
200
250
Power
(dBm/Hz)
-39
-39
-40
-40.5
Frequency
(kHz)
400
500
550
750
Power
(dBm/Hz)
-53
-66
-75
-76
315
-41
950
-84
Frequency
(kHz)
1120
1500
2000
> 2000
Power (dBm/Hz)
-95
-95
-107
3




f 2
-143 - 10 log10 
13 
.
1
134
10
×




Table 10 – PSD template definition for downstream transmission from a spectrum
management class 9 TU-C
FREQUENCY BAND (kHz)
0<f<4
4 < f < 25.875
EQUATION FOR LINE (dBm/Hz)
-101, with max power in the in 0-4 kHz band of +15 dBrn
-96 + 21 × log2 (f/4)
25.875 < f < 1104
1104 < f < 3093
-40
-40 – 36 × log2(f/1104)
3093 < f < 4545
–90 peak, with max power in the [f, f + 1 MHz] window of
(–36.5 –36 × log2(f/1104) + 60) dBm
4545 < f < 11040
-90 peak, with max power in the [f, f+1MHz] window of
-50 dBm
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table 11 - PSD template definition for upstream transmission from a spectrum
management class 9 or spectrum management class 5 TU-R
FREQUENCY BAND (kHz)
0<f<4
4 < f < 25.875
EQUATION FOR LINE (dBm/Hz)
-101, with max power in the in 0-4 kHz band of +15 dBrn
-96 + 21.5 × log2(f/4)
25.875 < f < 138
138 < f < 307
-38
-38 – 48 × log2(f/138)
307 < f < 1221
1221 < f < 1630
-903.5
–90 peak, with max power in the [f, f + 1 MHz] window of
(–90 – 48 × log2(f/1221) + 60) dBm
1630 < f < 11040
-90 peak, with max power in the [f, f+1MHz] window of
-50 dBm
Table 12 - Temination impedances
Spectrum management class
Class 1
Class 2
Class 3
Class 4
Class 5
Class 6
Class 7
Class 8
Class 9
Termination impedance R (Ohms)
135
135
135
135
100
100
TBD135
135
100
Table 13 - Resolution bandwidth for measuring a DUT PSD for conformance with
spectrum management classes 1, 2, 3, and 4.
Frequency Region
f <= 10 kHz
10 kHz <= f <= 3.1 MHz
3.1 MHz <= f <= 30 MHz
resolution bandwidth
1 kHz
3 kHz
100 kHz
Note: Values above 10 kHz are TBD
Table 14 – Resolution bandwidth for measuring a DUT PSD for conformance with
spectrum management class 5.
Frequency Region
f <= 10 kHz
10 kHz <= f <= 3.1 MHz
3.1 MHz <= f <= 30 MHz
resolution bandwidth
1 kHz
10 kHz
100 kHz
Note: values above 10 kHz are TBD
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table 15 – Resolution bandwidth for measuring a DUT PSD for conformance with
spectrum management class 6.
Frequency Region
f <= 10 kHz
10 kHz <= f <= 20 MHz
20 MHz <= f <= 30 MHz
resolution bandwidth
1 kHz
10 kHz
100 kHz
Note: values above 10 kHz are TBD
Table 16 – Summary of transverse balance testing criteria
SMC 1 SMC 2 SMC 3 SMC 4 SMC 5 SMC 6 SMC 7 SMC 8 SMC 9
90
TBD
90
90
90
ZL 500/90 500/90 500/90 500/90
(1)
(1)
(1)
(1)
135
135
135
100
TBD
135
135
100
ZM 135
VM 0.367 0.367 0.367 0.367 0.316 TBD 0.316 0.316 0.316
NOTES:
Numbers in this table are under study.
(1): The longitudinal impedance (ZL) shall be 500 ohms for frequencies from 200 Hz to 12 kHz and 90
ohms for frequencies above 12 kHz.
Table 17 - Maximum longitudinal output voltage limit
Applicable
Frequency
Range
Maximum Longitudinal
Output Voltage (rms) in all
4 kHz Frequency Bands
averaged over 1 second
Operating band
-50 dBV
From upper –30 dB
frequency to 4X the upper
-80 dBV
-30 dB frequency
– dB attenuation applied to PSD template for testing conformance of short-term stationary systems
Minimum percentage P of time
transmitting in any 4 second sliding
window
dDB attenuation
applied to PSD
template
100% ≥ P ≥ 10%
0.0 dB
10% > P ≥ 1%
3.0 dB
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
1% > P
Not allowed
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
– Frequencies over which the dB attenuation specified in Table 14 is applied for each spectrum
management class
Management class
Frequency range over which attenuation
in Table 14 is applied
SM class 1
200 Hz – 109 kHz
SM class 2
200 Hz – 238 kHz
SM class 3
200 Hz – 370 kHz
SM class 4
200 Hz – 450 kHz
SM class 5
4000 Hz – 3000 kHz
SM class 6
TBD
SM class 7
TBD
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
System A
Transmitter
First cable
section
+
Second cable
section
+
xTU-C
System A
Receiver
xTU-R
System B NEXT
noise generator
System B FEXT
and AWGN
noise generator
Figure 1 – Configuration for evaluation of effect of NEXT and FEXT into downstream
System A
Receiver
+
+
Test Loop
xTU-C
System A
Transmitter
xTU-R
System B FEXT
and AWGN
noise generator
System B NEXT
noise generator
Figure 2 – Configuration for evaluation of effect of NEXT and FEXT into upstream
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
-30
PSD Template (dBm/Hz)
-40
-50
-60
-70
-80
-90
-100
-110
0
100
200
300
400
500
600
700
800
Frequency (kHz)
Figure 3 - Spectrum management class 1 PSD template
-30
-40
PSD (dBm/Hz)
-50
-60
-70
-80
-90
-100
-110
0
100
200
300
400
500
600
700
800
Frequency (kHz)
Figure 4 - Spectrum management class 2 class PSD Template
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
-30
-40
PSD (dBm/Hz)
-50
-60
-70
-80
-90
-100
-110
0
200
400
600
800
1000
1200
Frequency (kHz)
Figure 5 - Spectrum management class 3 PSD template
-30
PSD (dBm/Hz)
-40
-50
-60
-70
-80
-90
-100
0
100
200
300
400
500
600
700
800
900 1000
Frequency (kHz)
Figure 6 - PSD mask for downstream transmission from a spectrum management class 4
TU-C
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
-30
P SD (dB m /Hz )
-40
-50
-60
-70
-80
-90
-100
0
100
20 0
300
40 0
5 00
600
Fr e q ue n c y (k H z)
Figure 7 - PSD mask for upstream transmission from a spectrum management class 4
TU-R
-40
PSD (dBm/Hz
-50
-60
-70
-80
-90
-100
-110
0
200
400
600
800 1000 1200 1400 1600 1800 2000 2200
Frequency (kHz)
Figure 8 - Spectrum management class 7 PSD template
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Figure 9 - Spectrum management class 8 PSD template
1 uF (min)
V
out
Resistive
Termination, R Ohms
(ground isolated
input)
DC
current
sink
Tip
Return loss
as per calibration
of the test circuit
Device
under test (DUT)
Ring
1 uF (min)
Figure 10 - PSD and total average power measurement setup
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
1:1
(+/- 1%)
Vout
20 mH (min)
(To highimpedance
load)
1 uF (min)
Tip
Return loss
as per calibration
of the the test circuit
DC
current
sink
R
Device
under test (DUT)
Ring
1 uF (min)
Figure 11 – Example PSD and total average power measurement setup
T1 (2)
1:1
S3
S2
S1
20 pF (5)
Tracking
Generator
EM
Equipment
Under Test
VM (6)
R2 (3)
R3 (7)
S2
R1 (1)
S3
EL
EL = longitudinal voltage
Spectrum
Analyzer (4)
EM = metallic voltage
1- Combined resistance of R1and tracking generator output resistance shall equal EUT impedance (I.e., 100 or 135 ohms).
2- Use center-tapped 1:1 transformer (e.g., Midcom 671-5767 or equivalent.
3- R2 provides the desired longitudinal impedance using 90 ohm or 500 ohm metal film or other non-inductive resistor.
4- High impedance spectrum analyzer or frequency selective voltmeter. It may be unbalanced.
5- Differential trimmer capacitor, 2.4 to 24.5 pF, Johnson 189-0759-005 or equivalent.
6- Any high impedance balanced or floating voltmeter with adequate frequency response. It need not be frequency selective.
7- R3 provides the desired calibration impedance. Should be a 100 or 135 ohm metal film or other non-inductive resistor.
Figure 12 - Illustrative test configuration for transverse balance conformance testing
42
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex A: Evaluation of interference from new technologies
into existing technologies
(normative)
A.1
Goals and framework for evaluation
The goal of spectral compatibility analysis described in this section is twofold:
a) to provide tests to validate that new services technologies will not interfere with existing
technologies, and
b) to allow sufficient flexibility to nurture innovation in new subscriber line transmission technologies
that further maximize the utility of the copper loop plant.
To achieve both goals simultaneously, this section describes computations that may be performed on new
signals to demonstrate spectral compatibility with existing technologies.
The rates and reaches of basis systems in this annex are provided only for analytical evaluation of
spectral compatibility with the basis systems.
Fitting within the spectrum management class PSD masks of the main body of this standard provides a
simplified test for spectral compatibility. However, this test alone would preclude large classes of new
transmission schemes which are spectrally compatible, and would stifle creativity for providing copper
access solutions. In order to nurture spectrally compatible innovation, this section describes a second,
more complicated evaluation (Test #2) that may be used to demonstrate compatibility technology by
technology with basis transmission technologies in the local loop. Test # 2 follows established industry
practices for demonstrating compatibility of new technologies during the definition of a standard. These
practices have been used successfully in the T1E1.4 working group for technical evaluation of services for
HDSL, HDSL2, ADSL, and VDSL, and would be sufficient for demonstrating compatibility of new
technologies. These analyses should be used to add to the spectrum management classes in this
standard at later dates. When followed rigorously, such analyses may be used as the basis for agreement
on spectral compatibility between parties sharing loop facilities, in the interim between updates of this
standard. Such agreements would be entirely between the parties and are outside the scope of this
standard.
The use of information in this Annex should be limited to the analysis of new technologies and proposed
Spectrum Management Classes. This Annex, including the definitions and performance criteria, does not
and is not intended to define “Significant Degradation”. Nor should any expectation of actual performance
be drawn or extrapolated from the information in this Annex.
The adoption of a new Spectrum Management Class shall require the determination of spectral
compatibility with all basis systems using the methods provided in this Annex. Furthermore, an analysis
should be performed to determine the spectral compatibility of a proposed new Spectrum Management
Class with existing transmission systems known to be representative of the existing Spectrum
Management Classes.
The number of Spectrum Management Classes should be the minimum necessary to serve the collective
needs of the industry. For this reason, during the development of future updates to this standard, the
removal certain Spectrum Management Classes should be considered when appropriate.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A.2
Analytical Method: Detailed crosstalk margin evaluations
Detailed margin calculations are required to demonstrate spectral compatibility of new technologies
outside of the established spectrum management categories. These calculations are described in this
section and must be calculated for each technology that may be interfered with. Because some
technologies are spectrally asymmetric, that is, use a different transmit spectrum in each direction,
evaluations must be performed in both the upstream and downstream directions.
The use of this section establishes non-interference with existing technologies by comparing the
performance of existing technologies in the presence of the new technology with industry-standard
reference performance levels in the presence of existing crosstalk. In this method, the established
reference disturbers are replaced in equal number by the new technology under trial, and the performance
margin of the technology being disturbed by the new technology is compared to the established reference
case. Appropriate reference evaluation loops, specified herein, are used for both the reference and new
technology disturber calculations. As noted in A1, the specified reference performance levels are not
intended to be performance targets for systems in the real world; they are only useful for comparing the
impacts on a basis technology due to crosstalk from a new technology and crosstalk from reference
technologies.
This analytical method evaluates the effect of crosstalk caused by metallic signals transmitted into the
loop plant by a new technology and assumes that no longitudinal voltages are transmitted or result from
inadequate transverse balance. This assumption is considered valid only if the new technology meets the
transverse balance specifications in Table 16 and the longitudinal voltage limits in Table 17 using the
testing methodologies for those parameters in Section 6.
This section is organized as follows. The subsections of A.2 describe the general methodology and the
specific margin calculations and methodology for a variety of technologies. Subsequent sections give the
transmission and performance parameters, and reference performance levels associated with each
existing technology. As new technologies become established, a subsection can be inserted into future
versions of this standard detailing the established performance benchmarks and method for calculating
compatibility with the new technology.
There are four three types of margin computations described in this section: DFE-based PAM signals
(e.g., 2B1Q ISDN and HDSL), DFE-based QAM/CAP signals and(e.g. CAP signals), DMT-based signals
(e.g., T1.413-1995 ADSL), and linear-equalization based signals (e.g., T1). Which computation is used
depends on the existing technology being affected, not the nature of the proposed technology.
A.2.1 General Methodology
The general model used for calculating both the reference performance levels and the performance of the
existing system in the presence of the new technology is shown in Figure A. 2. Note that the crosstalk
noise may be a mix of NEXT and FEXT from reference disturbers and/or new disturbers. When either
NEXT or FEXT is made up of different overlapping noise spectra, each should be constructed
independently of the other using the FSAN method. The NEXT may then be combined with the FEXT by a
simple power sum.
The simulation model for the downstream is shown in Figure A. 3. Note that the simulation model for the
upstream is exactly analogous to Figure A. 3, but with FEXT and NEXT transposed in the diagram.
The calculations to determine spectral compatibility with the basis systems proceed as follows:
1. For both upstream and downstream directions, calculate the reference performance levels for the
basis system per Figure A. 2 and Figure A. 3, using only reference disturbers. When calculating the
impact of the new technology on the basis system, the crosstalk noise from the new technology
replaces the appropriate number of reference disturbers.
2. When calculating the effect of new technologies on basis system performance, relative margin
calculations shall be performed. The same parameters used to calculate the reference performance
levels are used to calculate performance in the presence of crosstalk noise from the new technology.
Target loop length Z and target bit rate are input to the calculation and the resulting margin is output.
44
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
This new margin is compared with the margin used to obtain the target rates. If the new margin is no
more than some delta below the target margin for both upstream and downstream calculations (the
value of delta is TBD dB), then the new system is spectrally compatible with the basis system
3. If the upstream margin with the new system disturbers is below the target margin by some value
greater than delta, then the new system shall be considered not spectrally compatible with the basis
system under test, regardless of the outcome of the downstream calculation.
4. If the upstream margin passes the test but the downstream margin of the basis system is below the
target margin by some value greater than delta, then a new test shall be performed as shown in
Figure A. 4. Any reference disturbers are maintained on the Z kft. target loop, and the NEXT and
FEXT levels they present are unchanged. However, the new system disturbers are moved 100 feet
closer to the central office, so that they are on loops that are only Y = (Z – 0.1) kft. long. This reduces
the new NEXT level received at the TU-R since it is attenuated by the (Z-Y) kft. between the new
disturber and the TU-R, per Figure A. 5. The new FEXT level received by the TU-R is also reduced,
since the new system downstream signal couples only along the Y kft portion of the loop, but is
attenuated by the entire loop length Z.
The margin calculations shall be repeated for the downstream in an iterative fashion,
changingreducing the new loop length Y in increments of 100 feet but keeping Z constant, until the
largest value of Y is found that allows the basis system to maintain a margin which is no more than
delta below the target margin. The new system shall then be considered spectrally compatible with the
basis system under test only when the new system is deployed on loops Y kft. long or less. The
quantization step size for this process is 500 feet.
This process is depicted in Figure A. 6. Note that it is to be performed for each combination of test loop
and performance class indicated for each basis system in sections A3 and following. The new system
shall then be considered spectrally compatible only when the new system is deployed on loops Y’ kft. long
or less, where Y’ is the smallest value of Y, rounded to the nearest 500 foot increment, required for any
loop/performance scenario of all basis systems.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A.2.2 DFE-based PAM signals (e.g., 2B1Q ISDN and HDSL)
Margin for DFE-based PAM technologies is computed using an Optimal DFE calculation for PAM:
fbaud
Margin =
1 ⌠
fbaud 
⌡0
10 ∗ log10(1 + f _ SNR (f ))df − SNR_req dB
where f_SNR(f) is the folded received signal-to-noise ratio, defined as:
1
f _ SNR (f ) =
∑
n = −2
S(f + fbaud × n ) | H (f + fbaud × n ) |2
N (f + fbaud × n )
2
and S(f) is the desired signal’s (e.g., ISDN or HDSL) transmit power spectral density, |H(f)| is the
magnitude squared of the wireline loop transfer function, and N(f) is the total noise power spectral density
(crosstalk plus background noise) computed as described above. SNR folding, calculated out to 4 times
the Nyquist rate (twice the baud rate) is sufficient for all current xDSL signals. If future signals use more
bandwidth, they may require expansion of the range of n in the summation.
The C code in Table A.1 computes the optimal DFE SNR for PAM signals, from the given two arrays
containing received signal and received noise power spectral densities. By using the following code and
subtracting the required SNR from the result, one can compute PAM DFE margins as described above.
A.2.3 DFE-based QAM/CAP signals
Margin for DFE-base CAP/QAM technologies is computed using an Optimal DFE calculation for QAM:
Margin =
f +fbaud / 2
1 ⌠c
10 ∗ log10(1 + f _ SNR (f ))df − SNR _ req dB
fbaud 
⌡fc −fbaud / 2
where f_SNR(f) is the folded received signal-to-noise ratio, defined as:
3
f _ SNR (f ) =
∑
n =0
S(f + fbaud × n ) | H (f + fbaud × n ) |2
N (f + fbaud × n )
2
and S(f) is the desired signal’s transmit power spectral density, |H(f)| is the magnitude squared of the
wireline loop transfer function, and N(f) is the total noise power spectral density computed as described
above.
One important difference from the PAM calculation is that for QAM/CAP, S(f) = 0 for f < 0.
As in the PAM case, SNR folding is calculated out to 4 times the Nyquist rate, yet for QAM this is 4 times
the baud rate. As for PAM, future signals that use more bandwidth may require expansion of the range of
n in the summation.
Unlike PAM signals, it is important that the region of folding be sufficient to include any offset for the
carrier frequency of the QAM/CAP signal. This may be included either by changing the limits of integration
or by changing the limits on n in the SNR folding summation to adequately span the frequencies used by
the signal.
The C code in Table A. 2 computes the optimal DFE SNR for QAM/CAP signals, given two arrays
containing received signal and received noise power spectral densities. By using the code in Table A. 2
and subtracting the required SNR from the result, one can compute QAM/CAP DFE margins as described
above.
A.2.4 DMT margin computations
DMT systems allocate bits to individual carriers based on the Shannon capacity of the tones. Margin for
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
these systems is determined by the determining the Shannon capacity (minus appropriate SNR gap, and
plus coding gain), and then degrading the SNR at all frequencies until the capacity is equal to the desired
data rate. Capacity at an individual frequency is given by:
 S ( f ) | H ( f ) |2 

C(f ) = log 21 +


Γ
N
f
(
)


2
where S(f) is the received signal power spectral density at frequency f, |H(f)| is the magnitude squared of
the wireline loop transfer function, N(f) is the noise power spectral density at the receiver, as before, and Γ
is the effective SNR gap, as above. For coded systems, SNR gap is defined as (9.75 - (effective coding
gain)) dB. For the purposes of margin calculations, the effective SNR gap is increased by the desired
margin, and is defined as Γ = 9.75 - (effective coding gain) + Margin (dB).
Total capacity for the DMT system is then computed by integrating C(f) across the frequency band used
by the DMT system. Some DMT systems have a minimum number of bits per tone (such as T1.413-1995,
T1.413-1998, and ITU-T G.992.2, all of which support a minimum of 2 bits/tone (MINBITS=2)). In
calculations for these systems, C(f) must be further limited not to exceed the prescribed maximum.
When computing DMT capacity, the resulting integration is conditional at each frequency:
C=
∫DMT bandwidth C' (f )df ,
where C’(f) = min(C(f), MAXBITS) if C(F)>MINBITS , and C’(f) = 0 if C(f)<MINBITS, and DMT bandwidth
is the frequency range used by the data carrying tones of the desired DMT signal. It is worth noting that
implemented DMT systems go through a process of bit loading and adjustment of powers to each of the
tones. However, studies have shown that margins achieved by such algorithms closely match those
achieved by the less implementation dependent capacity calculation shown here.
The Matlab-code in Table A.3 and Table A. 4 computes DMT margins.
A.2.5 Margin computations for linear equalization systems (e.g., T1)
To be added later.
A.3
Compatibility with voicegrade services and technologies
A.3.1 Description of voicegrade services and technologies
Voicegrade services and technologies use the frequency spectrum from 0 to 4 kHz and often employ
various types of dc and ac signaling. There are several types of voicegrade signals and the impact of
crosstalk interference varies depending upon the type of disturbed signal. For example, voice systems
are concerned about the subjective effects of background noise during silent intervals when no speech is
present, whereas analog voiceband data systems are concerned about the signal-to-noise ratio during
data transmission.
Voicegrade services and technologies transmit signals that can be placed into one of five general
categories:
–
speech signals
–
single and dual tone signals
–
low frequency signals
–
digital data
–
analog data.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A.3.1.1 Speech signals
Speech signals include live voice as well as recorded announcements. Most of the speech energy is in
the frequency range from 200 to 3400 Hz. The most sensitive speech receiver is the human ear. It has
been found that background noise during silent intervals when no speech is present is the most disturbing
noise to the average listener. Background noise is measured with a C-message weighting filter that
simulates the effects of the average human ear with a 500-type telephone set. A background noise level
of 20 dBrnC or less is considered to be acceptable.
A.3.1.2 Single and dual tone signals
Single and dual frequency tones are used as network control and addressing signals, call progress
signals, and alerting signals. Network control and addressing signals include dual-tone Multi-frequency
(DTMF) signaling, multi-frequency (MF) signaling, single frequency (SF) signaling, and coin deposit
signals. Call progress signals include dial tone, busy tone, reorder tone, audible ring, special information
tones, and receiver off-hook tone. Call waiting tone is an example of a single frequency alerting tone that
is used with a supplemental feature on analog access lines.
Single and dual-tone signals range in frequency from 440 Hz to 2600 Hz and require signal-to-noise ratios
on the order of 16 to 28 dB for reliable detection.
A.3.1.3 Low frequency (< 100 Hz) signals
Ringing, maintenance signals, and subvoice data systems are examples of signals that use low (< 100
Hz) frequencies. The actual frequency range of the various signals is from about 17 to 83 Hz. These
signals have a relatively high tolerance for noise compared to other voicegrade signals.
A.3.1.4 Digital data
Digital data subrates use voiceband frequencies. The lowest digital data rates are entirely within the
voiceband. Digital data at 2.4 kb/s has nulls at 0 and 2.4 kHz with maximum power at 1.2 kHz. Digital
data at 3.2 kb/s has nulls at 0 and 3.2 kHz with maximum power at 1.6 kHz. Digital data rates at 4.8 kb/s
and above use bandwidths that are wider than the 4 kHz voiceband. For example, the 4.8 kb/s digital data
signal has nulls at 0 and 4.8 kHz with energy concentrated at 2.4 kHz.
The maximum loop loss for digital data services is 31 dB between 135-ohm terminations at the frequency
that represents one-half of the data rate. The minimum signal-to-noise ratio that provides acceptable
performance is 20 dB.
A.3.1.5 Analog data
Several types of analog data are used in the loop environment. The most common types are:
–
Low-speed frequency shift keying (FSK) associated with supplemental network features such as
Calling Number Delivery, Calling Name Delivery, and Visual Message Waiting Indicator.
–
Customer data using one of the ITU-T standards such as V.34 or V.90.
The network-originated FSK data messages associated with network supplemental features on analog
access lines generally require a signal-to-noise ratio of at least 25 dB. V.34 modems require a signal to
noise ratio of 39 dB. V.90 modems are the most sensitive voiceband data modems requiring a 50 dB
signal-to-noise ratio to operate at the maximum speed. The high signal-to-noise ratio makes the V.90
modem the most sensitive of all of the voicegrade technologies.
A.3.2 Voicegrade evaluation
Because of the subjective effects of speech crosstalk, particularly intelligible crosstalk, special
consideration must be given to crosstalk between loops that carry speech signals. In addition, voiceband
signals that have narrow spectral characteristics also require complicated evaluations to determine the
subjective effects of single frequency crosstalk interference on a human listener. This standard does not
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
provide guidance for evaluating the subjective effects of speech crosstalk or single frequency interference.
This standard assumes that the transmission system under evaluation is a DSL system that has spectral
energy that is dispersed across a portion of the voiceband and that the crosstalk noise from such a
system will have a Gaussian noise distribution.
The voicegrade spectral compatibility evaluation assumes that the V.90 modem is the victim technology. If
the DSL system under evaluation passes this evaluation, then it is unlikely that crosstalk interference
problems will result with the other, more robust, types of voicegrade systems.
It is convenient to evaluate V.90 performance in terms of the total crosstalk noise power that occurs in the
frequency band from 0 to 4 kHz.
A.3.2.1 Evaluation loop
The loop used for voicegrade evaluations shall be 15 kft of 26-gauge cable. This loop has a resistance of
1250 ohms and a 1 kHz loss of 7 dB when terminated at each end with 900 ohms.
A.3.2.2 Reference crosstalk environment
Spectral compatibility evaluations that use the V.90 modem as the victim technology shall assume fortynine disturbers in a 50-pair binder group.
A piece-wise linear crosstalk model is used for evaluations (see Figure A. 1 and Table A. 5). A simplified
49-disturber model that has 67 dB of loss at 20 kHz and a linear (log-log) slope of –4 dB per decade can
be expressed as:
NEXT49 = 10 log10[(f)
2/5
8
÷ 2.11 x 10 ]
where (f ) is in Hz from 200 to 20,000.
A.3.2.3 Crosstalk noise and peak power levels computation
Evaluations shall be performed in both the upstream and downstream directions. The DSL system under
evaluation shall be considered spectrally compatible with the V.90 modem, and voicegrade services and
technologies in general, if the NEXT caused by 49-disturbers in the same binder group meets the
voiceband NEXT PSD and total voiceband NEXT noise objective. The DSL system under evaluation shall
be considered spectrally compatible with voicegrade services and technologies in general, if the NEXT
caused by 49-disturbers in the same binder group meets the voiceband NEXT PSD requirement and total
voiceband NEXT noise requirement.
A.3.2.3.1
Voiceband NEXT PSD
The NEXT PSD at any frequency from 200 to 4,000 Hz caused by 49-disturbers on a victim pair in the
same binder group shall not exceed –97.5 dBm. To determine compliance, the 200 to 4,000 Hz PSD of
the system under evaluation is passed through the 49-disturber crosstalk model. The resultant NEXT
power level for each frequency is compared to the requirement.
PSDD(f) + 10 log10[(f )
2/5
8
÷ 2.11 x 10 ] ≤ – 97.5 dBm/Hz
The voiceband NEXT PSD requirement is met by any DSL system that has a transmit PSD is less than
-29 dBm/Hz across the frequency band from 200 to 4000 Hz.
If the voiceband NEXT PSD requirement is not met, the system under evaluation has failed to
demonstrate spectral compatibility with the V.90 modem, and voicegrade systems in general.
A.3.2.3.2
Total voiceband NEXT noise limit
The total NEXT noise on a victim pair caused by 49-disturbers in the same binder group should not
exceed –75 dBm (15 dBrn). To determine compliance, the 200 to 4,000 Hz PSD of the system under
evaluation is passed through the 49-disturber crosstalk model at each frequency and the NEXT noise at
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
each frequency is then summed on a power basis. The resulting total voiceband NEXT noise is then
compared to the requirement.
4000
10 log10
∫ PSDD (f ) × [(f )
2/5
200
]
÷ 2.11× 108 df ≤ −75dBm ;
where the PSD is expressed in linear units (e.g., mw/Hz).
This objective is met by any DSL system that has a transmit PSD that is less than –41 dBm/Hz across the
frequency band from 200 to 4000 Hz.
If the NEXT power level objective of ≤ -75 dBm is not met, the system under evaluation has failed to
demonstrate spectral compatibility with the V.90 modem. In order to demonstrate spectral compatibility
with voicegrade systems in general, the total NEXT noise in the frequency band from 1 to 4000 Hz on a
victim pair caused by 49-disturbers in the same binder group shall not exceed –66 dBm (24 dBrn). This
requirement is met by any DSL system that has a transmit PSD less than –32 dBm/Hz across the
frequency band from 200 to 4000 Hz.
A.3.3 Spectral compatibility of voicegrade systems with basis systems
The FCC has adopted rules and regulations in Part 68 for CPE in order to protect the network from harm.
One of the harms recognized by the FCC is crosstalk interference. The FCC has adopted signal power
limitations and longitudinal balance limitations to prevent crosstalk interference from being caused by
voicegrade CPE.
CPE that meets the voice or voiceband data signal power limitations in Part 68 will have spectral
compatibility with all of the basis loop transmission systems listed in 4.3.
Likewise, network equipment that meets the encoded analog content specifications in Part 68 will have
spectral compatibility with all of the basis loop transmission systems listed in 4.3.
A.4
Compatibility with Enhanced Business Services
A.4.1 Description of Enhanced Business Services
Enhanced Business Services use the frequency spectrum from 0 to 10 kHz and are used to transport
speech signals in the same way as done by traditional voicegrade services. A signalling channel is also
present that allows to perform all functions associated with the setting up and tearing down of voice calls
without the use of high voltage signalling.
A.4.1.1 Speech signals
The speech signals are carried in the 0 to 4 kHz band in the same way as done by voicegrade services
described in section A.3. Compatibility with speech signals must be assessed in the same manner as
described in section A.3.
A.4.1.2 Signalling functions
Signalling functions required to set up or tear down a call, and also to transmit information required to
implement service features such as Caller Id Display are transported over a digital signalling channel.
Data transmission over that channel is performed by modulating an 8-kHz carrier.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A.4.2 Enhanced Business Service Evaluation
Because the impact of the DSL system under evaluation on the speech signal of the victim Enhanced
Business Service line has been evaluated following the criteria for voicegrade services evaluation, only the
impact on the signalling channel needs to be assessed.
A.4.2.1 Reference crosstalk environment
Spectral compatibility evaluations that use the Enhanced Business Services as the victim technology shall
assume forty-nine disturbers of the DSL system under evaluation in a 50-pair binder group.
A piece-wise linear crosstalk model is used for evaluations (see Figure A.1). A simplified 49-disturber
model that has 66 dB of loss at 20 kHz and a linear (log-log) slope of –4 dB per decade can be expressed
as:
NEXT49 = 10 log10[(f)
2/5
8
÷ 2.11 x 10 ]
where (f ) is in Hz from 200 to 20,000.
A.4.2.2 Crosstalk noise and peak power levels computation
Evaluations shall be performed in both the upstream and downstream directions if the DSL system under
evaluation uses different PSD masks for each direction. Otherwise, only one direction suffices. The DSL
system under evaluation shall be considered spectrally compatible with the Enhanced Business Services,
if the NEXT caused by 49-disturbers in the same binder group meets the signalling band NEXT PSD and
if the voicegrade requirements of section A.3 are met.
A.4.2.2.1
Signalling Band NEXT PSD
The NEXT PSD at any frequency from 6,000 to 10,000 Hz caused by 49-disturbers on a victim pair in the
same binder group shall not exceed –96.0 dBm/Hz. To determine compliance, the 6,000 to 10,000 Hz
PSD of the system under evaluation is passed through the 49-disturber crosstalk model. The resultant
NEXT power level for each frequency is compared to the requirement.
PSDD(f) + 10 log10[(f )
2/5
8
÷ 2.11 x 10 ] ≤ – 96.0 dBm per Hz
The signalling band NEXT PSD requirement is met by any DSL system that has a transmit PSD less than
-29 dBm/Hz across the frequency band from 6,000 to 10,000 Hz.
If the signalling band NEXT PSD requirement is not met, the system under evaluation has failed to
demonstrate spectral compatibility with Enhanced Business Services.
A.4.2.3 Spectral compatibility of Enhanced Business Services with basis systems
The FCC has adopted rules and regulations in Part 68 for CPE in order to protect the network from harm.
One of the harms recognized by the FCC is crosstalk interference. The FCC has adopted signal power
limitations and longitudinal balance limitations to prevent crosstalk interference from being caused by
voicegrade CPE.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
CPE that meets the voice or voiceband data signal power limitations in Part 68 will have spectral
compatibility with all of the basis loop transmission systems listed in 4.3.
Likewise, network equipment that meets the encoded analog content specifications in Part 68 will have
spectral compatibility with all of the basis loop transmission systems listed in 4.3.
A.5
Compatibility with T1.410
ANSI T1.410-1992 (alternatively known as the Digital Data System or DDS) operates at rates from 2.4
kb/s to 64 kb/s, symmetrically, using simplex transmission over two non-loaded wire pairs. It is the primary
means for low rate connections for Frame Relay service, and is still quite popular, with over 200,000 new
installations each year. While 56 or 64 kb/s service is primarily used for Frame Relay, there still is a
significant deployment of subrate (2.4, 4.8 or 9.6 kb/s) service for automated teller machines and lottery
networks.
T1.410 uses 50% duty-cycle AMI transmission, similar to that of T1. The main lobe of the transmitted
spectrum lies in the frequencies between 0 and the signaling rate, with the peak at ½ the bit rate. As
st
specified in the standard, the transmit filter is 1 order, with a 3 dB point at 1.3 times the signaling rate (at
rates below 19.2 kb/s, some additional filtering is present.) Maximum transmit power is 6 dBm into 135
Ohms, except at the 9.6 kb/s rate, where the transmitted power is limited to 0 dBm (both number
computed for equal-probable 0s and 1s, since T1.410 does not employ data-randomizing scramblers). For
single channel service up to 56 kb/s, the signaling rate is the same as the service rate. For a service rate
of 64 kb/s, the signaling rate is 72 kb/s. Optionally, at rates of 56 kb/s and below, a secondary channel is
present, which increases the signaling rate by approximately 30%.
T1.410 specifies that transceivers operate on loops where the insertion loss at the 1/2 the signaling
frequency is 34 dB. Additional loop deployment practices limit the length of bridged taps that can be
present on the line. At rates below 19.2 kb/s, single and total bridged tap lengths are limited to 6 kft. At
rates of 19.2 kb/s and above, the total bridged tap length is limited to 2.5 kft with no single bridged tap
exceeding 2.0 kft.
A.5.1 Computation of DDS Performance – Margin Computation for AMI Transceivers
DDS uses AMI transmission with a 50% duty cycle. Historically, the receivers have used a rather simple
rd
structure, which incorporates a linear equalizer with only a single zero, and a 3 order lowpass filter (See
[1]).
The optimal (from a minimum mean squared error perspective) linear receiver for a 50% duty cycle pulse
can be obtained through the procedure described in [2]. For DDS, the resulting equalized channel
resembles a 60% raised cosine channel, which rolls off much faster than the third order lowpass filter
suggested in [1].
Since bipolar violations are used as control codes, the DDS receiver is not able to fully exploit the
correlation in the AMI signal for maximum performance. To derive the optimum receiver margin, we
assume a 2 level signal, and then increase the required SNR to compensate for the power difference
between the AMI and 2 level signals. (In fact, the result is nearly the same as we get if the correlation is
taken into account.) Starting from the work in [1], we can obtain
MSE =
fbaud / 2
1
M (f )
df ,
fbaud −fbaud / 2 M (f )L(f ) + 1
∫
where L(f) is defined as:
L(f ) =
∞
∑
n = −∞
G(f −fbaud ×n )H (f − fbaud ×n
2
N (fifbaud × n )
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and M(f) is spectrum of the message sequence (sin 2 shaped for AMI.) G(f) represents the transmitted
pulse shape; for AMI this includes the 50% duty cycle and any other filtering (1st order for DDS). H(f) is
the channel response, and N(f) is the noise spectrum. When M(f) is a constant (M0), this can be reduced
to the familiar margin equation for linear equalization:
fbaud / 2
 1
1
M arg indB = −10∗ log10
df
/
2
−
fbaud
f _ SNT ( f ) + 1
 fbaud
∫

 − SNR _ reqdB

and f_SNR, the folded SNR is given by
f _ SNR =
∞
∑
M0 G(f − fbaud × n )H (f − fbaud × n )
N (f − fbaud × n )
n = −∞
=
∞
∑
n = −∞
2
S(f − fbaud × n ) H (f − fbaud × n )
N (f − fbaud × n )
2
.
To account for the transmit power increase caused by the AMI correlation, we increase the required SNR
by 3 dB (the power difference for a ternary signal compared to a binary signal with the same level
-7
separation.) Then for a 10 error rate, the SNR_Required for a pseudo-optimum AMI receiver is
approximately 17.3 dB.
Since actual receivers have additional impairments (mis-equalization, timing jitter, etc.), the actual
required SNR is often higher than the 17.3 dB listed here. In addition, since the actual receive filters don’t
roll off as fast at the optimal receiver, additional noise power may reach the decision device, reducing the
actual SNR from that theoretically calculated. These conditions noted, we present the optimal calculation
as the basis for the relative performance measures to be used in this section.
A.5.2 Evaluation loops
The maximum metallic loop loss for T1.410 is 34 dB at ½ the signaling frequency (Nyquist frequency.)
Loop loss shall be calculated assuming 135-ohm terminations. Because DDS transceivers use linear
equalization, both upstream and downstream scenarios use the worst case loops listed below:
For the 56 kHz signaling rate, the Nyquist frequency is 28 kHz. ANSI T1.601 test loop 6 is representative
of a worst case loop, and is used for 56 kb/s evaluation.
For the 72 kHz signaling rate, the Nyquist frequency is 36 kHz. ANSI T1.601 test loop 10 is representative
of a worst case loop, and is used for 64 kb/s evaluation.
For the 9.6 kHz signaling rate, the Nyquist frequency is 4.8 kHz. 27 kft of 26 AWG is representative of a
34 dB loop, and is used for 9.6 kb/s evaluation.
A.5.3 Reference crosstalk environment
T1.410 is deployed today in the same loop plant with T1.601 ISDN. ISDN is the worst existing disturber for
DDS. To assess the effect of crosstalk from new technologies on DDS, a relative comparison will be made
with ISDN crosstalk. If a new technology produces the same or higher margins than that obtained with
ISDN crosstalk, then it is deemed compatible with DDS. The SM class 1 template will represent ISDN
crosstalk.
Since DDS only transmits on 1 of 2 pairs in use, spectral compatibility studies that use DDS as the
disturber technology should assume 24 disturbing DDS systems in a 50-pair binder group.
The reference crosstalk environment against which new technologies will be compared is 49 Spectrum
Management Class 1 disturbers. The two-piece Unger model for NEXT described in Figure A. 1 and Table
A. 5 is to be used for crosstalk into DDS due to the low frequency nature of the signal.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A.5.4 Margin computation
DDS relative margin is computed as described above (A.4.1) for AMI signals. For the new technology to
be considered spectrally compatible with DDS, the following scenarios must produce margins no lower
than that computed using the same loop and noise-coupling models with 49 spectrum management class
1 disturbers:
a)
49 new technology NEXT/FEXT
b)
24 SM class 1 NEXT/FEXT + 24 new technology NEXT/FEXT.
DDS evaluations at 9.6 kb/s and 64 kb/s should be sufficient to ensure spectral compatibility with all DDS
rates.
Required SNR (SNR_req) for DDS is 17.3 dB. The transmit signal spectrum used in the calculation is that
of a 50% duty cycle bipolar signal, balanced about DC (50% positive pulses, 50% negative pulses) and
passed through a 1 pole filter with 3 dB point at 1.3 times the signaling rate. Transmitted power is 6 dBm
for 56/64 kb/s, and 0 dBm for 9.6 kb/s. A frequency resolution of approximately 100 Hz (FDELTA=100 Hz)
should be used for 56/64 kb/s DDS margin calculations and 20 Hz for 9.6 kb/s DDS margin calculations
due to the narrow bandwidth of the signal.
A.6
Compatibility with ISDN DSL
Using the transmit spectrum for ISDN described in Annex B of T1.413-1995, spectral compatibility with
ISDN is verified by performing an Optimal DFE margin calculation for DFE-based PAM signals to
determine ISDN margin in the presence of the proposed signal. The remainder of this section defines the
test parameters.
A.6.1 Evaluation loops
Upstream Direction: Since ISDN DSL uses spectrally symmetric echo-canceled transmission, in the
upstream a worst-case near-end crosstalk event would occur when the ISDN loop is longest and the new
technology is crosstalking into the ISDN signal. ANSI T1.601 Loop 1 (18 kft comprised of 16.5 kft 26AWG
and 1.5 kft 24AWG) should be used for this test.
Downstream Direction: Evaluation should be performed on the shorter of either (a) the longest single
length of 26 AWG copper that the proposed technology will run on, or (b) ANSI T1.601 Loop 1 (as for the
upstream).
A.6.2 Reference Crosstalk environment
The reference crosstalk environment against which new technologies will be compared is:
49 Spectrum Management Class 1 template (self-NEXT) disturbers.
The two-piece Unger model for NEXT described in Figure A. 1 and Table A. 5 is to be used for crosstalk
into ISDN due to the low frequency nature of the ISDN signal.
A.6.3 Margin Computation
ISDN DSL margin is computed as described for DFE-based PAM signals. The computed margin for ISDN
against the proposed technology as a disturber should be compared against a calculation using the same
loop and noise coupling models for:
a) 49 new technology NEXT/FEXT, or
b) 24 SM class 1 Template NEXT/FEXT + 24 new technology NEXT/FEXT.
Spectral compatibility requires that computations for the maximum allowable numbers of the proposed
technology (based on self-crosstalk limitations) disturbing ISDN produce no lower ISDN margins than 49
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ISDN DSL NEXT. Required SNR (SNR_req) for ISDN is 21.4 dB. The baud rate for the 2B1Q ISDN
signal is 80 kHz. Model resolution of approximately 100 Hz (FDELTA=100 Hz) should be used for ISDN
margin calculations due to the narrow bandwidth of the signal.
A.7
Compatibility with HDSL
Using the transmit spectrum for HDSL described in Annex B of T1.413-1995, spectral compatibility with
HDSL is verified by performing an Optimal DFE margin calculation for DFE-based PAM signals to
determine HDSL margin in the presence of the proposed signal. The remainder of this section defines the
test parameters.
A.7.1 Evaluation loops
Upstream Direction: In practice, CSA4 has been shown to be a greater impediment to HDSL transmission
than the longest loops (CSA6 and CSA8). CSA4 should be used for margin evaluation.
Downstream Direction: Evaluation should be performed on the shorter of either (a) the longest single
length of 26 AWG copper that the proposed technology will run on, or (b) CSA4.
A.7.2 Reference crosstalk environment
The reference crosstalk environment against which new technologies will be compared is:
49 SM class 3 template disturbers.
Either the simplified T1E1 NEXT model or the two-piece Unger model may be used for crosstalk
evaluation. See Figure A. 1 and Table A. 5.
A.7.3 Margin computation
HDSL margin is computed as described for DFE-based PAM signals. The computed margin for HDSL
against the proposed technology as a disturber should be compared against a calculation using the same
loop and noise-coupling models for
a) 49 new technology NEXT/FEXT, or
b) 24 SM class 3 template NEXT/FEXT + 24 new technology NEXT/FEXT.
Spectral compatibility requires that computations for the maximum allowable numbers of the proposed
technology (based on self-crosstalk limitations) disturbing HDSL produce no lower HDSL margins than 49
SM class 3 template NEXT. The baud rate for the HDSL signal is 392 kHz. Required SNR (SNR_req) for
HDSL is 21.4 dB. Model resolution of at least 500 Hz (FDELTA <= 500) should be used for the HDSL
margin calculations.
A.8
Compatibility with ADSL and RADSL technologies
ADSL compatibility is inherently more complicated than for fixed-rate technologies. Compatibility with
ADSL must consider performance levels at different loop reaches, as appropriate to the deployment reach
of the technology being evaluated as a disturber to ADSL. This section addresses T1.413-1998,
CAP/QAM RADSL and ITU Recommendations G.992.1 and G.992.2.
A.8.1 Evaluation loops and performance levels
4 performance classes of ADSL are determined:
a) 5440 kb/s downstream, 640 kb/s upstream at reaches up to 9 kft 26 AWG (Loop CSA 6).
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
b) 1720 kb/s downstream, 176 kb/s upstream at reaches up to 13.5 kft 26 AWG (ANSI T1.601
Loop 7)
c) 1720 kb/s downstream, TBD kb/s upstream at reaches up to 12 kft 26 AWG.
d) 256 kb/s downstream, 96 kb/s upstream on T1.601 Loop 1 and on T1.601 Loop 2.
Downstream evaluation loops: For the downstream direction, evaluation will be on the shorter of either the
longest reach of the proposed system, or the reach of the desired ADSL performance level. In the case
where the evaluation is limited by the reach of the proposed system, the performance level required of
ADSL will be the next longer reach level (e.g., if the proposed system reaches 10 kft on 26 AWG, then the
performance level should be 1720 kb/s downstream, 400 kb/s upstream). In these cases, performance at
the limited reach is compared with margins given by the reference crosstalk environment at the targeted
reach of the desired performance level (in the example, at 12 kft 26 AWG).
Upstream evaluation loops: For the upstream direction, evaluation needs to consider all four performance
levels, regardless of the reach of the technology being evaluated. In practice, however, meeting level 1,
the moremost stringent of the three should be sufficient.
A.8.2 Reference crosstalk environments
Downstream:
Performance Class 1: (5440/640 kb/s, CSA reach): 20 SM class 3 template NEXT/FEXT disturbers.
Performance Classes 2&3: (1720/176 kb/s, 13.5 kft reach): 24 SM class 3 template NEXT/FEXT
disturbers. Performance class 4: (256/96 kb/s, T1.601 loops 1 & 2): 10 Spectrum Management Class 1
NEXT/FEXT disturbers.
Upstream:
Performance classes 1, 2 &3: 20 SM class 3 template NEXT/FEXT disturbers: 10
Performance class 4: Spectrum Management Class 1 template NEXT/FEXT disturbers
Either the simplified T1E1 NEXT model or the two-piece Unger model may be used for crosstalk
evaluation. See Figure A. 1 and Table A. 5.
A.8.3 Margin computation
T1.413-1998, G.992.1, and G.992.2 ADSL Margins are computed as described for DMT signals.
CAP/QAM RADSL margins are computed as described for CAP/QAM DFE signal.
For the purposes of these evaluations, the ADSL or RADSL transmitted PSD (and baud rates for RADSL)
defined in the relevant standards or recommendations, should be used for the ADSL or RADSL signals.
Evaluations will be performed for each type of ADSL (T1.413-1998, (non-overlapped
upstream/downstream spectra with the reduced NEXT transmit spectra of annex F), G.992.1 (also with
non-overlapped upstream/downstream spectra), G.992.2, and CAP/QAM, according to the parameters
within the relevant standards).
The data rates and noise models from A.7.2 are used in the formulas in the formulas from A.2. No
additional overhead is added to these rates; the results are relative, not absolute.
In order to reduce the sensitivity of this procedure to model accuracy, the computed margin for ADSL or
RADSL with the proposed new technology as a crosstalker should be compared against a calculation
using the same models for the reference crosstalkers for each performance/reach class, rather than
against a particular fixed minimum specified performance margin; e.g., 6 dB.
Downstream
Performance class 1:
a) 20 new technology NEXT/FEXT, or
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b) 10 SM class 3 template NEXT/FEXT + 10 new technology NEXT/FEXT.
Performance classes 2 & 3:
a) 24 new technology NEXT/FEXT, or
b) 12 Spectrum Management Class 1 template NEXT/FEXT + 12 new technology NEXT/FEXT.
Performance class 4:
a) 10 new technology NEXT/FEXT, or
b) 5 Spectrum Management Class 1 template NEXT/FEXT + 5 new technology NEXT/FEXT.
Upstream
Performance classes 1, 2, &3:
a) 20 new technology NEXT/FEXT, or
b) 10 SM class 3 template NEXT/FEXT + 10 new technology NEXT/FEXT.
Performance class 4:
a) 10 new technology NEXT/FEXT, or
b) 5 Spectrum Management Class 1 template NEXT/FEXT + 5 new technology NEXT/FEXT.
Spectral compatibility requires that computations for the same number of disturbers as in the reference
case (up to the maximum allowable number of the proposed technology based on self-crosstalk
limitations) disturbing ADSL produce no lower ADSL margins than the reference cases. Model resolution
of at least 4 times the tone spacing of the DMT signal should be used for ADSL margin calculations.
A.8.4 Compatibility with RADSL
Single carrier RADSL per TR-59 uses the same FDD PSD as ADSL per T1.413 Issue 2. This same PSD
mask is defined in asymmetric SM Class 5. The spectral compatibility conditions for ADSL are defined in
Section A.7. Since both RADSL and ADSL use the same PSD, the spectral compatibility conditions for the
two systems are equivalent.
To quantify the spectral compatibility into the upstream and downstream channels of single-carrier
RADSL, the DFE equations of section A.2.2 are applied to each of the test conditions. Table A. 6 and
Table A. 7 provide the spectral compatibility conditions of other services into RADSL.
A.9
Compatibility with HDSL2
Using the transmit spectrum for HDSL2 (PSD Mask 1) described in BSR T1.418, spectral compatibility
with HDSL2 is verified by performing an Optimal DFE Margin calculation for DFE-based PAM signals to
determine HDSL2 margin in the presence of the proposed signal. The remainder of this section defines
the test parameters.
A.9.1 Evaluation loops
If the new technology is proposed for operation on loops of CSA length or longer, CSA 6 should be used
for margin evaluation. If the new technology is proposed for use only on loops that are shorter than CSA
length, evaluation should be performed using CSA 6, with appropriate adjustments of the new technology
NEXT/FEXT to account for the difference in lengths (for an illustration, see Figure A. 5). Only the new
technology NEXT/FEXT is applied at its shorter length; the other (reference) disturbers are applied at CSA
length.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A.9.2 Reference crosstalk environment
The reference crosstalk environment against which new technologies will be compared is:
Downstream:
24 T1 template disturbers (defined in Annex B of T1.413-1995) and 24 SM class 4
template disturbers.
Upstream:
24 SM class 3 template disturbers and 24 SM class 5 template disturbers.
The simplified T1E1 NEXT model should be used for crosstalk coupling. See Figure A. 1 and Table A. 5.
A.9.3 Margin computation
HDSL2 margin is computed as described for DFE-based PAM signals. The HDSL2 margin with the
proposed technology as a disturber should be compared against calculations under the reference
crosstalk scenarios, with the same loop and noise coupling models used in each case. The following
crosstalk combinations, using the loop topologies described in Section A.9.1, should all be used to
compute margins with the proposed technology as a disturber:
Downstream:
(a) 49 new technology NEXT/FEXT,
(b) 24 T1 template NEXT/FEXT + 24 new technology NEXT/FEXT,
(c) 24 new technology NEXT/FEXT + 24 SM class 4 template NEXT/FEXT,
and
(d) 12 T1 template NEXT/FEXT + 12 SM class 4 template NEXT/FEXT + 24 new technology
NEXT/FEXT.
Upstream:
(a) 49 new technology NEXT/FEXT,
(b) 24 SM class 3 template NEXT/FEXT + 24 new technology NEXT/FEXT,
(c) 24 new technology NEXT/FEXT + 24 SM class 5 template NEXT/FEXT,
and
(d) 12 SM class 3 template NEXT/FEXT + 12 SM class 5 template NEXT/FEXT + 24 new technology
NEXT/FEXT.
Spectral compatibility requires that the computed HDSL2 margin, using each of the eight test crosstalk
combinations specified above, is not more than HDSL2_delta dB lower than the HDSL2 margin for the
corresponding reference case. The values of HDSL2_delta for the various test crosstalk combinations
shall be as specified in Table A. 8. The comparisons shall be done under the following conditions:
•
Required SNR (SNR_req) for HDSL2 is 27.7 dB – 5.1 dB for coding gain (i.e. 22.6 dB).
•
Model resolution of at least 500 Hz (FDELTA <= 500) should be used for the HDSL2 margin
calculations.
•
Noise floor is –140 dBm/Hz.
•
FSAN combination method for mixed disturbers.
•
The simplified T1E1 NEXT model is used for the noise-coupling model.
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A.10 Compatibility with 2B1Q SDSL
Using the transmit spectrum described in section A.10.4.1, spectral compatibility with 2B1Q SDSL is
verified by performing an optimal DFE calculation for DFE-based PAM signals to determine SDSL margin
in the presence of the proposed signal.
A.10.1 Evaluation loops and performance levels
Three performance classes of 2B1Q SDSL are determined:
1. 2B1Q SDSL at 400kb/s at reaches up to TBDkft.
2. 2B1Q SDSL at 1040kb/s at reaches up to TBDkft.
3. 2B1Q SDSL at 1552kb/s at reaches up to TBDkft.
Evaluation needs to consider all three performance classes regardless of downstream or upstream
direction.
A.10.2 Reference crosstalk environment
The reference crosstalk environment against which new technologies will be compared is:
a. 49 SM class 2 template disturbers for 2B1Q SDSL at 400kb/s.
b. 49 SM class 8 template disturbers for 2B1Q SDSL at 1040kb/s.
c. 49 SM class 7 template disturbers for 2B1Q SDSL at 1552kb/s.
Either the simplified T1E1 NEXT model or the two-piece Unger model may be used for crosstalk
evaluation. See Figure A. 1 and Table A. 5.
A.10.3 Margin computation
2B1Q SDSL margin is computed as described for DFE-based PAM signals. The computed margin for
2B1Q SDSL against the proposed technology as a disturber should be compared against a calculation
using the same loop and noise coupling models for
a. 49 new technology NEXT, or
b. 24 reference disturbers + 24 new technology NEXT.
Spectral compatibility requires that computations for the maximum allowable numbers of the proposed
technology (based on self-crosstalk limitations) disturbing 2B1Q SDSL produces no lower 2B1Q SDSL
margins than 49 SDSL SELF NEXT by 0.6dB. Required SNR for SDSL is 21.4dB. Model resolution of at
least 500 Hz should be used for the 2B1Q SDSL margin calculation.
A.10.4 2B1Q SDSL Technology Specification
2B1Q SDSL uses 4-PAM modulation. Symbol rate, baud rate, and power spectrum density at both HTUC
and HTUR transceivers are the same. Coding is optional. 2B1Q SDSL system may vary its data rate from
64kb/s to 2320kb/s. The granularity of data rate is not specified, but is expected to be in the range of 8kb/s
and 64kb/s. The startup process is specified in ITU G.991.1.
A.10.4.1 Power Spectrum Density
The power spectrum density of 2B1Q of SDSL systems at HTUC or HTUR can be approximated by
th
filtering a square pulse at the symbol rate followed by a 4 order Butterworth filter at 240/392 of the
symbol rate. It is described by:
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
2
  f

 sin  f  
sym 
2.7 × 2.7  
1
 ×
SDSLu ( f ) =

135 × f sym  f


f sym 

f

 1+ 
 240
f sym

 392






8
.
where fsym is the symbol rate. The PSD of 2B1Q SDSL at several data rates are plotted in Figure A. 7 and
Figure A. 8.
The actual PSD may differ from this template specification. However, it shall comply with the spectrum
management class (SMC) templates. Table A. 9 shows the spectrum management classes that 2B1Q
SDSL shall comply with.
A.10.4.2 Performance
Performance of 2B1Q SDSL at 160kbps is covered by basic rate ISDN, and performance of 2B1Q SDSL
at 784kb/s is covered by HDSL. Performance in this section does not apply to 2B1Q SDSL at these two
data rates.
At a given data rate, performance of 2B1Q SDSL is specified as a target reach on a test loop in the
-7
presence of crosstalk noise. At the target reach, SDSL transceivers shall achieve 10 bit error ratio (BER)
-7
with 3dB of noise margin. The required SNR at 10 bit error ratio for 4-PAM signal with 0dB of noise
margin is 21.4dB. This section describes the test loop, test setup, crosstalk noise, and reach target.
A.10.4.2.1 Test loops
Test loop lengths are in kilofeet units of Equivalent Working Length (e.g., 26AWG, with no bridge taps).
Test loops are given in Table A. 10. The parameters for the loop model are generated using the curve fit
documented in section B.3.1.7.2 and Table B. 2 of this standard.
A.10.4.2.2 Test Setup
Test setup is the same as for the HDSL2 noise impairment test given in BSR T1.418.
A.10.4.2.3 Crosstalk noise
The simplified 49 disturber NEXT model is used and is expressed by
NEXT49 = 8.818 × 10 −14 × (n / 49) 0.6 × f
3/ 2
where n is the number of disturbers. See Figure A. 1 and Table A. 5.
The crosstalk noise for 2B1Q SDSL at both HTUC and HTUR is specified as 49 SELF NEXT (n=49). The
PSD used for producing NEXT noise is specified in Section A.10.4.1.
A.10.4.2.4 Reach target
2B1Q SDSL shall have the target reach specified in Table A. 10 in the presence of crosstalk source
specified in Section A.10.4.2.3.
A.10.4.3 Return loss
The minimum return loss with respect to 135Ω over a frequency band of 1kHz to 1MHz shall be 12dB from
40kHz to fsym/2, with a slope of 20dB/decade below 40kHz and above fsym/2. An example of minimum
60
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
return loss for 784kbps system is shown is Figure A. 9.
A.10.4.4 Longitudinal Balance
2B1Q SDSL system shall meet the following longitudinal requirement:
•
40dB between 20KHz and fsym/2, with a slope of –20dB/decade below 20kHz and above fsym/2.
The requirement for784kbps system as an example is shown in Figure A. 10.
A.11 Combination of crosstalk sources: composite crosstalk model
See Section B.4.3.
A.12 Customer end-point separation
It is often the case that TU-Rs are in separate locations, and in this case the NEXT from a remote
upstream transmitter is attenuated by a length of cable before it couples into a remote downstream
receiver. To account for this customer end-point separation, calculations of remote receiver performance
shall it is reasonable to assume that attenuate the NEXT from remote transmitters is attenuated by a 150foot section of 24 AWG cable before this NEXT couples into the remote receiver. Furthermore, Thethis
model assumes used to derive this attenuation is only valid when the transmission paths closest to TU-Rs
are over distinct cable sheaths. Therefore thisthe customer end-point separation is not cumulative, and
instead it is equal to a fixed 150-foot length between the disturbed receiver and every crosstalker.
Numerous and detailed calculations during the course of this standards development have indicated that
utilizing this customer end-point separation model does not change compatibility results much from the
case of a simplified zero end-point separation (collocated customer end-points). Therefore, the simplified
zero end-point separation model shall be used.
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Table A.1 - Code for DFE PAM SNR computation
/* OPTIMAL DFE PAM SNR computation */
float pamsnr (
float *signal, /* array of received signal psd samples (resolution =
FDELTA Hz)*/
float *noise,
/* array of received noise psd samples (resolution =
FDELTA Hz) */
int
baud, /* PAM baud rate expressed in units of FDELTA (frequency
resolution) */
int
end, /* Maximum number of frequency samples */
int
in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in
linear units */
{
int
i,cnt;
float
snr,temp;
temp = 0;
i = 0;
cnt = 0;
while(i<end && i < baud) {
if( in_dB == 1 ) {
if (2*baud-i) < end){
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud}))+
pow(10.0,0.1*(signal[2*baud-i]-noise[2*baud-i]))+
+1.0);
) else if (i+baud < end){
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
+1.0);
} else if (baud-i < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[baud-i]-noise[baud-i]))+
+1.0);
} else {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+1.0);
}
} else {
if (2*baud-i < end {
temp += log(signal[i]/noise[i]+signal[baud-i] /
noise[baud-i] +signal[i+baud]/noise[i+baud] +
signal[2*baud-i]/noise[2*baud-i] +1.0);
} else if (i+baud < end){
temp += log(signal[i]/noise[i]+signal[baud-i] /
noise[baud-i] + signal[i+baud]/noise[i+baud] +1.0);
} else if (baud-i < end){
temp += log(signal[i]/noise[i]+signal[baud-i] /
noise[baud-i]+1.0);
} else {
temp += log(signal[i]/noise[i] +1.0);
}
}
cnt ++;
i++;
}
temp /= (float) cnt;
snr=10.0*temp*log10(exp(1.));
return(snr);
/* dB */
}
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Table A. 2 - Code for DFE QAM/CAP computation
/* OPTIMAL DFE QAM/CAP SNR computation */
float qamsnr (
float *signal, /* array of received signal psd samples (resolution =
FDELTA Hz)*/
float *noise,
/* array of received noise psd samples (resolution =
FDELTA Hz) */
int
baud, /* PAM baud rate expressed in units of FDELTA (frequency
resolution) */
int
end, /* Maximum number of frequency samples */
int
in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in
linear units */
{
int
i,cnt;
float
snr,temp;
temp = 0;
i = 0;
cnt = 0;
while(i<end && i < baud) {
if( in_dB == 1 ) {
if (i+3*baud < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
pow(10.0,0.1*(signal[i+2*baud]-noise[i+3*baud]))+
pow(10.0,0.1*(signal[i+3*baud]-noise[i+3*baud]))+
+1.0);
} else if (i+2*baud < end){
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
pow(10.0,0.1*(signal[i+2*baud]-noise[i+2*baud]))+
+1.0);
} else if (i+baud < end) {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+
pow(10.0,0.1*(signal[i+baud]-noise[i+baud]))+
+1.0);
} else {
temp += log(pow(10.0, 0.1*(signal[i]-noise[i]))+1.0);
}
} else {
if (i+2*baud < end){
temp += log(signal[i]/noise[i] +
signal[i+baud]/noise[i+baud]+
signal[i+2*baud]/noise[i+2*baud] +1.0);
} else if (i+baud < end){
temp +=log(signal[i]/noise[i] +
signal[i+baud]/noise[i+baud]+1.0);
} else {
temp += log(signal[i]/noise[i] +1.0);
}
}
cnt ++;
i++;
}
temp /= (float) cnt;
snr=10.0*temp*log10(exp(1.));
return(snr);
/* dB */}
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Table A.3 - Matlab code to set-up ADSL margin computation
-----------------------------------------------------------------------------% ADSL_margin.m
% This program shows how to set up the various parameters required to compute
% the DMT margin as per the spectrum management standard.
% Define_Xmit_PSD, Define_Loop_Function and Define_NEXTFEXT_Noise are user
% supplied functions
% Emphasis has been put on code portability rather than code efficiency
% set the direction, bit rate and loop length of the ADSL system
Direction = 'DN'; % either UP or DN
BitRate = 5184e3; % in bps
LoopLength = 9000; % in feet
% assumes that all signal computations are in the linear domain for the margin
% computation. This parameter should not be changed
in_dB = 0;
% set up the number of carriers and specifies the xmit PSD.
% The Xmit PSD will be the template corresponding to SMC5.
if (strcmp(upper(Direction),'DN'))
% carriers 33 to 255
CarrierStart = 33;
CarrierEnd = 255;
Carriers = [CarrierStart:CarrierEnd];
XmitPsd='SMC5-DN';
elseif (strcmp(upper(Direction),'UP'))
% carriers 6 to 31
CarrierStart = 6;
CarrierEnd = 31;
Carriers = [CarrierStart:CarrierEnd];
XmitPsd='SMC5-UP';
else
error(['Invalid direction,' direction])
end;
% set up the Coding gain, min and max number of bits per carrier, number of bits
% per symbol and frequency separation of the ADSL carriers
CODING_GAIN = 3.0;
MINBITS = 2;
MAXBITS = 14;
CarrierSpacing=4312.5;% Hz takes into account cyclic prefix
NPointsPerCarrier=4; % number of frequency points per carrier
BitsPerSymbol= BitRate/(4000/ NPointsPerCarrier); % baud rate is 4000 symbols/second
Deltaf=CarrierSpacing/NPointsPerCarrier;
% define frequency vector using NPointsPerCarrier frequency points distributed
% uniformly over each carrier
if (rem(NPointsPerCarrier,2)==0) % check remainder of a division by 2
% NPointsPerCarrier is even
StartFreq=CarrierStart*CarrierSpacing - (NPointsPerCarrier-1)*Deltaf/2;
EndFreq = CarrierEnd*CarrierSpacing + (NPointsPerCarrier-1)*Deltaf/2;
else
% NPointsPerCarrier is odd
StartFreq=CarrierStart*CarrierSpacing - (NPointsPerCarrier-1)/2*Deltaf ;
EndFreq = CarrierEnd*CarrierSpacing + (NPointsPerCarrier-1)/2*Deltaf ;
end;
Freq=StartFreq:Deltaf:EndFreq;
% define Xmit PSD
% psd_xmit is a vector that contains the value of the PSD corresponding
% to each frequency point defined in Freq. The units should be watts/Hz
% i.e. 10*log10(psd_xmit)+30 is in dBm/Hz.
% The Xmit PSD of ADSL should be the template of SMC5 either UP or DN
% i.e. -40dBm/Hz for example in the passband of the downstream direction
psd_xmit = Define_Xmit_PSD(Freq,XmitPsd)
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% define cable function
% loop is a vector that contains the magnitude squared of the insertion loss
% corresponding to each frequency point defined in Freq. The loop function
% should be on a linear scale.
% The termination impedance should be 100 ohms.
% See B.3.1.7 for insertion loss
% See table B.2 and B.6 for example cable parameters
loop = Define_Loop_Function(Freq,AWG26_length,termination_impedance)
% signal PSD is xmit PSD attenuated by loop
signal = psd_xmit .* loop;
% background noise at -140 dBm/Hz converted to a linear scale of Watts/Hz
Background_Noise = 1e-3.*(10.^(-140/10));
% NEXTFEXTNoise is a vector that contains the value of the NEXT plus FEXT PSD
% corresponding to each frequency point defined in Freq. The units should be watts/Hz
% see section 5.2 and associated tables and figures for a description of the SM
classes.
NEXTFEXTNoise =
Define_NEXTFEXT_Noise(Freq,etc,...)
% noise = NEXT + FEXT + Background noise
noise = NEXTFEXTNoise + Background_Noise;
[snr_margin resolution]= dmtmrgnTA3(signal,
in_dB, CODING_GAIN, MINBITS, MAXBITS);
noise,
BitsPerSymbol,
1,length(Freq),
% display the results used in the margin computation
fout=1; % redirect to std output
fprintf(fout,'\nADSL PARAMETERS: C.G.: %ddB Carriers: %d-%d MinBits: %d MaxBits:
%d',CODING_GAIN,Carriers(1),Carriers(length(Carriers)),MINBITS,MAXBITS);
fprintf(fout,'\nSIMULATION PARAMETERS:');
fprintf(fout,'\nNPointsPerCarrier:
%d
Freq.
Resolution:
%5.2f
Hz',NPointsPerCarrier,Deltaf);
fprintf(fout,'\nADSL
Direction:
%s
BitRate:
%dbps
LoopLength:
%dfeet',Direction,BitRate ,LoopLength );
fprintf(fout,'\nADSL MARGIN (resolution: %3.2fdB) : %2.1fdB\n',resolution,snr_margin);
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table A. 4 -- Matlab Code to compute a DMT margin
function [snr_margin, MARGIN_STEP]= dmtmrgnTA3(signal, noise, rate, f_start, f_end,
in_dB, CODING_GAIN, MINBITS, MAXBITS)
% signal -> psd of signal
% noise -> psd of noise
% rate
-> # bits per symbol
% f_start -> index of starting freq.
% f_end
-> inded of ending freq.
% in_dB = 1 if PSD is in dB or in_dB = 0 if PSD is in linear units
% CODING_GAIN -> self explanatory
% MINBITS -> min number of bits per carrier
% MAXBITS -> max number of bits per carrier
% snr_margin <- computed DMT margin with a resolution of MARGIN_STEP
%
%
%
%
%
%
%
%
%
%
%
Assumes that the margin is MAXIMUM_VALUE - MARGIN_STEP
Starts a brute force search from this point downward
Computes the capacity and compares to the target bit per symbol rate
If not enough, decrease the margin by MARGIN_STEP
if enough, then has found the correct margin
In case the margin is greater than MAXIMUM_VALUE - MARGIN_STEP,
will add 10 dB to MAXIMUM_VALUE until finds an initial guess that
is larger than the margin and proceeds as above.
Emphasis has been put on code portability rather than code efficiency.
To achieve greater speed, one can vectorize the various loops and use a root
finding algorithm such as, for example, fzero in matlab
MAXIMUM_VALUE =7.1;
% resolution of the margin computation will affect the speed
MARGIN_STEP = 0.1;
SNRGAP = 9.75 - CODING_GAIN;
snr_margin = MAXIMUM_VALUE;
firstpass = 1;
totcap = 0;
while (totcap < rate)
snr_margin = snr_margin - MARGIN_STEP;
% compute capacity
totcap = 0;
for j = f_start:1:f_end,
if (in_dB)
snr = signal(j) - noise(j);
else
snr = 10*log10(signal(j) / noise(j));
end;
delcap = log(1.+10.^((snr - snr_margin - SNRGAP)/10))/log(2);
if (delcap > MAXBITS) delcap = MAXBITS; end;
if (delcap < MINBITS) delcap = 0; end;
totcap = totcap + delcap;
end;
if ((totcap > rate) & (firstpass == 1))
snr_margin = snr_margin +10;
totcap = 0;
else
firstpass = 0;
end;
end;
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Table A. 5 – Data Points for Unger NEXT Model (see Figure A. 1)
No. of
Disturbers
1
10
49
0.2
88
80
74
2
82
75
70
Frequency in kHz
20
200
76
61
70
56
66
52
2000
46
42
38
Table A. 6 – Spectral Compatibility into downstream single-carrier RADSL
Crosstalk
Condition
Distance (26
AWG wire)
Downstream
Frequency Band
Symbol Rate
DFE Margin
Table A. 7 – Spectral Compatibility into upstream single-carrier RADSL
Crosstalk
Condition
Distance (26
AWG wire)
Downstream
Frequency Band
Symbol Rate
DFE Margin
Table A. 8 - HDSL2_delta (in dB) for various test crosstalk combinations
(a)
(b)
(c)
(d)
Downstream
49 New Technology
24 T1 + 24 New
24 New + 24 SMC 4
12 T1 + 12 SMC 4
+ 24 New
Upstream
49 New Technology
24 SMC 3 + 24 New
24 New + 24 SMC 5
12 SMC 3 + 12 SMC 5
+ 24 New
0.0
0.0
0.0
0.7
0.3
0.3
0.0
0.4
Table A. 9 - 2B1Q SDSL data rate and associated spectrum management classes
2B1Q SDSL data rate (kbps)
Data rate ≤ 288kbps
288 < data rate ≤ 528
528 < data rate ≤ 784
784 < data rate ≤ 1168
1168 < data rate ≤ 1552
SMC
1
2
3
8
7
Table A. 10 -– 2B1Q SDSL reach target at sample data rates
SDSL data rate (kb/s)
400
1040
1552
Reach target (kft, EWL)
13.0
9.0
7.0
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
90
1
Simplified Model for 49 Disturbers
(57 dB at 80 kHz; -15 dB/Decade)
10
80
-6
1% NEXT Loss - dB
49
70
-5
Number of
Disturbers
-4
-15
60
-15
-14
slope
50
-15
40
20000 Hz
30
100
1000
10000
slope
100000
1000000
Frequency - Hz
Notes:
1.
Terminated with cable characteristic impedance Z0 at each frequency
2.
NEXT disturbers in the same cable binder unit of 50 pairs
3.
See Table A. 5 for data points of Unger NEXT model
Figure A. 1 – Unger NEXT model and simplified NEXT model of 1% NEXT for 18kft of
22GA PIC
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Ref-C
Ref-R
TU-C
TU-R
New-C
New-R
Z kft, 26 gauge
Figure A. 2 – Crosstalk into a Basis System: NEXT and FEXT
Reference
FEXT
Reference
NEXT
+
TU-C
Z kft
New
FEXT
+
TU-R
New
NEXT
Figure A. 3 – Simulation Model for Reference and New Crosstalk into Downstream
Receiver
Ref.-C
Ref. -R
TU-C
TU-R
New-C
New-R
Z-Y
kft
Y kft
Z kft
Figure A. 4 – Crosstalk into Basis System: NEXT & FEXT with reduced new loop length
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Reference
FEXT
Z kft Coupling
TU-C
+
New
FEXT
Y kft Coupling
Reference
NEXT
Y kft
+
Z-Y kft
+
TU-R
New
NEXT
Figure A. 5 – Simulation Model for Self- and New Crosstalk into Downstream Receiver
with reduced new loop length
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Reference
disturbers, test
loop, and
basis system
performance
class
New system is not
compatible with basis
system and not spectrally
compatible in general
No
2. Upstream:
replace
appropriate
reference
disturbers with
new techology
disturbers
1. Calculate
reference
performance level
of basis system,
up & downstream
3. Calculate new
upstream
performance level
of basis system
New level >
(ref. level - delta)?
Yes
6. Go to new
loop/
performance
scenario
8. Go to new
basis system
No
All
loop/performance level
scenarios for this basis
system calculated?
No
4. Downstream:
replace appropriate
reference disturbers
with new techology
disturbers
Yes
All
basis systems
calculated?
Yes
New level >
(ref. level - delta)?
5. Calculate new
downstream
performance level
of basis system
No
6. Move new TU-R
interferer 500 feet
closer to CO
7. Note total new
distance Y
between CO and
new TU-R
interferer
Yes
New system is
considered
spectrally
compatible
If steps 6 and 7are employed for any loop/
performance scenario, the new system is
considered spectrally compatible only when
deployed on loops Y' or shorter, where Y' is the
smallest value of Y required for any loop/
performance scenario of all basis systems.
Figure A. 6 - Process flow for spectral compatibility calculations
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Figure A. 7 – 2B1Q SDSL PSD at several data rates
Figure A. 8 - 2B1Q SDSL PSD at several data rates
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Figure A. 9 - Minimum return loss for 784kbps system
Figure A. 10 - Longitudinal balance for 784kbps system
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex B: Loop Information
(informative)
B.1
General
B.1.1 The loop environment
There are 700 million metallic twisted pair cables delivering communications services to customers
around the world. It is predicted that most of this embedded base of copper wire will eventually be
replaced with wider bandwidth transmission media such as optical fiber and coaxial cable. However,
4)
twisted pair copper wire will be the main method of delivery for several years to come .
Recent years have seen the increase in demand for customer information bandwidth escalate dramatically
from 3 kHz analog voice services to digital services requiring several megabits per second. Advances in
integrated circuit density, digital signal processing techniques and information compression algorithms are
resulting in the introduction of ever higher bandwidth twisted pair transmission systems that can transport
these new services.
These new systems must fit into a outside loop transmission environment with several existing
transmission systems and other systems that may be introduced later. For the voice frequency services
the twisting of the wire pairs and construction of the cables such that no two pairs traveled together for
very long, helped to control crosstalk coupling. Interference between pairs was held to acceptable levels.
As signal bandwidth increases, the crosstalk coupling between pairs increases at the same time as the
transmission loss increases making the circuits more susceptible to interference. Interference can come
from other transmission systems of the same kind or from different type systems that overlap the signal
spectrum.
The 1.5Mb/s T1 line system was originally developed for application in the intra-office cable plant whose
construction is very carefully controlled. To control crosstalk interference between T1 systems, T1 signals
in the two directions were placed on separate pairs located in different binder groups that had shields
between them. T1 repeaters were spaced and placed to minimize differences in signal levels. When T1
systems began to be deployed in the customer outside loop plant, the situation became much more
challenging. As will be described later, the loop plant is designed and constructed to deliver voice services
to customers at acceptable quality and minimum cost. In recent years, many T1 lines have been deployed
in the outside plant to deliver 1.5 Mb/s services to business customers. The engineering design and
construction of these lines is a challenge in minimizing interference and cost.
Over the years several high bandwidth analog carrier systems were also deployed in the outside plant with
mixed results for compatibility and interference.
The use of the loop plant to transport high rate digital signals was not envisioned at the beginning. Indeed,
for over 100 years the loop plant has been optimized for the reliable delivery of voice frequency services at
lowest cost and acceptable quality. In the last several years, the design of new loop plant has been
modified slightly to ease the introduction of digital transmission.
As newer digital transmission systems have been developed for the loop plant, each one has been
subjected to hard scrutiny for potential interference with like systems. ISDN Basic Access digital
subscriber line (DSL) systems had to account for other DSLs in the cable and existing systems like T1
lines and Digital Data Service (DDS). In turn, high-bit-rate DSL (HDSL) had to show compatibility with
DSL, T1 and DDS. Asymmetric DSL (ADSL) had to account for all of the above.
Development and deployment of these new transmission systems is very costly in time and money. It
would be very desirable to predict how a system will perform before it is actually built. Testing a system
–––––––
4)
As will be discussed later, not all telephony wire is made up of twisted pairs. Some single-pair aerial/overhead drop
wire uses parallel/flat/non-twisted conductors for lengths up to 700 feet. Also, not all of the wire is copper.
Copperclad steel and copper-cadmium mixes are used where strength is needed in drop cables.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
against all reasonable cases of interferers on all reasonable loop configurations is not feasible. To test the
performance of a system and the mutual interference with other systems, a combination of analysis,
simulation, laboratory and field testing is done. Analysis and computer simulation are the first steps in
developing new systems. Needed are accurate models for: the transmission systems (the new proposed
system as well as the other possible conflicting systems), the primary transmission constants of the
cables, and the crosstalk coupling for the frequency spectrum, representative set(s) of test loop
topologies, a reasonable set of interferer systems (types and numbers and combinations), broadband
background (thermal) noise models, impulse noise models, etc.
B.1.1.1 Background noise
According to a recent Bellcore study, the residence background noise level in the band of interest could be
at a level of around -140 dBm/Hz. This background noise level is higher than that achievable by a receiver
front end electronic circuit. On the other hand, attention still has to be paid to make the receiver front end
electronic circuit noise level below the assumed -140 dBm/Hz level.
B.1.1.2 Impulse noise
Impulse noise is of major concern for higher speed twisted wire pair type systems, especially due to the
higher subscriber loop loss. Compared with the very weak received signal, a majority of impulses collected
by the same Bellcore study would cause receiver detection error. It has been shown that forward error
correction coding is effective at minimizing the impact of impulse noise. The effect of impulse noise needs
to be included in transmission performance simulation. Forward error correction codes are typically used
to handle impulse noise. Section tbd and Appendix tbd describe the results of field measurements of
impulse noise.
B.1.1.3 Radio frequency interference (RFI)
In addition to coupling within a cable, radio frequency interference (RFI) also becomes a concern as the
signal frequency increases, the wavelength shortens and approaches the dimensions of the cable
structure components, and overlaps radio services. Radio frequency energy may radiate from a wire pair
and interfere with radio services (egress). Radio frequency energy may enter a wire pair and interfere with
the wire pair transmission system (ingress).
Modern wire pair systems operate with signals in a "metallic" mode where currents in the two conductors
are equal and opposite in direction thus tending to reduce radiation either entering or exiting. Currents that
travel on both conductors in the same direction are said to be in a "longitudinal" mode. These longitudinal
currents are much more likely to radiate. External radio frequency fields tend to couple to the pair in the
longitudinal mode. The balance of the individual wire pair conductors and the connecting circuitry relative
to the environment determines the conversion of the normal metallic signal conduction to longitudinal
currents and the conversion of longitudinal currents to metallic signals.
B.1.1.4 Structural cable faults
Structural cable faults (degraded splices, shorts, opens, grounds, crossed pairs, conductor pair
reversals, … do occur and will prevent a transmission system from working. Such mechanical cable faults
are beyond the scope of this effort.
B.1.1.5 The loop environment
Early digital twisted pair transmission systems needed to have cables with very simple make-ups. T1
carrier system was originally intended for the interoffice cable plant to replace the loaded cable voicefrequency pairs. Interoffice voice-frequency cable used only one gauge of wire. No bridged taps were
allowed. The T1 repeater spacing matched that of the loading coils starting at 3000 feet and at intervals of
6000 feet afterwards to take advantage of the loading coil mounting locations between central offices.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
When T1 carrier began to be deployed in the subscriber loop plant for connections to Digital Loop Carrier
systems and for high-capacity digital services to business customers, it encountered a much tougher
environment in terms of cable makeup. Repeater spacing had to be reduced to 4000 or 3000 feet. Bridged
taps had to be removed.
Modern digital twisted pair transmission systems are intended to deliver digital information to average
households and businesses through the copper loop plant as it exists without modifying the makeup.
Depending on the desired information rate and noise environment, the serving distance from a central
office (CO) or a remote terminal (RT) could be different. To have a low overall cost, the deployment
procedure for a new digital transmission system should be as simple as possible. In other words, it would
be ideal if the system terminals could simply be installed on the selected loop and turned on. Additional
engineering work such as field trips and loop qualification should be avoided. From the telephone
company point of view, a service using the transmission system as a delivery vehicle should be prequalified for a known type of loop plant, such as resistance design range, CSA, etc. Any loop qualification
should be on a bulk area basis, not for each individual loop.
B.1.1.6 Telephone cable and subscriber loop structures
This section describes the nature of the structure of the outside loop plant in the US.
5)
A subscriber loop consists of sections (typically 500 feet long) of copper twisted pairs of different gauges.
A section of a subscriber loop could be aerial (hung on poles), buried (directly in the ground), or
underground (pulled inside protective conduit). Electrical joints, called splices, for cable sections could be
made on a telephone pole for aerial cables or in a manhole for underground cables. These splices are not
soldered as in most electronic circuits, but are made with some form of compression technique. For many
years the most common splice was made by stripping the insulation from the wire ends, hand twisting the
bare wire ends together and covering the splice with tape. Modern splices use connectors which use a
hand compression tool to generate the force to penetrate the insulation and make a solid connection.
Properly performed, the compression splice results in a metal to metal connection that is impervious to
liquid or gas.
Twisted pair cables have large cross sections near the central office. There could be 12, 13, 25, 50 or 100
10, 25 or up to 50 pairs in a cable binder group and up to 50 binder groups per cable. Binder groups are
combined to form cables of from 50 pairs to several thousand pairs. Cables share a common electrical
and physical structure, with metallic electrical sheathing and plastic covering. Cables intended for
application near the customer premises may have fewer pairs.
Functionally, a subscriber loop can be divided into portions that belong to feeder cable, distribution cable,
6)
and drop wire . Wiring inside the customer premises that connects to the drop wire at the network
7)
customer interface does not count as part of the network loop . The interface between the network loop
and the customer premises wiring is usually made as close as practical to the point of entry to the
premises.
For large multi-tenant buildings and campuses, the network may provide cabling past the minimum point
of entry if permitted by state regulations.
Feeder cables provide links from a central office to a
concentrated customer area. Distribution cables then carry on from feeder cables to potential customer
sites. Since the loop plant construction is completed before customer service requests, distribution cables
are usually made available to all existing and potential customer sites. Hence, it is a common practice to
connect a twisted pair from a feeder cable with more than one distribution cable to maximize the
probability of reaching a potential customer. These multiple connections from a feeder or a distribution
cable to more than one customer location are called "bridged taps." At any one time only one customer is
–––––––
5)
As noted elsewhere, an exception to twisted pair cable is single-pair aerial drop wire.
6)
The term "drop" refers to the drop downward from a pole to a house. Today, most "drop" wires in new construction
are buried.
7)
Of course, any significant lengths of customer premises wiring were included before a transmission system
terminal would obviously contribute to transmission effects. Customer premises wiring can vary from a few feet to
thousands of feet in length. The analysis here assumes that the terminal is at the interface.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
8)
connected and the other taps are left open . As customers connect and disconnect service, these bridged
tap appearances allow the operating company flexibility in the use of the wire.
At voice frequency the transmission effects of bridged taps are relatively small and can be controlled
within acceptable limits by design. The loop plant design rules, such as Resistance Design and CSA, limit
the total bridged tap length to minimize adverse effects, mainly loss and spectrum distortion, on POTS
transmission. From a transmission line point of view, these bridged taps are open open-ended shunts.
Above the voice band the transmission effects become more significant as the frequency increases and
the signal wavelengths approach the tap lengths.
Connection points between feeder cables and distribution cables are commonly located in cabinets, called
Feeder Distribution Interfaces (FDI). Connection points in distribution cables are commonly in pedestals
for underground cables or terminals for aerial cables.
Single aerial drop wires often consist of parallel copper-clad steel wires, sometimes called "flat pairs." For
new construction in recent years, multiple (2, 4 or 6) twisted copper pairs are being used, and are buried if
possible. The drop wire is usually short and has a proportionately small effect on the loop transmission
characteristics except for potential radiation effects. A typical rule of thumb was to allow the drop wiring to
be less than 700 feet or 25 ohms in resistance.
The loop and drop wire potentially could pick up other high frequency radiation noises. It could also could
radiate signals to other high frequency electronic devices.
B.1.2 Loop plant design rules: resistance design
Most of the embedded outside loop plant in the US has been constructed using the guidelines called
Resistance Design or one of its variations.
POTS loop plant design must accomplish three goals: ensure that there is sufficient direct current flow
from the network battery plant to operate station sets, allow dc/low-frequency call process signaling
(dialing, ringing), and limit transmission loss and frequency roll-off to acceptable levels. As mentioned,
telephone cables are designed with different gauges of wire from 26 AWG (thin, with higher resistance) to
19 AWG (thicker with lower resistance). These different gauges are designed to have close to the same
capacitance between conductors per unit length (nominal 0.083 (µf/mile). It happens that limiting the
maximum dc resistance also controls the maximum voice frequency loss and roll off with frequency.
9)
For modern switching systems a maximum loop resistance (DC resistance) of 1500 ohms meets
powering, signaling and transmission objectives. The maximum transmission loss at 1004 Hz is about 9
dB with a roll-off of 6 dB at 2804 Hz. From survey data, the average loop has a dc resistance of 600 ohms
with 4 dB of loss at 1 kHz.
Since distances from a central office to each customer are different, distribution cables of different gauges
are utilized to keep the amount of copper (and dollars) used to a minimum while meeting design
guidelines. To reduce overall loop resistance the end sections of a long subscriber loop tend to have
coarser twisted pairs, whereas finer gauge twisted pairs are used closer to the central office in order to
reduce the diameter of cables in crowded ducts and minimize cost.
However, some customers are so far away from the central office that a direct implementation of twisted
pair cables would result in a dc resistance much higher than the specified 1500 ohms and hence a poor
–––––––
8)
Of course, old fashioned party lines had all the customers on the line tied to the same loop back to the Central
Office.
9)
Different types of switching equipment have different dc loop resistance limits, depending on the battery feed
voltage, the feed resistance, the typical set resistance and the desired minimum current. Step-by-step (SXS)
switches with nominal 48 volt dc batteries (and a 41.5 volt emergency minimum) typically have a 1300 ohm design
limit to achieve a minimum of 23 mAdc through a rotary dial 500-type station set with about 150 ohm dc resistance.
Thus, references are common to "1300 ohm Resistance Design" even though SXS switches have been retired in all
major operating companies. Newer electronic switches typically have a 1500-ohm loop design limit from a nominal
52-volt battery, for a minimum of 20 mAdc through 400 ohm-dc Touchtone station sets.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
voice channel service quality. A procedure of installing loading coils and coarser gauge cables has been
used to extend the central office serving distances for the voice channel. Inductive loading results in a loop
with reduced loss within the voice band for a given gauge of cable and acts as a low pass filter above
10)
11)
3000 Hz. .The original rule for loading cable was 18 kft working length excluding any bridged taps.
Under the current rules, loading coils are installed for cables with a total length exceeding 15 kft including
bridged taps. For the most common loading plan, called "H88" with 88mH inductors, the first loading coil is
installed at 3 kft from the central office. Loading coils are installed every 6 kft thereafter. There may be no
bridged taps between loading coils. Bridged taps on the end sections at the central office and customer
ends may be left connected, up to a total tap length of 6 kft.
B.1.3 Loop plant design rules: carrier serving area (CSA)
DLC systems were originally developed to serve POTS customers beyond the Resistance Design range.
Early DLC systems are based on copper twisted pairs using the 1.5 Mb/s T-carrier, T1-Line technology.
Twenty-four voice channels are carried on one T1-line by use of time division multiplex. With the use of
outside plant digital repeaters/regenerators it is possible to reach out 100 miles. Fiber based DLC systems
are now more popular. Depending on the cost of DLC electronics, it becomes more economical to serve
customers with DLC systems beyond a certain distance. This "prove in" distance has been decreasing as
DLC electronics costs have come down.
The concept of Carrier Serving Area (CSA) engineering guidelines was originally developed in the early
1980's to support 56 kb/s Digital Data Service (DDS) delivery to customers served by DLC systems. The
concept was then revised very slightly and has been used as the guide for voice grade special services
and POTS deployment from the DLC remote terminal. A CSA is roughly defined as a serving distance of 9
kft for 26 gauge loops and 12 kft for 24 gauge loops from a DLC remote terminal the term is also applied
to loops that originate from a central office as well if they meet CSA guidelines. Short loops around a
central office may be consistent with CSA rules even though constructed using Resistance Design rules. A
recent (1991) survey shows that over 60% of DLC loops meet the CSA guidelines. (References in this
document to "CSA" loops or "CSA-type" loops mean wire pairs that meet CSA design guidelines whether
they originate from a central office or from a network remote terminal site.)
As the operating company have deployed DLC systems CSA rules have proven a useful rule of thumb for
HDSL system deployment. They were also chosen as the loop reach target for 6Mb/s ADSL-3 systems.
Carrier serving area wire pairs from the remote terminal of a DLC system to the network interface on the
customer's premises are expected to meet the following design guidelines.
a)
Non-loaded cable only
b)
Multi-gauge cable is restricted to two gauges (excluding short cable sections used for stubbing or
fusing).
c)
Total bridged tap length may not exceed 2.5 kft. No single bridged tap may exceed 2.0 kft.
d)
The amount of 26 gauge cable (used alone or in combination with another gauge cable) may not
exceed a total length of 9 kft including bridged tap.
e)
For single gauge or multi-gauge cables containing only 19, 22 or 24 gauge cable, the total cable
length including bridged tap may not exceed 12 kft.
f)
The total cable length including bridged tap of a multi-gauge cable that contains 26 gauge cable
may not exceed
–––––––
10)
Transmission analysis shows that loss is minimum for certain ratios of resistance, conductance, capacitance and
inductance. Normal cable has a small inductance relative to resistance and capacitance. Lumped inductive loading
achieves close to the ratios for minimum cable loss within the voice band.)
11)
The length of the cable that connects directly from the network to the customer, excluding any bridged taps, is
called the "working length." The working length of the cable corresponds to the dc resistance path from the network
battery to the customer interface. Bridged taps are open-circuited to dc flow.)
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12 −
3(L26 )
kft
9 − LBTAP
where L26 is the total length of 26 gauge cable in the cable (excluding any 26 gauge bridged tap)
and LBTAP is the total length of bridged tap in the cable. All lengths are in kilofeet (kft).
The limits defined above are the maximum permissible outer bounds for a CSA. Nothing in the CSA
concept prohibits the restriction of CSA cables to shorter lengths.
CSA guidelines do not include central office wiring on the switch side of the protector frame, drop wire or
customer building wiring. Detailed statistics for central office wiring or customer premises are not
available. Central office cabling is typically 24 or 26 gauge and may be up to 1 kft long. Customer
premises wiring is typically 26 gauge and may be up to 1kft or more. In some installations, some drop and
wiring inside the customer premises may be part of the network. Although not a transmission requirement,
it is suggested that no more than two gauges of cable be used. Note: all wire gauge references in this
document are American Wire Gauge (AWG).
B.1.4 Distribution area (DA)
A CSA is often further divided into 1 to 6 Distribution Areas (DA). A DA is characterized by a single Feeder
Distribution Interface (FDI) where cross-connects are located. A DA typically serves about 500 customers.
The cable pair group from a RT to all DAs could have different service capacity than that of all DAs
combined. Distribution cables emanating from an FDI usually have a 1.5 to 2 pairs for all potential
customer living units. On the other hand, cable pairs from RT to FDI are installed based on the number of
real customer lines with a smaller spare ratio. This strategy is aimed at an overall minimized installation
12)
cost. The average serving distance of each DA is usually significantly shorter than that of a CSA. A
recent (1991) survey shows that most DA distribution loops are less than 6 to 8 kft in length (26 and 24
gauge respectively) or about 2/3 of the maximum CSA lengths.
B.1.5 Loop statistics
The Resistance Design and Carrier Serving Area design do not define how much of each type of cabling
is actually used. Major surveys of loop topology in the old Bell System were conducted in 1976 and 1983.
B.2
AWG and metric cable: diameters and DC resistance and capacitance
Test loop sets have been developed for AWG and metric cables by T1E1.4 and ETSI for ISDN DSLs,
HDSL, ADSL and VDSL. It is sometimes useful for interested parties who are familiar with one set of
cables, but not the other, to make a rough judgment on which cable in one set compares to which cable in
the other set, if any. One can get into the right ballpark or at least out of the wrong one, by comparing
conductor diameters and diameters, DC (0 Hz) resistance and DC capacitance and insulation materials.
Table B.1 summarizes this data for the most common types of metric and AWG telephony cables. Nontelephony 18 and 20 AWG gauges are also included for comparison because their conductor diameters
are close to 0.8mm and 1.00 mm metric cables.
Attenuation versus frequency data (say at 1 kHz, 10 kHz, 100 kHz, 1 MHz, 10 MHz and 30 MHz) would
allow further contrasts and comparisons. Polyethylene is the most common insulation for feeder and
distribution cables. Polyethylene is a very good dielectric whose properties change very little with
frequency. PVC is the most common insulation for single-pair, overhead/aerial drop wires exposed to the
external environment. PVC dielectric properties vary much more with frequency than those of
polyethylene.
Only AWG 26 PIC and metric 0.40 PE are really close in transmission characteristics.
–––––––
12)
However, the maximum serving distance of a DA might still be very close to that of a CSA.
This is a draft document and thus, is still dynamic in nature.
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B.3
Cable primary constants (RLGC) characterization
It is not feasible to perform laboratory or field tests to represent all likely environments that a transmission
system will encounter. Computer simulation provides a means to test schemes against anything that can
be quantified numerically. Fundamental to the simulation of wire systems are accurate models of the
transmission characteristics of the wire itself versus frequency and temperature.
The primary constants of resistance (R, ohms/km), inductance (L µH/km), capacitance (C, nF/km), and
conductance (G, mho/km) are used to model most transmission lines. Secondary parameters such as
impedance, attenuation and phase or the chain parameters ABCD may be calculated from the primary
constants. These "constants" actually vary in value with frequency, temperature and humidity. To a first
order, signal attenuation increases as the square root of the frequency. Variation of the "constants" and
inductive reactance becoming larger with frequency relative to resistance and capacitance result in the
actual attenuation versus frequency curve being more complex.
Chain parameters ABCD allow cascading of models of two port electronic devices such as wire pairs.
Complex loop topologies with changes of gauge and bridged taps can be constructed with ABCD
matrices. See Figure B.1.
The existing primary constant RLCG models of the common AWG PIC cables were based on careful
measurements and curve-fitting in the early 1970s. They were believed to be valid to 10 MHz and to
represent nominal values for expected manufacturing variations. VDSL and newer proposed schemes
may well have spectral components to 30 MHz. It is vital to have models that reflect the transmission
behavior of the cables in the real world to the frequency and temperature ranges needed.
The primary constant data can be presented in either as R, G, C and G values versus frequency or as
parameters to equations that have been curve fitted to measured data. (See T1E1.4/96-015.)
B.3.1 Transmission-Line Characterization
This section directly addresses the transmission characteristics of twisted-pair phone lines.
Most twisted-pair phone lines can be well-modeled for transmission at frequencies up to at least f<30 MHz
by using what is known as two-port modeling or “ABCD” theory. Such ABCD theory is well covered in
basic electromagnetic texts, but is often not in a form convenient for use in DSLs. Werner presented
essential results of such translation to DSLs in a 1991 JSAC paper “The HDSL Environment” (August
1991) and this section essentially repeats that effect, but provides more detail along with updates based
on various studies in standards bodies that have led to DSL characterization to at least 30 MHz.
Section B.3.1.1 first describes ABCD modeling in general before Section B.3.1.2 specializes to the case of
twisted-pair transmission lines. Section B.3.1.5 considers the special case of bridge taps before Section
B.3.1.6 shows how to compute the transfer characteristics of a subscriber loop consisting of many
sections. Section B.3.1.7 shows how to measure RLCG parameters for loop characterization as well as
lists models for several popular twisted-pair types.
B.3.1.1 “ABCD” modeling
Figure B. 2 shows a general two-port linear circuit. There is a voltage at each port and a current on the
upper path on each port. The voltages and currents will depend on the source (port 1) and load (port 2)
impedances and voltage source(s), but nevertheless always relate to each other by the matrix relationship:
V1 = AV2 + BI 2
V1   A B  V2 
V 
or
⋅   = Φ ⋅  2
 =

I1 = CV2 + DI 2
 I1  C D   I 2 
 I2 
where Φ is a 2 × 2 matrix (nonsingular in all but trivial situations not of interest) of 4 possibly frequencydependent parameters, A, B, C, and D, which all depend only on the network and not on external
connections. The quantities have circuit definitions as in the table below:
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A
open-load voltage ratio
B
shorted-load impedance
C
open-load admittance
D
shorted-load current ratio
The transformation is reversed by Φ −1 so that
V2 
1
 =
I
−
AD
BC
 2
D

− C
− B  V1
−1 V1
 ⋅   = Φ ⋅ I  .
A   I1 
 1
When Φ = I , that is an identity, the network is a trivial connection of the upper path and lower path across
the network, essentially meaning there is no network. A relationship of interest is the ratio
V
T (f ) = 2
V1
where the frequency dependence is shown explicitly for T(f), but not for the other voltages to simplify
V
notation. This ratio depends on the load impedance attached at port 2, or the ratio ZL = Z 2 = 2
I2
T (f ) =
1
A+B
=
ZL
ZL
A ⋅ ZL + B
can be related to a transfer function H (f ) between an input voltage supply VS (with finite internal
impedance ZS ) to the output voltage VL = V2 (across a load ZL = Z 2 ).
VL (f )
V (f ) V (f )
Z1
= H (f ) = L ⋅ 2
=
⋅ T (f ) ,
V2 (f ) VS (f ) Z1 + ZS
VS (f )
V
where Z1 = 1
is the input impedance of the terminated two-port. Z1 must be computed as in the
I1
second equation below and is the ratio of input voltage to current when load ZL is attached at the output.
A cascade of two-ports has a two-port matrix that is the product, in order, of the two ports
V1
VN 
VN 
  = Φ1 ⋅ Φ 2...⋅ Φ N −1 ⋅   = Φ ⋅   ,
 I1 
 IN 
 IN 
allowing for the calculation of transfer functions, and insertion losses of more complicated networks as
long as a two-port model can be found for each subsection in the cascade. The inverse is found by
reversing the order and taking the product of the inverse matrices. The input impedance of the two-port is
B
V
AZL + B
ZL
.
Z1 = 1 =
=
D
I1
CZL + D
C+
ZL
A+
Two-port networks are very useful in the analysis of twisted-pair transmission lines as in the next several
sections. In these sections, the transmission line is modeled as a cascade of two ports that are
characterized by resistance, inductance, capacitance, and conductance per unit length, and by the length
of the transmission-line segment.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
B.3.1.2 Transmission-line RLCG characterization
The two-port characterization of a transmission line derives from the per-unit length two-port model in
Figure B.3. The R, L, C, and G parameters represent resistance, inductance, capacitance, and
conductance per unit length of the transmission line.
A segment of transmission line can be viewed as a cascade of such sections that are infinitesimally small
in length. At any point x, the two-port voltages and currents relate through the differential equations
dV
= (R + jωL ) ⋅ I
dx
dI
−
= (G + jωC ) ⋅ V
dx
−
at any given frequency ω = 2πf .
V and I are phasor quantities representing peak amplitudes of
sinusoids at frequency f (or amplitudes of the complex exponential e j 2πft ). The R, L, C, and G
parameters themselves can vary with frequency, but are presumed constant with respect to length at any
given frequency in the analysis to follow. This set of differential equations is equivalent to the pair of
second-order differential equations
d 2V
dx 2
d 2I
= γ 2 ⋅V
,
2
= γ ⋅I
dx 2
where
γ = α + jβ =
(R + jωL ) ⋅ (G + jωC ) =
Z ⋅Y
is the frequency-dependent propagation constant for the twisted pair, and characterizes the segment of
transmission line. The impedance per unit length, Z, and the admittance per unit length, Y, are also
defined in Figure B.3. The attenuation constant is α and the phase constant is β. The attenuation
constant is very important for twisted-pair. As can be inferred from equations to come, the attenuation of a
twisted-pair is approximated by 8.668 α dB per unit length at the frequency of interest. The phase
constant is related to speed of propagation on the twisted pair: At each frequency ω = 2πf , a sinusoid
propagates on the twisted pair with phase given by
θ(ω, x ) = ωt − βx
and has envelope amplitude attenuated as e −αx . The wavelength is the length (at fixed frequency and
time) over which the sinusoid undergoes a full cycle and is thus given by
λ=
2π
.
β
Remembering that β is tacitly a function of frequency, different frequencies thus have different
wavelengths. The sinusoidal wave at frequency appears to propagate along the twisted pair at phase
velocity
vp =
ω
,
β
and the phase delay per unit length at this same frequency is τ p = 1v = β ω . When β is a linear function
p
of frequency, the channel is said to have linear phase and the phase velocity and delay are constant over
all frequencies. An example is the case where R=G=0, and then β = ω LC - and (when L and C are
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constant with respect to frequency) means that all frequencies move at the same phase velocity
. Such a transmission line is said to be dispersionless. Note that it is possible to subtract out
vp = 1
LC
the linear (proportional) part of β without introducing error to the time domain response of a cable pair only
when there is no reflected wave being propagated along the pair. Such a condition (no reflection) occurs
only when the impedance of the load is matched to the characteristic impedance of the cable pair. This
may be particularly important when modeling bridged taps. In practical DSLs, dispersionless transmission
never occurs and different frequencies travel with different velocities, leading to dispersion of signal
energy (and to the intersymbol interference). For a dispersive transmission line, it is of interest to
investigate the speed at which a group of frequencies centered around propagates. To understand this
concept of “group” or “envelope” velocity, suppose one investigates the differing speeds of the two
frequencies ω ± ∆ω where the offset or difference is small and the corresponding values of β ± ∆β , but
both have the same amplitude. The resultant sum waveform is
A cos[(ω + ∆w )t − (β + ∆β)x ] + A cos[(ω − ∆w )t − (β − ∆β)x ] = 2 A cos[∆w ⋅ t − ∆β ⋅ x ]⋅ cos[ωt − βx ] ,
the right side this equation is an “envelope-modulated” sinusoid, a product of two sinusoids. When the
phase velocity is constant and there is no dispersion, the phase velocity of the first term on the is the
same as that of the second term, and the phase velocity equals the group velocity. However, when phase
velocity is not constant, the first term moves at a different (often much slower) speed given by ∆ω / ∆β .
This slower speed is the group velocity and in general computed by the inverse of the group delay
τg =
dβ
dω
or
.
Group delay in essence measures the spread in delay between the fastest and slowest moving
frequencies in the immediate vicinity of ω . The greater the group delay, the greater the dispersion in the
transmission line.
The solution to the set of differential equations is easily modeled as the sum of two opposite-direction
voltage/current waves:
V (x ) = V0+ ⋅ e − γx + V0− ⋅ e γx
I (x ) = I0+ ⋅ e − γx + I 0− ⋅ e γx
.
By insertion of either of these solutions into the appropriate first-order voltage/current differential
equations, the ratio of the positive-going voltage to the positive-going current, as well as the (negative of
the) ratio of the negative-going voltage to the negative-going current is equal to a constant characteristic
impedance of the transmission line
V+
V−
Z0 = 0 = − 0 =
I0+
I0−
R + jωL
Z
.
=
G + j ωC
Y
One easily verifies that the R, L, C, and G parameters are equal to
R = ℜ{γ ⋅ Z0 }
1
ℑ{γ ⋅ Z0 }
ω
1  γ 
C = ℑ  .
ω  Z0 
L=
 γ 
G = ℜ 
 Z0 
For twisted-pair transmission and DSLs, it is rare that any of these 4 parameters are zero and so
simplifications in textbooks or other developments that lead to so-called “lossless transmission lines” or
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
“dispersion-less” transmission are not of interest for DSLs. Furthermore, these parameters are frequencydependent for transmission lines and are best determined by measurement as in section B.3.1.7.1.
A segment of transmission line of length d has solution VL = Vd and I L = Id and thus
VL = V (d ) = V0+ ⋅ e − γd + V0− ⋅ e γd
I L = I (d ) = I 0+ ⋅ e − γd + I 0− ⋅ e γd
.
Since the two voltage waves in each direction are related to the same-direction current waves by the
common ratio Z0 , one can solve the above two equations for V0+ and V0− to get:
V0+ = 21 (VL + IL ⋅ Z0 ) ⋅ e γd
V0− = 21 (VL − I L ⋅ Z0 ) ⋅ e − γd
.
By substituting these constants back into the solution in general and evaluating for the voltage and
currents at x=0 in terms of those at x=d , one obtains the following two-port representation
Z0 ⋅ sinh(γd )
 cosh(γd )
V (0 ) 
 V (d ) .

 =  1 ⋅ sinh(γd ) cosh(γd )  ⋅ 

 I (0 )   Z
 I (d ) 

 0

The ABCD entries can be read from the matrix, or equivalently, can be computed from the R, L, C, G
values through in relations for γ and for Z0 . Then, for a given length of transmission line d, the engineer
may model that transmission line as a single “lumped” two-port, replacing the distributed model in Figure
B.3.
Knowing the load impedance so that V (d ) / I (d ) = ZL , the insertion loss then becomes
T =
1
 Z0
cosh(γd ) + 
 ZL

 ⋅ sinh(γd )

=
sech(γd )
1 + Z0 tanh(γd )
L
Z
.
The input impedance of the two-port is V(0)/I(0) or
Z + Z0 ⋅ tanh(γd )
Z1 = Z0 ⋅ L
.
Z0 + ZL ⋅ tanh(γd )
The input impedance of a very long line reduces to Z1 = Z0 , since tanh(γd ) → 1 for large d.
The transfer function in any case becomes
H=
Z0 ⋅ sech(γd )
Z1
T =
.
Z0
Z
Z1 + ZS

ZS ⋅ Z + tanh(γd ) + Z0 ⋅ 1 + Z0 ⋅ tanh(γd )



 L
L
Thus, this type of model applies to the upper example in Figure B.5. Note also there the two-port models
that characterize the source and load. Thus, general principle of multiplying matrices when cascading
two-ports can be directly applied. If several transmission line segments with different R, L, C, and G were
cascaded, then each would have its own two-port model. This situation corresponds to connection of
twisted pairs (splicing) with different gauges.
B.3.1.3 Power for transmission lines
A sinusoid at any frequency on a transmission line represented by the phasor voltage V and phasor
current I has average (rms) power
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P (f ) =
[ ]
1 ℜ VI *
2
.
Figure B.4 shows a simple phase circuit having input current I and voltage V across a load with
impedance ZL = RL + jX L . From basic circuit theory, a sinusoidal current with peak amplitude I delivers
power
2
[ ]
V
2
P (f ) = 21 I RL = 21
RL = 21 ℜ VI *
ZL
,
thus providing interpretation for the relation in the previous equation.
Maximum power is transferred from the power supply to the load when the source impedance is the
conjugate of the load impedance in Figure B.4, ZS,opt = ZL* = RL − jX L . This corresponds to one-half the
total power of the source being dissipated in the load. An example of the use of this maximum-powertransfer result is when one investigates the termination of a twisted-pair transmission line. To transfer
maximum power from the line to the load, the load impedance should be designed to be the conjugate of
the line impedance viewed going back into the line. When the line is long, this impedance will be the
characteristic impedance of the line itself, meaning the best loading is
ZL,opt ≅ Z0* ,
meaning half the power in the line is transferred to the load (with the other half dissipated within the line
itself). Similarly, the optimum driving impedance is the conjugate of the line impedance, which again for
long lines is the characteristic impedance, so
ZS,opt = ZL,opt ≅ Z0*
.
Again, half the source power will be delivered to the line. For a lossless transmission line, the half of the
source power delivered to the line is the same half of power delivered to the load. At higher frequencies,
all transmission lines become lossless and so the best load and source impedances become resistive and
equal to the (real) characteristic impedance of the line.
The condition for maximum power transfer is not the same condition for elimination of reflections (see
next subsection) unless the line is lossless.
B.3.1.4 Reflection coefficients
When the load impedance is equal to the characteristic impedance (and not the conjugate of the
characteristic impedance), the negative-going wave constant V0− = 0 in the above equations. There is
then no reflected wave and all the above relationships simplify somewhat. In practice, such matching is
not likely to occur, and the solution for the differential equation at x=d has general ratio of positive-going
wave to negative-going wave as
V − ⋅ e − γd
Z − Z0
.
ρ= 0
= L
+
γd
Z
V0 ⋅ e
L + Z0
This reflection coefficient is clearly zero when the transmission line is “matched” or terminated in its own
impedance, ZL = Z0 . The return loss is defined as the inverse of the reflection coefficient for any
interface to a two-port, and usually expressed as a positive quantity in decibels. This situation prevents
“bouncing” of signals on a transmission line and thus reduces the dispersion (relative delay) of signals on
the line. In this case of ZL = Z0 , the input impedance is then also Z1 = Z0 . When the transmission line
impedance is approximately real, then the situation of no bouncing corresponds also to maximum energy
transfer in section B.3.1.3 from the line into the matched load. However, when (as usual for twisted pairs),
the line characteristic impedance is complex, then maximum energy transfer occurs when the load is the
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
conjugate of the characteristic impedance, and thus elimination of bouncing does not guarantee maximum
energy transfer for lossy lines. On many lines as the frequency increases, the R and G terms become
negligible and so for these frequencies, maximum energy transfer and elimination of bouncing occur when
the load impedance is matched to ZL = Z0 ≈ L .
C
A similar source reflection coefficient can be written as
ρS =
ZS − Z 0
.
ZS + Z 0
This source reflection coefficient measures the reflected positive-going wave amplitude with respect to a
negative-going wave that flows into the source impedance. The return loss at the interface to between the
source and line is therefore the inverse (in dB) of the source reflection coefficient. Note that the source
impedance that leads to maximum power transfer into the line ZS = Z1* again is not necessarily the same
as that leading to no reflection at the source end. A wave launched from a source will traverse the loop
with phase and group velocities, will be reflected at one end, reflected again at the source end, and so on.
This series of reflections leads to a transient on the loop, unless the loop is terminated in a load
impedance equal to the characteristic impedance of the line. Again when the line can be approximated
over the used frequency range as lossless, and thus having real characteristic impedance, then the
maximum energy transfer and reduction of bouncing objectives coincide.
Formally the return loss of a transmission line is the inverse ratio of reflected power to incident power on
the load (or next section of circuitry). This return loss is simply the square of the reflection coefficient, thus
1
return loss = 10 log10
ρ
2
dB.
B.3.1.5 Characterization of a bridge-tap section – a three-port
For modeling of loops, a bridge-tap is a three-port section, but one of the ports appears as a load
impedance to the line, between the two sections on each side of the bridge tap. Such a situation can be
modeled by the two-port with ABCD matrix shown in the last example of Figure B.5.
The impedance of the tap section Zt is computed according to the formula above for the input impedance
of a section of transmission line terminated with an open circuit ( ZL = ∞ ), which simplifies to
Zt = Z0t ⋅
cosh(γd )
.
sinh(γd )
If the tap were not terminated in an open circuit, then the general formula for the input impedance Z1
(above)of the section should be used.
Circuits with bridge-taps on bridge-taps have an impedance that is calculated by working backwards from
all open taps to points of the taps, modeled as the two-tap section’s impedances in parallel. The resultant
impedance then becomes a termination (load) impedance for the next section working backwards towards
the main transmission pair of interest. While perhaps tedious, the calculation process is straightforward
and recursive.
B.3.1.6 Computation of transfer function
The computation of the transfer functions for twisted-pair transmission lines with multiple sections then
simply becomes a process of multiplying in cascade the corresponding two-port ABCD matrices for each
section. Some examples are provided in Figure B.5, with the corresponding two-port matrices below each
example. The matrices are multiplied left to right in the natural order of appearance in the figure. That is
the overall two port is just
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Φ = Φ 0 ⋅ Φ1 ⋅ ⋅ Φ N
where the source voltage divider is modeled by the two-port
1 ZS 
Φ0 = 
.
0 1 
The final output voltage and current are related by the usual VL = IL ⋅ Z L , which allows the transfer
function to be computed from the ratio
V
H= L .
VS
In the upper example of Figure B.5, a simple section of twisted pair with characteristic impedance Z0 and
propagation constant γ is modeled by the cascade of a two-port matrix description Φ1 for a length d and
the source two-port matrix Φ 0 . This upper example is straightforward application of the two-port theory.
The lower example additionally has a bridge-tap section with Z02 and γ 2 of length d 2 and a second
section of the transmission line with yet a third characteristic impedance and propagation constant. The
two sections of transmission line are modeled as usual, where the impedance and propagation constant
can be computed for each frequency from the known R, L, C, G parameters for each section. The bridgetap section is modeled as a parallel (shunt) impedance that is computed according to the formula for an
open-ended transmission line of length d 2 (if the tap were terminated, the impedance shown need only be
replaced by the more general expression for the inverse of the input impedance of that section). The
overall two-port matrix is simply the product of the 4 two-port matrices shown.
A variety of simplifications are sometimes studied assuming each section is very long and so appears to
be terminated in its own characteristic impedance, leading to expressions for the transfer function and
input impedance in various situations. While sometimes useful for interpretation, with modern day signal
processing analysis tools (for instance, matlab, etc.), it is often easier to compute the transfer function
without simplifying assumptions and then analyze the corresponding results.
B.3.1.7 Relationship of transfer function and “insertion loss”
Transmission engineers sometimes also directly measure the transfer characteristics of a transmission
line at several frequencies. It is hard to measure the transfer function directly because of loading effects,
but it is possible to measure easily the insertion loss, from which the transfer function can be computed if
load and source impedances for the measurement are known.
The insertion loss is computed using a configuration in Figure B.4 by first measuring the voltage Vno , and
then inserting the transmission line at the point where Vno was measured initially and again measuring
VL , the voltage across the load with the line inserted. Thus the insertion loss is
TIL (f ) =
VL (f )
.
Vno (f )
The desired transfer function is instead H = VL / VS so
V
V
ZL
⋅ TIL (f )
H (f ) = no ⋅ L =
VS Vno ZS + ZL
.
Note that when Z1 = ZL , meaning the line is terminated in its own impedance as often in practice, then the
equation can be rewritten in terms of the T(f) as
V V
Z1
⋅ T (f ) ,
H (f ) = 1 ⋅ L =
VS V1 ZS + Z1
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
which also then shows that in the matched-termination case, T (f ) = TIL (f ) . In most cases of interest in
DSL, the line is long and so the source impedance is matched to the characteristic impedance (which
equals the input impedance of the line when the line is long) and all impedances are real over the higher
frequencies used for DSL transmission. In this case, the transfer function is simply 6 dB lower than the
insertion loss. A program that computes transfer functions of twisted pairs using two-port theory is on the
second author’s worldwide web page at http://www-isl.stanford.edu/people/cioffi.
A crucial point of note: When the transfer function is computed for a circuit using RLCG parameters, then
the insertion loss may be computed from the transfer function and is roughly 6 dB higher under the
approximations above. The insertion point is exactly the point at which a transmit power constraint
applies. Thus for instance, input voltage levels computed from a power constraint for a DSL (for instance,
in performance calculation or SNR computation) undergo a channel that is the insertion loss, and not the
transfer function. A common mistake is to compute data rates and performance as if the transmit power
were 6 dB lower by incorrectly using the transfer function instead of the insertion loss.
Transmission lines are characterized in this Appendix by 4 parameters, the Resistance R in Ohms/km, the
Inductance L in Henrys/km, the Capacitance C in Farads/km, and the Conductance G in Mhos/km.
The RLCG parameters in this appendix were provided by the following measurement and curve-fitting
procedures:
B.3.1.7.1
Measurement Procedure
The open-circuit impedance, ZOC , and short-circuit impedance, ZSC , for a length, l , of twisted-pair
transmission line are measured versus frequency. An l=10 m length is used for measurements below 2
MHz and an l=1 m length is used for measurements between 2 MHz and 30 MHz. The characteristic
impedance and propagation constant are computed from the measured impedance according to:
characteristic impedance:
Z0 = ZOC ⋅ ZSC
propagation constant:
 ZSC 
1

γ = tanh −1
 ZOC 
l


From the characteristic impedance and propagation constant, RLCG can be computed as:
R = ℜ(γZ0 )
L=
1
ℑ(γZ0 )
ω
C=
1
ω
 γ 

ℑ

 Z0 
 γ 

G = ℜ

 Z0 
B.3.1.7.2
.
Curve-fitting
Because of error in practical measurements of the impedance, the RLCG values may not follow smooth
curves with frequency so parameterized (smooth) models of RLCG are then fit to the measured values.
The models are:
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
1
R (f ) =
1
4r4
OC
+ aC ⋅ f
2
+
1
4r4
OS
+ aS ⋅ f 2
where r0C is the copper DC resistance and r0s is (any) steel DC resistance, while ac and as are constants
characterizing the rise of resistance with frequency in the “skin effect.”
l 0 + l ∞  f 
 fm 
L(f ) =
b
1 +  f 
 fm 
b
where l0 and I ∞ are the low-frequency and high-frequency inductance, respectively and b is a parameter
chosen to characterize the transition between low and high frequencies in the measured inductance
values.
C ( f ) = c ∞ + c0 ⋅ f −
ce
where c ∞ is the “contact” capacitance and c0 and ce are constants chosen to fit the measurements.
G(f ) = g 0 ⋅ f +
ge
where g0 and ge are constants chosen to fit the measurements.
Further information on smoothing of test measurements is found in ASTM D 4566.
B.3.2 TP1
TP1 is representative of .4 mm or 26-gauge phone-line twisted pair. The specific cable measured was
provided by Bell South to BT and measurements were validated by GTE to produce an acceptable fit
between measured responses and projected insertion loss as computed from the parameters in Table B.
2 using methods in B.3.1.7. The primary constants produced using the parameters are given in Table B.3.
Measurements by Bellcore, whose results are listed in Table B.4 and Table B.5, have indicated that their
results for 26-AWG PIC lines have found strong agreement with the values in the model of this document.
B.3.3 TP2
TP2 is representative of .5 mm or 24-gauge phone-line twisted pair. Parameters found in Table B.6
computed using methods in B.3.1.7. Primary constants are found in Table B.7. Measurements by
Bellcore, whose results are listed in Table B.8 have indicated that their results for 24-AWG PIC lines are in
strong agreement with the values in the model of this document.
B.3.4 22-Gauge Phone-Line Twisted Pair
Measurements by Bellcore for 22-gauge twisted pair are found in Table B.9.
B.3.5 TP3
TP3 is representative of DW10 Reinforced cable with .5 mm copper PVC-insulated conductors, PVCinsulated steel strength member, and Polyethylene sheath. Parameters computed using methods in
B.3.1.7 are found in Table B.10. Primary constants are found in Table B.11.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
B.3.6 FP
FP is representative of ETSI 1.14 mm flat (no twists) phone-line twisted pair. Parameters computed using
methods in B.3.1.7 are found in Table B.12. Primary constants are found in Table B.13.
B.3.7 Category-5 Twisted Pair
Table B.14 gives parameters and Table B.15 gives primary constants computed using methods in 3.1.7
for cables that meet or exceed EIA/TIA Category 5 twisted-pair specifications.
B.3.8 Two-Pair Twisted Drop
Bellcore measured a two-pair twisted service drop cable where the tip and ring are twisted for each pair,
and the two pairs are then twisted together. The conductor gauge is 22 AWG, and the tested cable was of
length 228.6 m = 750 ft. The values for R, L, C and α are averaged over the two pairs since they
exhibited a high degree of symmetry.
The measurements were made with an HP-3577A Network Analyzer connected to an HP-356711A SParameter test set. Measurements at equally spaced log frequencies between 772 kHz and 40 MHz were
obtained.
C and α were directly measured with the network analyzer. The short and open complex impedances,
Zsc and Zoc of the drop cable were also measured with the network analyzer, and the characteristic
impedance, Z0 , calculated in the usual fashion.
It was not feasible to obtain accurate measurements of the conductance G due to a lack of a precise
measurement of impedance angle. This is a result of measurement equipment limitations and the
transformer baluns used to perform the impedance conversion. Additionally, the drop wire jackets directly
contact the pair insulation, hence altering the effective dielectric constant and tan delta. Moreover, the
capacitance is not flat over the entire frequency range. Fortunately, at high frequencies, G is of little
importance for transmission.
Using the relationship
Z0 =
L
,
C
which holds when G<< ω C and R<< ω L, the inductance values are calculated.
Using the relationship
α = 4.34 ∗ R / Z0
the resistance values over the range 0.772 - 40 MHz are evaluated. Results are given in Table B.16.
B.3.9 Two-Pair Quaded Drop
Bellcore measured a two-pair quaded service drop cable where the four conductors comprising the two
pairs are twisted together as a unit. The conductor gauge is 22-AWG, and the tested cable was of length
228.6 m = 750 ft.
Test equipment, measurement setup and the equations used to perform the calculations are identical to
those used in B.3.8 on Two-Pair twisted drop. Results are found in Table B.17.
B.3.10 Flat-Pair Drop
Bellcore measured a flat-pair service drop cable where the tip and ring conductors of a single pair are
parallel. The conductor gauge is 18-1/2 AWG, and the tested cable was of length 291 m = 954 ft.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Test equipment, measurement setup and the equations used to perform the calculations are identical to
those used in B.3.8 on Two-Pair twisted drop. Results are found in Table B.18.
B.3.11 Additional Models
For additional models for European and other types of cable, see ETSI STC TMC6 Permanent Document
# TM6(97)02.
B.4
Cable crosstalk models
Accurate models of crosstalk coupling between pairs in typical cable structures are as vital as the primary
RLGC constant models to system simulation. For the DSL family of transmission systems the limiting
factor on loop range has been crosstalk coupling of signal energy from like or unlike transmission systems
on other pairs in the cable and not from the end-to-end attenuation of the signal.
The current crosstalk models were developed in the 1980s based on computer simulations of the physical
structure of the cables and later compared with measurements. Quantitative crosstalk models for less
than full binder groups or small cables are not available.
B.4.1 Near end crosstalk, NEXT
Telephone twisted pairs are organized in binder groups of 12, 13, 25, 50 and 100 10, 25, or 50 pairs.
Many binder groups share a common physical and electrical shield in a cable. Due to capacitive and
inductive coupling, there is crosstalk between each twisted pair even though pairs are well insulated at
DC. The crosstalk in voice frequency band is minimal, i.e. one can hardly hear the voice energy from an
adjacent pair because the crosstalk loss is usually more than 80 dB, compared with a voice channel loss
of less than 20 dB.
In general the effect of cable crosstalk is minimized not only by the use of good insulation material
between pairs but also by adapting different twist distances among different pairs in a binder group. The
binder groups are also twisted such that no two groups are adjacent for long runs. For digital
communication via digital subscriber line technology, where the signal bandwidth reaches into the MHz
range, the crosstalk is a limiting factor to the achievable throughput.
Near-end-crosstalk (NEXT) is defined as the crosstalk effect between transmit and receive pairs at the
same end of a cable section. In other words, NEXT is a measure of the crosstalk effect between a
transmitter and a receiver at the same end of a twisted pair cable. See Figure B.6. NEXT is usually
considered for full duplex digital subscriber line systems such as DSL and HDSL where the transmit and
receive spectra at each end are the same (or overlap).
NEXT is strongest on the cable at the point where the transmitter of the crosstalking signal puts the signal
on the pair. Any receivers near to this transmitter will receive NEXT as well as the intended signal. The
NEXT path attenuates the unintended signal greatly, but the relevant issue is the signal to noise ratio
between the intended signal and the NEXT. Therefore, NEXT becomes a problem if the intended signal is
attenuated enough. Symmetrical systems such as the ISDN DSL have transmitters at both ends of every
pair on which it is installed. The worst case NEXT is then usually the NEXT produced by a binder group
full of similar collocated transceivers. The NEXT received from similar systems, i.e. DSL to DSL, HDSL to
HDSL, or T1 to T1, in this way is called “self-NEXT.”
For DSL and HDSL, full duplex communication on a single pair is achieved by the use of the echo
cancellation technique. This requires transmit and receive signal paths be as fully separated as practical
13)
with signal processing techniques even though transmit and receive signals share the same frequency
spectra. However, transmit signals in other adjacent pairs are not available to the particular receiver.
–––––––
13)
Two-to-four-wire hybrid circuits that act as balanced bridge networks perform the first level of separation between
transmit and receive signals. Ten to twenty dB of isolation can be achieved with active and passive analog
compromise balance impedance networks. Digital echo cancellers can provide 30 to 40 dB of additional isolation.)
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Thus, any energy coupled into a pair used by a transmission system can not be effectively removed from
the received signal.
For the T1 line system, bipolar (AMI) encoding of the 1.544 Mb/s signal results in a transmitted in a
transmitted spectrum centered on 772 kHz. This 772 kHz signal is much higher in frequency than voice
signals and crosstalk coupling is much higher. For T-carrier or T1 system, the full duplex communication
is based on two separate twisted pairs. In the interoffice cable plant, special cables are used with a shield
between the binder groups. T1 signals going in one direction from all T1 systems are placed in single
binder groups. All signals going the other direction are in the other binder group with the shield between
them. This binder group separation of transmit and receive pairs and shielding greatly reduce, but does
not eliminate, the NEXT effects.
In the outside customer loop plant, the special cables are not readily available. Binder group separation is
practiced as much as possible for T1 in the loop plant. Shorter repeater spacing and very careful attention
to placing of repeaters relative to other T1 systems helps compensate somewhat for the much more
severe crosstalk environment in the loop plant.
For ADSL systems using FDM to separate the upstream and downstream transmissions, there is no selfNEXT to limit transmission range, as is the case for DSL and HDSL. For EC-based ADSL systems using
echo cancellation with overlapping downstream and upstream spectra, there will be self-NEXT in the
overlap region. Analyses indicate that self-NEXT will not be a limiting factor for EC-based ADSL loop
range, but compared to non-overlapped systems (e.g., FDM), overlapped systems would cause
substantially more crosstalk into ADSL upstream and downstream transmissions on other pairs.
There could be 1225 different NEXT values at a particular frequency for a 50 pair binder group, assuming
pair-to-pair NEXT is symmetrical. The measured NEXT can be approximated with a gamma or a
truncated Normal distribution on log scale. The truncated Normal distribution has a better physical
meaning since the number of NEXT pairs is limited. In practice, we might be concerned about NEXT from
more than one disturber. We need to calculate a power sum for multiple disturbers. We have 50, 3.16 ×
15
11
10 and 4.1 × 10 different power sum NEXT values for 49 disturbers, 24 disturbers, and 10 disturbers
respectively in a 50 pair binder group. The manipulation of large numbers of power sums for 24 and 10
disturbers is not easy. Hence, a direct computer simulation approach has been used in the past.
NEXT is dependent on frequency as well as on the relative location of the pairs in the binder group.
However, location is not relevant for a full binder group. Cables differ from one another with respect to
NEXT due to the cable design and manufacturing variations. The NEXT loss at any given frequency, is
usually stated as the power sum of crosstalk from signals in all other pairs of the cable binder group. The
NEXT model used for studies such as the one reported here is stated as expected 1% worst case power
sum crosstalk loss as a function of frequency. This means that on the average, 1% of the cables tested
have power sum crosstalk loss worse (less) than the model at the given frequency. Such a model is a
smooth curve Vs frequency, in which the loss decreases at about 15 dB per decade of frequency.
Individual pair-to-pair loss on a single sample of cable is not a simple curve, and individual pairs generally
exhibit different loss Vs frequency curves. The power sum loss for less than a full binder group depends
on the distribution of the pairs on which the crosstalking signal appears. Measurements for a 25-pair
binder group of a 24-AWG PIC cable are given in Figure B.7:
The study of transmission issues related to T1 systems established a first step in dealing with NEXT
modeling for simulations. The study not only tried to model NEXT loss with mean and standard deviation
but also initiated the use of 1% worst case NEXT value for overall system requirements. The reason is
-6
that people were expecting better than 95% satisfactory T1 service at an error ratio of less than 10 . The
use of the 1% worst case for transmission engineering would allow multiple spans of T1 systems in an
end-to-end service connection and also provide room for some unforeseen impairments.
The same better than 95% satisfactory service objective also applies to other digital subscriber line
systems such as DSL, HDSL, and ADSL. The 1% worst case NEXT model has also been used for DSL
and HDSL simulation studies and test procedures. The piece-wise linear (log-log scale) NEXT models
used for DSL and HDSL have loss values of 57 dB, 61 dB, and 67 dB for 49 disturbers, 10 disturbers, and
1 disturber, respectively, at a frequency of 80 kHz. A simplified 49 disturber NEXT model that has 57 dB of
loss at 80 kHz and a linear (log-log scale) slope of -15 dB/decade has been most frequently used by ANSI
T1E1.4 and can be expressed by
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
NEXT49 = x n × f 3 / 2
where x n = 8.818 × 10 −14 × (n 49 )0.6 and n is the number of disturbers. Experimental results for a 25-pair
binder group of a 24-AWG PIC cable, support this model. In Figure B.8, those results are shown fitted to
the model. The difference between the fitted results and the ANSI model can be explained as the
difference between a 50-pair binder group of 22-AWG PIC cable (ANSI model) and a 25-pair binder group
of 24-AWG PIC cable.
B.4.2 Far end crosstalk, FEXT
Far-end crosstalk (FEXT) is defined as the effect of crosstalk due to adjacent transmitters. In other words,
FEXT is due to crosstalk from adjacent transmitters at the transmitter end that couples to the receiver of
another system. See Figure B.9. FEXT loss is similar but not equal to the combination of NEXT and the
subscriber loop channel losses over the coupling length. FEXT was also considered during T1
transmission engineering efforts but was classified as a minor factor compared with NEXT. The effect of
FEXT for DSL and HDSL is very small and, hence, has been omitted in test procedures.
The effect of ADSL system self-FEXT can not be simply ignored. At high frequencies and for upstream
transmitter disturbers on short loops, ADSL self-FEXT noise power can exceed that of HDSL NEXT and
white background noise combined. A simplified FEXT model has been most frequently used by ANSI
T1E1.4 and is expressed by
2
FEXT49 = Hchannel (f ) × klf 2
Where Hchannel (f ) is the channel transfer function, k = 8 × 10 × (n 49 )0.6 , n = number of disturbers, l
= the loop length in feet, and f = frequency in Hz. Experimental results for a 25-pair binder group of a 24AWG PIC cable, support this model. In Figure B.10, those results are shown fitted to the model. The
difference between the fitted results and the ANSI model can be explained as the difference between a
50-pair binder group of 22-AWG PIC cable (ANSI model) and a 25-pair binder group of 24-AWG PIC
cable.
2
-20
The simplified FEXT model assumes the channel transfer function and length of the coupling path match
those of the disturbed system or more simply that the disturber system FEXT sources (transmitters) are
co-located with the transmitter of the disturbed system. In the upstream direction, this underestimates the
FEXT where the disturbers are closer to the central office than the victim signal transmitter.
B.4.3 Method for combining crosstalk contributions from unlike types of disturber
B.4.3.1 Base models for NEXT and FEXT
The modelling of interference contributions to an access DSL system due to crosstalk from other DSL
systems in the same cable is a fundamental part of spectral compatibility studies. The widely accepted
base models due to work by Werner and others for near end crosstalk (NEXT) and far end crosstalk
(FEXT) which are commonly used (see B.4.2) for this modelling are of the form:
Next [f , n ] = S[f ] X N f
3
2
n 0.6
Fext [f , n, l ] = S[f ] H 2 [f ] X F f 2 l n 0.6
These expressions are for that interference power likely to be exceeded in 1% or less of cases where f is
frequency, n is the number of disturbing systems, l is the length of the cable, XN and XF are scalar
constants, S[f ] is the PSD of the interfering systems and H[f ] is the pair signal transfer function. There is
an implicit assumption in these models that all the pairs involved are in the same binder group of the same
cable and have a common length and also that all the interferers are of the same type.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
0.6
There is a counter-intuitive aspect of these models relating to the n term. Intuitively it would be
expected for the interference power to be proportional to the number of disturbers (since the disturbers
0.6
are independent) but instead there is the n factor. This is due to the fact that the quantity being dealt
with is not an average value or an expectation of any sort, but a 1% worst case.
If the proximity of pairs in a cable segment is maintained along its length, certain pairs (usually the
proximate ones) contribute much more to the interference in a given pair than others do. When there are
few interferers (n small) if a single member is one of the proximate pairs the contribution to interference is
disproportionately increased. For this reason the model has to be biased for small numbers of interferers
and this is the reason for the exponent of n being less than unity.
A difficulty arises when modelling complex access network scenarios though, where there may be many
types of interferer. Suppose for example that the NEXT from n1 systems of spectrum S1[f ] and n2
systems of spectrum S2[f ] is considered. The obvious way of extending the model to cope with this is to
add the crosstalk power contributions according to the base model for each:
Next [f ] = S1[f ] X N f
3
2
n10.6 + S2 [f ] X N f
3
2
n20.6
The difficulty here is that each term in this expression is pessimistic enough for the 1% worst case, but
their joint probability is much lower, so the combined model is excessively pessimistic. This can be seen
by taking this expression and allowing S2[f ] = S1[f ] (the interferers have become of the same type). In this
case the expression can be simplified to:
Next [f ] = S1[f ] X N f
3
2
(n
0.6
1
+ n20.6
)
whereas the base model would in this case predict the lesser interference of:
Next [f ] = S1 [f ] X N f
3
2
(n1 + n 2 )0.6
This appendix describes the recommended method for calculation of NEXT and FEXT contributions from
groups of unlike disturbers. The method avoids making an over pessimistic calculation of total crosstalk
contribution which arises when assuming that all sub-groups of n interfering systems are using the worst n
pairs in a multi-pair cable. It does so without treating any sub-group differently so that there is only one
way of making the computation. The computation is such that in various limiting or trivial cases it
converges asymptotically to the base model for the reduced state. Also it never predicts a lower crosstalk
level when more disturbers are added.
The method is equally applicable to the calculation of NEXT and FEXT models.
B.4.3.2 Combining crosstalk from mixed disturber types
Instead of directly adding the crosstalk power terms, each term is first arbitrarily raised to the power 1/0.6
before carrying out the summation. Then, after the summation, the resultant expression is raised to the
power 0.6. There is no simple physical justification for this process but it has been shown both analytically
below and elsewhere by means of Monte Carlo simulations that the method has many sound and realistic
properties.
B.4.3.3 Application to two NEXT terms
Take the example from B.4.3.1. The combined NEXT power would take the form:
1
1


3
3


0.6  0.6
0.6  0.6 
2
2
Next [f ] =   S1[f ] X N f n1 
+  S2 [f ] X N f n2 







0.6
The first sound property is that if either inner term vanishes the model returns to the base model.
Suppose for example that S2≡0 or n2=0. In this case the second term would vanish. This would leave the
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two arbitrarily introduced exponents acting on a single expression, so that they cancel out, returning the
expression to the base model.
The second sound property arises when S2≡S1. In this case the common factors S1[f ] XN f
taken out of the two inner terms, and further brought outside the enclosing brackets, leaving:
Next [f ] = S1[f ] X N f
3
2
3/2
can be
0.6
1
 0.6 10.6

0
.
6
0
.
6
 n1

+ n2




( )
( )
This in turn quickly collapses to:
Next [f ] = S1 [f ] X N f
3
2
(n1 + n 2 )0.6
which is identical to the base model for the case of n1+n2 identical disturbers.
The same process can be applied to collections of more than two interference contributions.
B.4.3.4 Application to FEXT terms
The same process can also be applied to collections of FEXT interferers.
Take the case of three sources of FEXT at a given receiver. In this case there are n1 systems of
spectrum S1[f ] at range l1, a further n2 systems of spectrum S2[f ] at range l2 and yet another n3 systems
of spectrum S3[f ] at range l3.
The expected crosstalk is built in exactly the same way as before, taking the base model for each source,
raising it to power 1/0.6, adding these expressions, and raising the sum to power 0.6:
0.6
1
1


 S1[f ] H12 [f ] X F f 2 l1 n10.6 0.6 + S2 [f ] H 22 [f ] X F f 2 l 2 n20.6 0.6

Fext [f ] = 

1

2
0.6 0.6 
2
[
]
+
S
[
f
]
H
f
X
f
l
n
F
3
3
3 3


(
)
(
(
)
)
In this case it is assumed that H1[f ] is the transfer function of the length l1 etc.
Even in this more complex case the same sound properties appear.
The first sound property is that if any of the inner terms vanishes the model returns to the simpler case
until when there is only one inner term left it returns to the base model. For FEXT though there are many
more ways in which a term can disappear. Instead of just S2≡0 or n2=0 there are also the possibilities l2=0
2
and l2→∞. The latter arises because the product l2 H2 [f ]→0 as l2→∞. In any of these cases the second
term would vanish, and the equation is exactly as it would appear if the second crosstalk subgroup had
not been considered in the first place. If in addition the third term disappears, for example because n3=0,
the resulting equation is easily reduced to the base model for just the first subgroup of interferers.
The second sound property arises when for example S2≡S1 and l2=l1. This means that the first two terms
actually relate to identical system types causing FEXT at the same location. As l2=l1 it can be assumed
2
1/0.6
2
2
2
that H2 [f ]≡H1 [f ]. In this case the common factors (S1[f ] H1 [f ] XF f l1)
can be taken out of the first
two inner terms, leaving the expression:
(
) ( )
( )
(
)
1 
1
1
1



Fext [f ] =  S1[f ] H12 [f ] X F f 2 l1 0.6  n10.6 0.6 + n20.6 0.6  + S3 [f ] H32 [f ] X F f 2 l3 n30.6 0.6 








0 .6
The exponents around n1 and n2 now collapse to yield the sum n1+n2 which can then be taken back inside
the common factor to yield:
This is a draft document and thus, is still dynamic in nature.
95
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
0.6
1
1

2
0.6 0.6 
2
0.6 0.6
2
2

[
]
[
]
Fext [f ] = S1[f ] H1 f X F f l1 (n1 + n2 )
+ S3 [f ] H3 f X F f l3 n3




(
)
(
)
This is exactly the form that would be obtained if the new method were applied to the simplified modelling
situation (of an increased number of identical disturbers at the same location) in the first place.
In addition if the terms subscripted with 3 were to vanish, for example because n3=0, then the expression
would further simplify to the base model for the remaining interferers.
B.4.3.5 Crosstalk is non-decreasing
It will be apparent that the exponentiation operations, which are applied in this process, are applied to
quantities of dimension power. This means of course that they are applied to real positive functions.
After exponentiation the functions are still real and positive. As adding more disturbers is modelled by
adding together these real positive functions and then applying a monotonic mapping to the sum (the
subsequent exponentiation with exponent 0.6) it follows that adding more disturbers always increases the
crosstalk.
B.4.3.6 All disturbers are treated equally
It should be apparent from the absolute symmetry of the method that all disturbers are treated equally. It
does not matter what order the disturbers are taken in the resulting expression is the same.
B.4.3.7 Adding NEXT and FEXT
The method should be separately applied to the NEXT terms and the FEXT terms to arrive at separate
NEXT and FEXT disturbance power spectra. These power spectra should then be added.
The method should not itself be used for adding NEXT to FEXT. This is because it is perfectly feasible
for the same proximate disturbing pair to contribute both NEXT and FEXT powers from different disturbing
transceivers, whereas it cannot contribute two lots of NEXT or two lots of FEXT from different disturbing
transceivers.
96
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.1 – American wire gauge (AWG) and metric wire
Metric & AWG Wire Gauges: R & C (0Hz/DC, 20deg C or 70deg F)
mm
(0.253)
0.30 PE
0.32 PE
AWG
30
Ω/km
677
494
409
Ω/mi
1090
792
658
Ω/kft NOTES
206
150
125
0.40 PE
(0.405)
(0.455)
50
26 PIC 15.94 mil 51.6
40.0
25 MAT
80.5
83.0
64.3
280
273
213
451
439
343
84.4
83.1
64.9 Metro. Area Trans., PIC
50.9
55
50
51.6
82.0
88.5
64.4
83.0
181
179
179
172
291
288
288
276
55.1
55.6
54.5
52.3
40
45
22 PIC 25.35 mil 51.6
64.4
72.4
83.0
123
113
108
198
182
174
37.5
34.5
32.9
40
64.4
90
145
27.5
40
64.4
69
66.6
111
107
21.0
20.3 for comparison, not telephony
51.1
40
51.6
72-118
82.3 55.5
64.4 55
83.0 53.8
116-19 141
89.3
88.5
86.6
227
16.9
16.8
16.4
35.89
43
PVC, copperclad steel, parallel
22.7
36.5
66.3 12.6 PVC, copper &cadmium
67.4 12.8 for comparison, not telephony
28
mils
nF/km nF/mi
10.03 mil 40
64.4
40
64.4
12.64 mil 40
64.4
0.50 DW10
0.50 DUG
0.50 PE
(0.511) 24 PIC 20.10 mil
0.60 PE
0.63 PE
(0.644)
0.70 PE
0.80 PE
(0.812)
20
31.96 mil
0.90 DW12
0.90 PE
(0.912) 19 PIC
1)
(0.965) 18 ½
1.0 DW8
(1.024) 18
40.3 mil
41.2
41.9
Loss
DW1
28.0
45.1 63.5 102 19.4 PVC, copper & cadmium
DW3
24.4
39.3 266
428 81.0 PVC, copperclad steel
DW5
29.3
47.2 258
415 78.6 PVC, copperclad steel
DW6
27.9
44.9 200
322 61.0 PVC, copperclad steel
NOTES: ( ) = AWG conductor diameter → (mm) = not a normal metric size
PE = metric Polyethylene insulated cable
PIC = AWG Polyethylene insulated cable, sometimes called "plastic insulated cable"
as contrasted to older pulp or paper insulated cable.
PVC = Polyvinyl chloride insulated cable
DW = European drop wire, overhead/aerial
DUG = European underground drop cable
1)
F Drop Wire, AT-8668, aerial, parallel (flat, not twisted) 18 ½ AWG copperclad steel
conductors, solid black PVC insulation, oval cross section, conductor diameter = 0.038 inch,
43 ohm/kft = 227 ohm/mi, C = 0.116 µf/mi dry, C = 0.190 µf/mi wet (US Drop wire limits: 700
feet or 25 ohms)
Attenuation/Loss at 1 kHz, 10 kHz, 100 kHz, 1 MHz, 10 MHz, 30 MHz: FUTURE?
Table B. 2 - Cable model parameters for TP1 (0.4 mm or 26-gauge twisted pair)
Resistance
r0c
r0s
ac
This is a draft document and thus, is still dynamic in nature.
ax
97
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
(value)
Inductance
286.17578 Ω/km
l0
∞ Ω/km
l∞
0.14769620
b
0.0
fm
(value)
488.95186
µH/km
c0
0.92930728
806.33863 kHz
Capacitance
675.36888
µH/km
c∞
(value)
Conductance
49 nF/km
g0
0.0 nF/km
ge
0.0
(value)
43 nS/km
.70
ce
Table B.3 - Primary constants for TP1 (0.4 mm or 26-gauge twisted pair)
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
10.5e6
30.e6
Resistance
Ω/km)
(Ω
286.21516
286.3332
286.8039
290.03566
300.77488
626.85069
1.9606119e3
2.0090081e3
3.3955368e3
Inductance
(H/km)
673.7277e-6
672.26817e-6
669.55152e-6
662.28605e-6
651.94136e-6
572.86886e-6
505.33352e-6
504.66857e-6
495.20494e-6
Capacitance
(F/km)
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
49.e-9
Conductance
(S/km)
16.701192e-6
27.131166e-6
44.074709e-6
83.70424e-6
135.97794e-6
681.50407e-6
3.4156114e-3
3.5342801e-3
7.3697598e-3
98
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.4 - Cable parameters for 26-AWG PIC air core
MHz
R
G
L
C
α
β
(Ω
Ω/Km) (µ
µS/Km) (mH/Km) (nF/Km) (dB/Km) (rad/Km)
Ω)
Z0 (Ω
0.304
0.327
0.357
0.388
0.418
0.456
0.496
0.534
0.582
0.633
0.682
0.743
0.809
0.871
0.949
1.033
1.112
1.212
1.319
1.421
1.548
1.684
1.814
1.977
2.151
2.317
2.525
2.748
2.959
3.225
3.509
3.78
7.874
8.58
9.337
10.06
10.96
11.92
12.84
14
15.23
16.4
17.87
19.45
354.02
358.78
364.39
372.51
380.64
392.35
405.63
420.44
440.01
462.03
485.14
512.91
542.09
568.12
596.53
622.32
643.03
665.05
687.17
709.85
741.08
776.22
809.18
844.21
873.33
898.41
936.27
978.03
1014.30
1051.68
1094.11
1135.57
1610.79
1679.19
1747.18
1809.56
1883.19
1959.05
2030.18
2113.47
2204.55
2285.22
2381.34
2484.60
108.05
107.71
107.37
107.07
106.84
106.59
106.35
106.16
105.94
105.72
105.53
105.32
105.09
104.90
104.67
104.43
104.22
103.98
103.73
103.51
103.25
102.98
102.76
102.49
102.22
101.99
101.71
101.44
101.21
100.94
100.67
100.44
98.30
98.07
97.84
97.66
97.44
97.23
97.06
96.86
96.66
96.50
96.32
96.14
95
103
112
123
133
145
158
170
186
203
218
238
258
278
303
329
354
385
419
451
492
535
576
628
683
736
803
874
942
1026
1118
1205
2529
2758
3003
3238
3532
3846
4148
4524
4926
5313
5793
6309
0.579
0.579
0.577
0.576
0.576
0.575
0.574
0.573
0.571
0.569
0.568
0.565
0.562
0.559
0.556
0.552
0.550
0.547
0.544
0.542
0.539
0.536
0.534
0.531
0.528
0.526
0.523
0.521
0.519
0.516
0.514
0.512
0.494
0.492
0.490
0.489
0.487
0.485
0.484
0.483
0.481
0.480
0.479
0.477
49.61
49.87
50.09
50.29
50.46
50.60
50.71
50.82
50.89
50.92
50.96
50.93
50.88
50.83
50.74
50.64
50.60
50.55
50.53
50.56
50.56
50.56
50.57
50.54
50.53
50.56
50.59
50.60
50.63
50.65
50.68
50.73
51.11
51.15
51.19
51.25
51.29
51.34
51.40
51.44
51.48
51.55
51.58
51.62
14.26
14.50
14.78
15.16
15.52
16.04
16.63
17.27
18.11
19.06
20.05
21.25
22.50
23.63
24.87
26.01
26.94
27.93
28.94
29.97
31.37
32.95
34.43
36.03
37.38
38.56
40.30
42.23
43.91
45.67
47.66
49.59
72.20
75.49
78.77
81.79
85.37
89.07
92.53
96.60
101.05
105.00
109.73
114.80
10.23
11.05
12.05
13.13
14.16
15.44
16.80
18.10
19.71
21.41
23.04
25.04
27.16
29.18
31.67
34.32
36.86
40.03
43.44
46.71
50.78
55.11
59.24
64.34
69.81
75.08
81.63
88.62
95.29
103.60
112.51
121.02
248.56
270.44
293.84
316.25
344.13
373.99
402.58
438.13
476.19
512.65
557.99
606.52
This is a draft document and thus, is still dynamic in nature.
99
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.4 (concluded)
MHz
20.95
22.83
24.84
26.76
29.16
31.73
34.17
37.24
40
R
G
L
C
α
β
(Ω
Ω/Km) (µ
µS/Km) (mH/Km) (nF/Km) (dB/Km) (rad/Km)
2577.25 6803
0.476
51.69
119.36
653.04
2691.58 7419
0.475
51.72
125.00
710.86
2811.07 8079
0.474
51.76
130.90
772.74
2915.22 8711
0.473
51.82
136.07
832.08
3058.27 9499
0.472
51.85
143.12
905.83
3189.29 10343
0.470
51.89
149.65
984.79
3320.49 11153
0.470
51.94
156.16
1060.47
3469.12 12160
0.469
51.98
163.58
1154.59
3606.84 13077
0.468
52.03
170.45
1240.11
Ω)
Z0 (Ω
95.99
95.82
95.65
95.52
95.36
95.21
95.09
94.95
94.83
Table B.5– Cable parameters for 26-AWG filled PIC
MHz
0.304
0.327
0.357
0.388
0.418
0.456
0.496
0.534
0.582
0.633
0.682
0.743
0.809
0.871
0.949
1.033
1.112
1.212
1.319
1.421
1.548
1.684
1.814
1.977
2.151
2.317
2.525
2.748
R
G
L
C
α
β
Z0
(ohm/Km) (µ
µS/Km) (mH/Km) (nF/Km) (dB/Km) (rad/Km) (ohms)
397.8
48.3
0.685
46.44
14.267
10.767 121.409
398.7
52
0.682
46.77
14.376
11.618
120.79
399.5
56
0.68
47.09
14.488
12.677 120.134
400.3
60.2
0.677
47.38
14.592
13.813
119.54
401.6
64.9
0.676
47.64
14.701
14.902 119.076
403.9
69.7
0.674
47.9
14.854
16.259 118.579
407.3
75.1
0.672
48.15
15.042
17.715 118.127
413.9
80.8
0.671
48.37
15.336
19.112
117.77
423.1
86.9
0.67
48.59
15.734
20.851 117.386
437.7
93.6
0.668
48.78
16.328
22.712 117.032
454.6
101
0.667
48.95
17.005
24.487 116.751
478.8
108
0.665
49.07
17.959
26.678 116.446
506.4
117
0.663
49.14
19.045
29
116.164
533.3
125
0.661
49.18
20.1
31.197 115.937
565.9
135
0.658
49.15
21.376
33.902 115.689
595.1
145
0.654
49.08
22.528
36.772 115.457
616.4
156
0.652
49.05
23.378
39.512
115.27
635.4
168
0.649
48.98
24.153
42.923 115.064
649.9
181
0.646
48.95
24.758
46.595 114.871
665
195
0.644
48.97
25.376
50.143 114.713
688.1
209
0.643
48.99
26.309
54.578 114.539
721.9
226
0.641
49
27.65
59.311 114.374
758
243
0.639
48.99
29.074
63.801 114.239
796
261
0.637
48.93
30.583
69.341 114.089
821.3
281
0.634
48.86
31.607
75.261 113.947
840
302
0.633
48.86
32.377
80.973
113.83
871.4
326
0.632
48.85
33.646
88.12
113.699
913.4
351
0.63
48.82
35.321
95.73
113.575
100
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.5 (concluded)
MHz
2.959
3.225
3.509
3.78
4.119
4.482
4.827
5.26
5.724
6.165
6.718
7.31
7.874
8.58
9.337
10.06
10.96
11.92
12.84
14
15.23
16.4
17.87
19.45
20.95
22.83
24.84
26.76
29.16
31.73
34.17
37.24
40
R
G
L
C
α
β
Z0
(ohm/Km) (µ
µS/Km) (mH/Km) (nF/Km) (dB/Km) (rad/Km) (ohms)
948.7
377
0.628
48.81
36.73
102.987 113.472
979.9
406
0.627
48.77
38.004
112.019 113.358
1018
436
0.625
48.75
39.56
121.726 113.248
1057
471
0.624
48.75
41.111
131.005 113.158
1100
506
0.623
48.72
42.837
142.545 113.056
1153
545
0.621
48.69
44.956
154.887 112.959
1196
586
0.62
48.68
46.712
166.675 112.879
1243
630
0.619
48.66
48.622
181.378 112.789
1300
679
0.618
48.63
50.907
197.094 112.702
1347
730
0.617
48.62
52.81
212.146 112.631
1403
786
0.616
48.6
55.087
230.88 112.551
1467
846
0.614
48.57
57.697
250.927 112.473
1519
909
0.614
48.57
59.805
270.094 112.41
1581
980
0.613
48.54
62.34
293.981 112.338
1650
1054
0.612
48.52
65.163
319.541 112.269
1712
1134
0.611
48.52
67.691
344.03 112.212
1785
1222
0.61
48.5
70.701
374.474 112.148
1872
1312
0.609
48.48
74.235
407.112 112.086
1943
1415
0.608
48.48
77.176
438.268 112.035
2024
1522
0.608
48.46
80.519
477.129 111.978
2129
1637
0.607
48.43
84.794
518.684 111.923
2206
1763
0.606
48.44
87.994
558.535 111.877
2307
1894
0.605
48.42
92.156
608.048 111.826
2431
2042
0.605
48.4
97.247
661.129 111.776
2513
2196
0.604
48.4
100.693 711.891 111.736
2636
2363
0.603
48.38
105.808 774.997 111.69
2759
2545
0.603
48.36
110.912 842.743 111.646
2886
2734
0.603
48.37
116.164 907.614 111.609
2996
2947
0.602
48.35
120.82
988.173 111.568
3149
3170
0.601
48.33
127.189 1074.618 111.529
3301
3411
0.601
48.35
133.523 1157.406 111.496
3470
3673
0.6
48.33
140.573 1260.329 111.46
3671
3946
0.6
48.34
148.837 1353.774 111.429
This is a draft document and thus, is still dynamic in nature.
101
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.6 – Cable model parameters for TP2 (0.5 mm or 24-gauge twisted pair)
Resistance
r0c
r0s
ac
ac
(value)
Inductance
174.55888 Ohms/km
l0
∞ Ohms/km
l∞
0.053073481
b
0.0
fm
(value)
Capacitance
617.29539 µH/km
c∞
478.97099 µH/km
c0
1.1529766
ce
553.760 kHz
(value)
Conductance
50 nF/km
g0
0.0 nF/km
ge
0.0
(value)
234.87476 fMho/km
1.38
Table B.7 – Primary constants for TP2 (0.5 mm or 24-gauge twisted pair)
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
30.e6
Resistance
Ω/km)
(Ω
174.62121
174.8078
175.54826
180.48643
195.44702
482.06141
1.5178833e3
2.6289488e3
Inductance
(H/km)
616.69018e-6
615.95674e-6
614.35345e-6
609.15855e-6
600.41634e-6
525.43983e-6
483.72215e-6
480.34357e-6
Capacitance
(F/km)
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
50.e-9
Conductance
(S/km)
29.882364e-9
77.774343e-9
202.42201e-9
716.82799e-9
1.8656765e-6
44.754463e-6
1.0735848e-3
4.8894913e-3
102
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.8 – Cable parameters for 24-AWG PIC air core
MHz
R
G
L
C
α
β
Z0
Ω)
(Ω
Ω/Km) (µ
µS/Km) (mH/Km) (nF/Km) (dB/Km) (rad/Km) (Ω
0.304 269.87
98
0.581
51.3
11.05
10.42 106.39
0.327 280.59
105
0.578
51.22
11.51
11.19 106.24
0.357 291.94
115
0.574
51.13
12.01
12.14 105.99
0.388 302.05
125
0.57
51.07
12.46
13.16 105.67
0.418 310.67
134
0.567
51.09
12.86
14.14 105.35
0.456 321.11
146
0.563
51.12
13.34
15.36 104.96
0.496 333.53
159
0.559
51.19
13.92
16.67 104.53
0.534 346.45
172
0.556
51.29
14.51
17.92 104.16
0.582 362.84
188
0.552
51.36
15.27
19.47 103.71
0.633 378.68
204
0.548
51.41
16.01
21.12 103.27
0.682 391.83
220
0.545
51.45
16.63
22.68 102.89
0.743 405.72
240
0.54
51.48
17.29
24.63 102.46
0.809 420.66
262
0.537
51.53
18.01
26.71 102.03
0.871 436.52
282
0.534
51.62
18.76
28.72 101.69
0.949 455.25
308
0.53
51.66
19.64
31.2 101.29
1.033 472.06
335
0.526
51.69
20.45
33.85 100.92
1.112 487.37
362
0.524
51.74
21.18
36.38 100.62
1.212 507.85
394
0.521
51.78
22.15
39.54 100.28
1.319 527.94
429
0.518
51.8
23.11
42.91 99.96
1.421 545.55
463
0.515
51.84
23.95
46.14
99.7
1.548 568.5
504
0.513
51.86
25.03
50.15 99.42
1.684 590.64
549
0.51
51.87
26.09
54.44 99.16
1.814 611.79
592
0.508
51.9
27.09
58.54 98.95
1.977 636.98
645
0.506
51.9
28.28
63.65 98.72
2.151 662.83
702
0.504
51.9
29.5
69.11 98.51
2.317 686.62
756
0.502
51.92
30.62
74.34 98.35
2.525 715.83
823
0.5
51.9
32
80.84 98.17
2.748 744.89
896
0.498
51.88
33.37
87.78
98
2.959 772.18
965
0.497
51.9
34.65
94.44 97.87
3.225 804.85 1051
0.495
51.87
36.19
102.72 97.73
3.509 838.52 1143
0.494
51.84
37.77
111.57 97.61
3.78 869.26 1231
0.493
51.84
39.21
120.05 97.51
4.119 905.99 1341
0.492
51.81
40.93
130.59 97.41
4.482 944.23 1458
0.49
51.77
42.73
141.86 97.31
4.827 978.86 1570
0.489
51.77
44.35
152.67 97.24
5.26 1021.06 1709
0.488
51.72
46.33
166.1 97.16
5.724 1064.19 1859
0.487
51.68
48.35
180.48 97.1
6.165 1104.2 2002
0.487
51.67
50.22
194.25 97.05
6.718 1152.18 2179
0.486
51.63
52.47
211.37 96.99
7.31 1201.2 2369
0.485
51.58
54.77
229.7 96.95
7.874 1246.16 2551
0.484
51.57
56.88
247.25 96.91
8.58 1300.4 2778
0.484
51.52
59.43
269.08 96.88
9.337 1356.48 3020
0.483
51.47
62.06
292.44 96.85
This is a draft document and thus, is still dynamic in nature.
103
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.8 (concluded)
MHz
R
(Ω
Ω/Km)
10.06 1407.6
10.96 1469.55
11.92 1533.1
12.84 1591.69
14 1662.07
15.23 1735.64
16.4 1802.69
17.87 1886.11
19.45 1973.16
20.95 2049.06
22.83 2145.43
24.84 2251.42
26.76 2338.22
29.16 2453.81
31.73 2573.37
34.17 2678.96
37.24 2816.8
40
2935
G
(µ
µS/Km)
3252
3540
3849
4144
4512
4905
5282
5750
6251
6732
7329
7968
8581
9342
10157
10938
11909
12792
L
C
α
β
Z0
Ω)
(mH/Km) (nF/Km) (dB/Km) (rad/Km) (Ω
0.482
51.46
64.46
314.84 96.82
0.482
51.41
67.37
342.67 96.8
0.481
51.37
70.36
372.47 96.78
0.481
51.35
73.12
401.04 96.77
0.48
51.3
76.44
436.54 96.76
0.48
51.26
79.91
474.55 96.75
0.48
51.25
83.08
510.99 96.75
0.479
51.2
87.03
556.28 96.74
0.479
51.15
91.14
604.76 96.74
0.479
51.14
94.75
651.24 96.74
0.478
51.09
99.32
709.04 96.75
0.478
51.05
104.34 770.89 96.75
0.478
51.04
108.48 830.23 96.76
0.477
51
113.98 903.98 96.76
0.477
50.95
119.68 982.89 96.77
0.477
50.94
124.73 1058.57 96.78
0.477
50.9
131.31 1152.64 96.79
0.477
50.9
136.97 1238.16 96.79
104
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.9 – Cable parameters for 22-AWG PIC air core
MHz
R
G
L
C
α
β
(Ω
Ω/Km) (µ
µS/Km) (mH/Km) (nF/Km) (dB/Km) (rad/Km)
0.304 197.3
96
0.537
50.27
8.33
9.92
0.327 202.3
104
0.537
50.33
8.55
10.69
0.357 207.61
113
0.536
50.41
8.79
11.65
0.388 214
123
0.534
50.51
9.09
12.67
0.418 220.28
133
0.533
50.64
9.38
13.64
0.456 227.97
145
0.531
50.76
9.74
14.86
0.496 236.78
159
0.528
50.89
10.16
16.15
0.534 246.08
171
0.527
51.03
10.59
17.39
0.582 258.3
187
0.524
51.15
11.16
18.93
0.633 271.25
204
0.521
51.27
11.76
20.56
0.682 285.09
220
0.519
51.38
12.41
22.13
0.743 301.4
240
0.516
51.45
13.17
24.06
0.809 317.76
262
0.513
51.51
13.93
26.11
0.871 333.2
282
0.51
51.56
14.66
28.07
0.949 349.26
308
0.507
51.58
15.42
30.5
1.033 363.23
335
0.504
51.59
16.09
33.09
1.112 374.44
361
0.502
51.64
16.64
35.58
1.212 386.35
394
0.499
51.67
17.23
38.68
1.319 399.1
429
0.497
51.72
17.85
42.01
1.421 412.51
462
0.495
51.79
18.5
45.21
1.548 431.41
504
0.493
51.83
19.4
49.19
1.684 452.73
549
0.491
51.86
20.42
53.43
1.814 471.26
592
0.49
51.89
21.3
57.47
1.977 492.08
645
0.488
51.89
22.3
62.49
2.151 508.82
701
0.486
51.89
23.11
67.88
2.317 525.41
756
0.485
51.93
23.91
73.06
2.525 552.67
824
0.483
51.94
25.21
79.49
2.748 580.94
896
0.482
51.92
26.55
86.35
2.959 602.38
965
0.481
51.92
27.57
92.89
3.225 621.3
1052
0.479
51.9
28.5
101.06
3.509 650.94 1144
0.478
51.9
29.91
109.84
3.78 679.27 1233
0.477
51.9
31.25
118.21
4.119 704.13 1342
0.476
51.87
32.45
128.6
4.482 733.72 1460
0.475
51.85
33.87
139.78
4.827 765.09 1573
0.475
51.85
35.36
150.44
5.26 795.07 1713
0.474
51.82
36.8
163.73
5.724 835.86 1862
0.473
51.78
38.74
177.94
6.165 861.63 2006
0.472
51.78
39.99
191.54
6.718 900.91 2184
0.471
51.75
41.87
208.49
7.31 937.53 2375
0.471
51.71
43.63
226.63
7.874 974.62 2558
0.47
51.71
45.41
243.97
8.58 1016.13 2786
0.47
51.67
47.41
265.58
9.337 1057.87 3029
0.469
51.63
49.42
288.69
10.06 1097.92 3262
0.469
51.63
51.35
310.85
Z0
(Ω
Ω)
103.38
103.28
103.08
102.83
102.57
102.25
101.89
101.58
101.21
100.83
100.52
100.15
99.79
99.5
99.16
98.84
98.58
98.29
98.02
97.8
97.56
97.34
97.15
96.96
96.78
96.63
96.48
96.33
96.22
96.1
95.99
95.9
95.81
95.73
95.67
95.6
95.54
95.5
95.45
95.41
95.37
95.34
95.31
95.29
This is a draft document and thus, is still dynamic in nature.
105
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.9(concluded)
MHz
R
(Ω
Ω/Km)
10.96 1145.72
11.92 1191.83
12.84 1236.98
14 1290.75
15.23 1349.41
16.4 1403.08
17.87 1464.42
19.45 1534.46
20.95 1591.96
22.83 1668.08
24.84 1737.61
26.76 1808.44
29.16 1889.94
31.73 1994.92
34.17 2070.38
37.24 2175.93
40 2278.32
G
(µ
µS/Km)
3552
3862
4159
4529
4925
5304
5775
6280
6763
7364
8008
8624
9391
10212
10999
11977
12866
L
C
α
β
(mH/Km) (nF/Km) (dB/Km) (rad/Km)
0.468
51.59
53.66
338.4
0.468
51.55
55.9
367.9
0.468
51.54
58.08
396.17
0.467
51.51
60.69
431.33
0.467
51.47
63.54
468.97
0.467
51.46
66.14
505.05
0.466
51.42
69.14
549.89
0.466
51.38
72.54
597.91
0.466
51.38
75.36
643.91
0.465
51.34
79.08
701.15
0.465
51.3
82.51
762.44
0.465
51.3
85.99
821.18
0.465
51.26
90.02
894.26
0.465
51.23
95.14
972.48
0.465
51.23
98.9
1047.44
0.464
51.19 104.11 1140.68
0.464
51.19 109.13 1225.38
Z0
(Ω
Ω)
95.27
95.26
95.24
95.23
95.23
95.22
95.22
95.21
95.21
95.21
95.22
95.22
95.22
95.23
95.23
95.24
95.25
Table B.10 – Cable model parameters for TP3 (DW10 reinforced .5 mm copper PVC-insulated
steel strength member, polyethelene sheath)
Resistance
r0c
r0s
ac
ax
(value)
Inductance
180.93 Ohms/km
l0
∞ Ohms/km
l∞
.0497223
b
0
fm
(value)
Capacitance
728.87 µH/km
c∞
543.43 µH/km
c0
.75577086
ce
718888 Hz.
(value)
Conductance
51 nF/km
g0
63.8 nF/km
ge
.11584622
(value)
89 nMho/km
.856
106
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.11 – Primary constants for TP3 (DW10 reinforced .5 mm copper PVC-insulated steel
strength member, polyethelene sheath)
Frequency
(Hz)
5000
10000
20000
50000
100000
1.5e6
10.e6
30.e6
Resistance
Ω/km)
(Ω
180.98245
181.13951
181.76372
185.96294
199.01927
579.72026
1.493348e3
2.5864318e3
Inductance
(H/km)
724.62777e-6
721.81902e-6
717.26896e-6
707.04768e-6
694.78496e-6
611.02577e-6
565.7413e-6
553.86667e-6
Capacitance
(F/km)
74.722723e-9
72.886768e-9
71.192474e-9
69.151688e-9
67.745589e-9
63.21713e-9
60.79255e-9
59.613734e-9
Conductance
(S/km)
130.65969e-6
236.50605e-6
428.0977e-6
938.00385e-6
1.6978732e-3
17.246997e-3
87.504681e-3
224.11821e-3
Table B.12 – Cable model parameters for FP (1.14 mm flat cable)
Resistance
r0c
r0s
ac
ax
(value)
Inductance
41.16 Ohms/km
l0
∞ Ohms/km
.001218
b
0
fm
(value)
Capacitance
1000 µH/km
c∞
911 µH/km
c0
1.195
ce
174.2 kHz
(value)
Conductance
22.68 nF/km
g0
31.78 nF/km
ge
.1109
(value)
53 nMho/km
.88
l∞
Table B.13 – Primary constants for FP (1.14 mm flat cable)
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
30.e6
Resistance
Ω/km)
(Ω
41.268736
41.589888
42.805363
49.316246
62.284991
186.92411
590.76171
1.023223e3
Inductance
(H/km)
998.73982e-6
997.16583e-6
993.76481e-6
983.62766e-6
969.66713e-6
920.40732e-6
911.20963e-6
910.69563e-6
Capacitance
(F/km)
35.041871e-9
34.127572e-9
33.280903e-9
32.257008e-9
31.548702e-9
29.550852e-9
28.003118e-9
27.392833e-9
Conductance
(S/km)
95.360709e-6
175.49949e-6
322.98493e-6
723.38496e-6
1.3312998e-3
10.098942e-3
76.608308e-3
201.43854e-3
This is a draft document and thus, is still dynamic in nature.
107
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.14 – Cable model parameters for category 5 twisted pair
Resistance
r0c
r0s
ac
ax
(value)
Inductance
176.6 Ohms/km
l0
∞ Ohms/km
l∞
.0500079494
b
0.0
fm
(value)
Capacitance
1090.8 µH/km
c∞
504.5 µH/km
c0
0.705
ce
32570 kHz
(value)
Conductance
48.55 nF/km
g0
0.0 nF/km
0.0
(value)
1.47653 nS/km
.91
ge
Table B.15 – Primary constants for category 5 twisted pair
Frequency
(Hz)
5000
10000
20000
50000
100000
1.e6
10.e6
30.e6
Resistance
Ω/km)
(Ω
176.656720
176.826554
177.501041
182.020084
195.898798
475.172462
1.4954809e3
2.5901370e3
Inductance
(H/km)
967.308142e-6
913.078780e-6
847.551900e-6
753.691218e-6
687.417012e-6
552.634084e-6
514.663928e-6
509.228994e-6
Capacitance
(F/km)
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
48.55e-9
Conductance
(S/km)
3.430086e-6
6.445287e-6
12.110988e-6
27.880784e-6
52.389261e-6
425.835904e-6
3.461324e-3
9.406382e-3
108
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.16 – Cable parameters, two-pair twisted drop
MHz
R
(Ω
Ω//Kft)
L
(mH/Kft)
C
(nF/Kft)
α
(dB/Kft)
(ohm)
0.772
0.819094
0.869062
0.922077
0.978327
1.038008
1.101329
1.168514
1.239797
1.315428
1.395673
1.480813
1.571148
1.666992
1.768684
1.876579
1.991056
2.112517
2.241387
2.378118
2.523191
2.677113
2.840425
3.0137
3.197545
3.392605
3.599564
3.819149
4.052129
4.299321
4.561593
4.839864
5.13511
5.448368
5.780735
6.133378
6.507533
6.904512
7.325709
7.772599
8.246752
8.749829
112.6412
116.4074
120.112
124.0752
128.3016
132.5838
137.0843
142.5063
147.1853
151.9907
157.6941
163.022
168.7727
174.1886
180.6494
187.1776
194.3419
201.1778
209.5537
216.4363
223.9826
232.2693
240.6053
250.1631
259.1278
268.8572
279.5037
290.5045
300.6599
311.7282
323.8352
336.0791
349.7153
363.3367
377.6032
393.3168
407.97
426.5291
443.104
461.1752
479.3825
499.3066
0.143576
0.141203
0.140489
0.140184
0.139989
0.139838
0.139706
0.139581
0.139459
0.139336
0.13921
0.137695
0.138949
0.138813
0.138676
0.138536
0.138396
0.138254
0.138115
0.137974
0.137835
0.137699
0.137566
0.137436
0.137309
0.137186
0.137066
0.13695
0.136837
0.136727
0.13662
0.136515
0.136413
0.136312
0.136212
0.136114
0.135994
0.135918
0.135701
0.135721
0.135621
0.13552
14.15747
13.94644
13.89817
13.88958
13.89123
13.89671
13.90334
13.91021
13.91676
13.92271
13.92782
13.79328
13.93559
13.93821
13.94017
13.94142
13.94219
13.94235
13.94233
13.94175
13.94097
13.94009
13.93908
13.9381
13.93699
13.93593
13.93491
13.9339
13.93283
13.93179
13.93077
13.92969
13.92858
13.92734
13.92597
13.92445
13.92044
13.92074
13.90626
13.91594
13.91307
13.90983
4.85443771
5.02088326
5.18481854
5.36006449
5.54683894
5.73618274
5.93512422
6.17415048
6.38116738
6.59381465
6.84559689
7.0812544
7.3354461
7.57525379
7.86066953
8.14921318
8.46564642
8.7679671
9.13761936
9.44235495
9.77621976
10.142589
10.5113113
10.9336262
11.3302283
11.7604684
12.2310533
12.7173706
13.1668983
13.6566036
14.1920396
14.7337063
15.3366472
15.939193
16.5702788
17.2651257
17.9136803
18.7340146
19.4674802
20.266952
21.072676
21.9541433
100.7043
100.6214
100.5408
100.4626
100.3867
100.313
100.2415
100.172
100.1046
100.0391
99.97558
99.91388
99.85398
99.79583
99.73937
99.68456
99.63135
99.5797
99.52955
99.48086
99.43359
99.3877
99.34315
99.2999
99.25792
99.21715
99.17758
99.13916
99.10186
99.06565
99.0305
98.99637
98.96324
98.93107
98.89984
98.86953
98.84009
98.81152
98.78378
98.75685
98.7307
98.70532
Z0
This is a draft document and thus, is still dynamic in nature.
109
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.16 (concluded)
MHz
R
(Ω
Ω//Kft)
L
(mH/Kft)
C
(nF/Kft)
α
(dB/Kft)
(ohm)
9.283595
9.849923
10.4508
11.08833
11.76475
12.48244
13.2439
14.05182
14.90903
15.81852
16.7835
17.80735
18.89365
20.04622
21.2691
22.56658
23.94321
25.40382
26.95353
28.59778
30.34234
32.19331
34.1572
36.2409
38.4517
40
518.4216
539.9883
562.2894
586.0891
610.2553
635.0644
662.0816
691.5593
720.1421
749.6236
783.4812
817.8867
851.7503
890.3597
925.5264
969.5871
1010.435
1060.989
1104.402
1152.924
1191.673
1244.775
1319.738
1367.755
1441.865
1488.43
0.135417
0.135312
0.135205
0.135096
0.134984
0.13487
0.134752
0.134631
0.134508
0.134382
0.134252
0.13412
0.133985
0.133846
0.133706
0.133561
0.133416
0.133265
0.133115
0.132962
0.132811
0.132654
0.132489
0.132332
0.132167
0.132058
13.90623
13.9022
13.89775
13.89286
13.88752
13.88174
13.87545
13.86865
13.86144
13.85377
13.84553
13.83685
13.82779
13.81817
13.80831
13.79778
13.78702
13.77562
13.76416
13.75225
13.74044
13.72787
13.71443
13.70164
13.68794
13.67882
22.8003059
23.7545736
24.7414418
25.7945582
26.8641089
27.962258
29.1579437
30.462329
31.7276262
33.0328382
34.5312289
36.0541353
37.5534985
39.2624684
40.819977
42.7701295
44.5789511
46.816403
48.7392081
50.8878273
52.6053964
54.9569524
58.2741802
60.4020976
63.682754
65.7446239
98.68068
98.65676
98.63353
98.61098
98.58909
98.56784
98.54721
98.52718
98.50774
98.48886
98.47053
98.45274
98.43547
98.4187
98.40242
98.38661
98.37127
98.35637
98.34191
98.32787
98.31424
98.30101
98.28816
98.27569
98.26358
98.25571
Z0
110
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.17 – Cable parameters, two-pair quad drop
MHz
0.772
0.82
0.87
0.923
0.979
1.038
1.103
1.169
1.241
1.317
1.396
1.483
1.572
1.668
1.771
1.877
1.993
2.114
2.243
2.381
2.523
2.68
2.843
3.016
3.201
3.393
3.604
3.822
4.055
4.304
4.562
4.845
5.139
5.453
5.788
6.133
6.515
6.91
7.331
7.782
8.247
8.76
9.291
R
(Ω
Ω//Kft)
Pair 1
125.4
130.1
135
139.5
143.4
148.3
153
158.1
163.8
170.3
176.3
180.7
187.2
193.5
200.4
207.1
214
221.6
229.6
237.2
245.6
254.9
263.5
273.1
283.3
293.1
303.9
314.7
326.9
338.9
351.8
364.6
377.6
391.6
407
424.1
439.6
456.8
474.7
494.7
513.7
533.8
553.8
Pair 2
129.2
134.7
139.9
145.4
149
154.1
158.3
164.2
170.3
176.6
182.6
188.3
194.3
202.3
209.2
215.6
223
232.1
239.2
248.3
257.7
267.4
276.3
287.4
297.2
307.7
318.5
331.8
343.3
355.5
370.7
384.7
400.6
413.7
429.1
445.9
463.9
482.3
501.6
520.4
542.6
565.7
594.5
L
(mH/Kft)
Pair 1
0.156
0.14
0.145
0.146
0.146
0.146
0.146
0.142
0.146
0.146
0.146
0.146
0.146
0.146
0.145
0.145
0.145
0.145
0.145
0.145
0.145
0.145
0.145
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.144
0.143
0.143
0.143
0.142
0.143
0.143
0.143
0.143
Pair 2
0.178
0.141
0.148
0.149
0.15
0.151
0.151
0.151
0.148
0.153
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.15
0.149
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
C
(nF/Kft)
Pair 1
12.53
11.28
11.67
11.75
11.79
11.81
11.82
11.5
11.84
11.84
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.85
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.84
11.77
11.79
11.83
11.83
11.83
Pair 2
13.69
10.83
11.38
11.51
11.57
11.6
11.62
11.63
11.43
11.75
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.64
11.65
11.59
11.46
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
α
(dB/Kft)
Pair 1
4.879
5.065
5.259
5.439
5.592
5.786
5.971
6.174
6.402
6.658
6.896
7.07
7.33
7.579
7.851
8.118
8.39
8.691
9.01
9.312
9.642
10.01
10.35
10.74
11.14
11.53
11.96
12.39
12.87
13.34
13.85
14.36
14.88
15.43
16.04
16.72
17.34
18.02
18.73
19.52
20.27
21.07
21.87
Pair 2
4.923
5.132
5.331
5.54
5.676
5.87
6.031
6.256
6.487
6.729
6.957
7.175
7.404
7.71
7.969
8.214
8.497
8.843
9.116
9.46
9.819
10.19
10.53
10.95
11.32
11.72
12.13
12.64
13.08
13.54
14.12
14.66
15.26
15.76
16.35
16.99
17.68
18.38
19.11
19.83
20.67
21.55
22.65
This is a draft document and thus, is still dynamic in nature.
Z0
(ohms)
Pair 1 Pair 2
111.5 113.9
111.5 113.9
111.4 113.9
111.3 113.9
111.3 113.9
111.2 113.9
111.2 113.9
111.1 113.9
111.1 113.9
111
113.9
111
113.9
110.9 113.9
110.9 113.9
110.8 113.9
110.8 113.9
110.7 113.9
110.7 113.9
110.6 113.9
110.6 113.9
110.6 113.9
110.5 113.9
110.5 113.9
110.5 113.9
110.4 113.9
110.4 113.9
110.4 113.9
110.3 113.9
110.3 113.9
110.3 113.9
110.2 113.9
110.2 113.9
110.2 113.9
110.2 113.9
110.1 113.9
110.1 113.9
110.1 113.9
110.1 113.9
110
113.9
110
113.9
110
113.9
110
113.9
109.9 113.9
109.9 113.9
111
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.17 (concluded)
MHz
9.857
10.46
11.09
11.78
12.49
13.25
14.07
14.91
15.84
16.8
17.82
18.92
20.05
21.29
22.58
23.96
25.43
26.95
28.63
30.37
32.22
34.2
36.24
38.5
40
R
(Ω
Ω//Kft)
Pair 1
578.8
596.9
620.5
646.7
672.1
701.4
728.4
757.8
789.1
821.2
854.9
892.9
925.2
964.6
1012
1053
1104
1142
1194
1229
1306
1352
1395
1482
1530
Pair 2
610.9
635.1
661.2
685.5
712.6
740.6
772.8
804.1
837.8
872.1
906.3
945.9
991.2
1028
1078
1120
1169
1226
1283
1319
1378
1439
1507
1555
1622
L
(mH/Kft)
Pair 1
0.143
0.143
0.143
0.143
0.143
0.143
0.143
0.143
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
0.142
Pair 2
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
0.151
C
(nF/Kft)
Pair 1
11.83
11.83
11.83
11.83
11.83
11.83
11.82
11.82
11.82
11.82
11.82
11.82
11.82
11.82
11.81
11.81
11.81
11.81
11.81
11.81
11.81
11.8
11.8
11.8
11.8
Pair 2
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
11.63
α
(dB/Kft)
Pair 1
22.85
23.57
24.51
25.55
26.55
27.72
28.79
29.95
31.2
32.47
33.81
35.32
36.6
38.16
40.03
41.66
43.69
45.19
47.26
48.65
51.71
53.53
55.26
58.69
60.6
Pair 2
23.28
24.2
25.19
26.12
27.15
28.22
29.44
30.64
31.92
33.23
34.53
36.04
37.77
39.18
41.08
42.67
44.56
46.71
48.9
50.26
52.49
54.83
57.4
59.24
61.81
112
This is a draft document and thus, is still dynamic in nature.
Z0
(ohms)
Pair 1 Pair 2
109.9 113.9
109.9 113.9
109.9 113.9
109.9 113.9
109.8 113.9
109.8 113.9
109.8 113.9
109.8 113.9
109.8 113.9
109.8 113.9
109.7 113.9
109.7 113.9
109.7 113.9
109.7 113.9
109.7 113.9
109.7 113.9
109.7 113.9
109.7 113.9
109.6 113.9
109.6 113.9
109.6 113.9
109.6 113.9
109.6 113.9
109.6 113.9
109.6 113.9
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.18 – Cable parameters, flat-pair Drop
MHz
0.772
0.82
0.87
0.923
0.979
1.038
1.103
1.169
1.241
1.317
1.396
1.483
1.572
1.668
1.771
1.877
1.993
2.114
2.243
2.381
2.523
2.68
2.843
3.016
3.201
3.393
3.604
3.822
4.055
4.304
4.562
4.845
5.139
5.453
5.788
6.133
6.515
6.91
7.331
7.782
8.247
8.76
9.291
9.857
R
(Ω
Ω//Kft)
170.5
173.7
177.6
182.4
188
194.5
201.8
209.9
219
228.9
239.8
251.5
264.2
277.8
292.3
307.8
324.2
341.6
359.9
379.1
399.4
420.6
442.8
466
490.1
515.3
541.4
568.5
596.6
625.7
655.8
686.9
719.1
752.2
786.3
821.4
857.5
894.7
932.8
972
1012
1053
1095
1139
L
C
α
(mH/Kft) (nF/Kft) (dB/Kft)
0.129
0.13
0.131
0.132
0.133
0.135
0.136
0.138
0.139
0.141
0.142
0.144
0.145
0.147
0.148
0.149
0.151
0.152
0.153
0.155
0.156
0.157
0.158
0.159
0.16
0.161
0.162
0.163
0.164
0.165
0.166
0.167
0.168
0.169
0.169
0.17
0.171
0.171
0.172
0.173
0.173
0.174
0.175
0.175
10.5
10.43
10.39
10.36
10.34
10.33
10.33
10.32
10.33
10.34
10.35
10.36
10.38
10.39
10.41
10.42
10.43
10.45
10.46
10.47
10.49
10.49
10.5
10.51
10.52
10.52
10.53
10.54
10.54
10.54
10.55
10.55
10.55
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
10.56
6.681
6.76
6.87
7.012
7.185
7.389
7.625
7.892
8.19
8.519
8.88
9.273
9.696
10.15
10.64
11.16
11.7
12.28
12.9
13.54
14.21
14.92
15.66
16.43
17.23
18.06
18.92
19.81
20.74
21.69
22.68
23.7
24.75
25.83
26.95
28.09
29.27
30.47
31.71
32.98
34.28
35.62
36.98
38.38
Z0
(ohms)
110.8
111.5
112.2
112.9
113.6
114.2
114.8
115.5
116.1
116.6
117.2
117.7
118.3
118.8
119.3
119.7
120.2
120.7
121.1
121.5
121.9
122.3
122.7
123.1
123.5
123.8
124.2
124.5
124.9
125.2
125.5
125.8
126.1
126.4
126.6
126.9
127.2
127.4
127.7
127.9
128.1
128.3
128.6
128.8
This is a draft document and thus, is still dynamic in nature.
113
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Table B.18 (concluded)
MHz
R
(Ω
Ω//Kft)
10.46
11.09
11.78
12.49
13.25
14.07
14.91
15.84
16.8
17.82
18.92
20.05
21.29
22.58
23.96
25.43
26.95
28.63
30.37
32.22
34.2
36.24
38.5
40
1183
1228
1274
1322
1370
1419
1469
1521
1573
1626
1680
1736
1792
1849
1908
1967
2027
2089
2151
2214
2279
2344
2410
2455
L
C
α
(mH/Kft) (nF/Kft) (dB/Kft)
0.176
0.176
0.177
0.177
0.178
0.178
0.178
0.179
0.179
0.18
0.18
0.181
0.181
0.181
0.182
0.182
0.182
0.182
0.183
0.183
0.183
0.184
0.184
0.184
10.56
10.56
10.56
10.55
10.55
10.55
10.54
10.55
10.55
10.54
10.55
10.54
10.54
10.54
10.53
10.53
10.53
10.53
10.53
10.52
10.51
10.52
10.51
10.5
39.8
41.26
42.75
44.27
45.82
47.41
49.02
50.67
52.34
54.05
55.79
57.57
59.37
61.2
63.07
64.96
66.89
68.85
70.84
72.86
74.92
77
79.12
80.55
Z0
(ohms)
129
129.2
129.4
129.6
129.7
129.9
130.1
130.2
130.4
130.6
130.7
130.9
131
131.1
131.3
131.4
131.5
131.7
131.8
131.9
132
132.1
132.2
132.3
114
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Z in =
AZ L + B
CZL + D
Zout =
DZS + B
CZS + A
Vin = VS
Z in
ZS + Z in
Vout = VL = VS
Pin =
ZL
AZ L + B + CZS ZL + DZS
 1 
1
| Vin |2 Re 

2
 Z in 
Pout =
 1 
1
| Vout |2 Re  
2
 ZL 
Vout
ZL
=
Vin
AZ L + B
Loop Insertion Loss =
Z L + ZS
AZ L + B + CZS ZL + DZS
Mean Squared Loss (MSL) =
1 N Pout (fi )
∑
N i =1 Pin (fi )
Figure B.1 – Loop ABCD parameters, impedance and voltages
This is a draft document and thus, is still dynamic in nature.
115
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
I1
ZS
Vs
I2
+
V1
–
+
–
A B 
C D 


Source
+
V2=VL
–
ZL
Load
Two-port Network
Figure B. 2 – Two-port network model.
I(x)
I(x+dx)
+
V(x)
–
Rdx
Ldx
Cdx
Gdx
+
V(x+dx)
–
Z = R + jωL
Y = G + jωC
Two-port Network
Figure B.3 – Incremental section of twisted-pair transmission line.
ZS = RS + jX S
+
–
VS
+
V
I
ZL = RL + jX L
–
Figure B.4 – Simple load circuit for power analysis.
116
This is a draft document and thus, is still dynamic in nature.
T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Source
ZS
VS
+
Line Length d
Z0 , γ
+
–
VL
ZL
–
1 ZS 
Φ0 = 

0 1 
Load
 cosh(γd )
Z0 ⋅ sinh(γd )

Φ1 =  1 ⋅
sinh(γd )
cosh( γd ) 
 Z0

Source
ZS
VS
+
–
Line Length d2
Z02 , γ2
+
VL
Line Length d3
Z03 , γ3
Line Length d1
Z01 , γ1
1 ZS 
Φ0 = 

0 1 
 cosh(γ1d1)
Z01 ⋅ sinh(γ1d1)

Φ1 =  1 ⋅
sinh(γ1d1)
cosh(γ1d1) 
 Z01


1
0
Φ 2 =  1 ⋅ tanh(γ d ) 1


2 2

 Z02
ZL
–
 cosh(γ 3d3 )
Z01 ⋅ sinh(γ 3d3 )

Φ 3 =  1 ⋅
sinh(γ 3d3 )
cosh(γ 3d3 ) 
 Z03

Load
Figure B.5 – Examples of two-port cascades for twisted-pair transmission line configurations
Same Binder Group
Transmit
NEXT
Receive
Figure B.6 – Near end crosstalk (NEXT)
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
P air1
P air2
P air3
P air4
P air5
P air6
P air7
P air8
P air9
P air10
P air11
P air12
P air13
P air14
P air15
P air16
P air17
P air18
P air19
P air20
P air21
P air22
P air23
P air24
P air25
1% C ase
N EXT PO W ER SU M LO SS(dB )
1000 FT,24 AW G PIC
70
60
50
40
30
20
10
0
0.1
1
10
100
FR EQ U EN CY(M H z)
Figure B.7 – NEXT power sum losses for 25 pairs of PIC cable binder group
1% N E X T P O W E R S U M LO S S
1000 FT,24 A W G P IC
70
60
.3 - 40M H z fit
50
AN SI
40
1.5 - 30M H z fit
30
20
10
0
0.1
1
10
100
FR E Q U E N C Y (M H z)
Figure B.8 – Comparison of ANSI NEXT with Measured NEXT
118
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Same Binder Group
Transmit
FEXT
Receive
Figure B.9 – Far end crosstalk (FEXT)
1% FE X T P O W E R S U M LO S S
1000 FT,24 A W G P IC
70
60
50
40
.3 -40M H z fit
30
AN SI
20
1.5 -30M H z fit
10
0
0.1
1
10
100
FR E Q U E N C Y (M H z)
Figure B.10 – Comparison of ANSI FEXT with Measured FEXT
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex C: Probability of error estimation
(Informative)
As in all digital transmission, the most common measure of DSL performance is the probability of error.
Usually, probability of bit error is desired, but sometimes probability of symbol error is also of interest. In
either case, the engineer attempts to measure the probability of error by observation of a system’s
performance – this is usually achieved with transmission hardware through the use of a BERT. Typically,
the bert allows a test engineer to select one of a number of different bit streams (typically various lengths
of pseudorandom patterns). The bert essentially synchronizes to the receiver output bit pattern and
compares it to the input pattern, while counting the number of error positions. The number of errors
accumulated is periodically divided by the total number of bits measured to estimate the probability of bit
error. As time increases, this average bit error rate should converge to the actual system value (when the
system is not time-varying). A reset button allows bit error counts to be restarted at zero when necessary.
In general, bit error rate measurements become more reliable with time. The designer then needs to
know how long the bit error rate needs to be observed before any derived bit error rate is sufficiently
accurate. This is a basic statistical problem that involves measurement of a distribution. Let us suppose
that bit (or symbol) errors are made with some unknown, but fixed, probability p. One measures p by
th
counting errors in successive observations of the channel output. Let the k experiment be denoted by
pk where
error measured ( p )
1
.
pk = 
0 no error measured (1 − p )
Then, an estimate of the probability of error, based on N independent measurements, is
pˆ (N ) =
1 N −1
⋅ ∑ pk .
N k =0
This estimate has an average value
E [pˆ (N )] = p
and a variance about this average of
σˆ 2p = var [pˆ (N )] =
p
(1 − p ) ≈ p .
N
N
Clearly, this estimate converges to the true probability of error as N gets large. However, N can be much
larger than sometimes expected. For instance, the standard deviation is the square root of the variance.
-7
Thus, for a system where p=10 , for the probability of error estimate deviation from to have a standard
deviation of 10% of the value of p , then 10 −8 =
10 −7
N
or
N = 10 9 .
In fact, a single standard deviation may not be sufficient to guarantee good accuracy of measured
probability of error.
The distribution of the random variable pˆ (N ) has a binomial distribution given by
N 
f pˆ (k ) = Pr {N ⋅ pˆ (N ) = k } =  (1 − p )N −k p k .
k 
The test engineer desires to ensure that the probability of error estimate deviates less than an amount
p
ε=
from the true value with a high degree of confidence. Let us say that we desire (1 − δ ) (equal say
L
90%) confidence that the measurement deviates less than ε from the true value. Corresponding to this
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
value of ε is a range of values for the index k (ε ) such that the estimate is close enough, mathematically
stated precisely as
{
}
N 
Pr pˆ (N ) − p < ε > 1 − δ= ∑  (1 − p )N −k p k .
k (ε )  k 
Clearly, just to have a non-trivial set for k (ε ) , then N ≥ L / p . Evaluation of the sum can be excessively
intensive and so a rough use of the central limit theorem is applied to the distribution to say that for large
N , the distribution is approximately Gaussian and so The probability is then approximated by
{
}
Pr pˆ (N ) − p < ε ≈
p
⌠L
−p
⌡
L
1
2π
−
e
x2
2p L
 Np 
 = 1− δ ,
dx = 1 − 2Q
 L 


or then
 Np  δ
= .
Q
 L  2


For 90% confidence, δ = .1 that the error is less than p/L%, then the above equation produces
N≥
14.8 ⋅ L2
,
p
so, for instance, 10% accuracy at p = 1 e −7 , requires that nearly 3 billion bits be tested. Thus, at a speed
of 10 Mbps, this takes about 300 seconds, or approximately 5 minutes. For 1 Mbps transmission, the test
would require 3.5 hours. The measurement time can be reduced most easily by reducing L to 2, which
corresponds to only about a .2 dB SNR difference. Even then, 1 Mbps DSL transmission at 1e-7 error
rate may take 2 minutes for a measurement, while a lower speed of 100 kbps would take 20 minutes.
Such measurement intervals are typical in, for instance, performance comparision tests sponsored by
standards groups like ANSI.
C.1
Effect of input bit sequence
Clearly, the input bit sequence will need to be periodic for any practical implementation of a bert. The
period of this sequence should be such that it exceeds the memory of the transmission system
significantly. Such sequence length is necessary to ensure that all possible channel output conditions are
excited. Given that DSL transmission systems may have long memory, a 24-bit pseudorandom pattern is
24
most likely used (with a period of 2 -1 bits and running through all 24-bit sequences once and only once
per period. Some sequences with lengths greater than 24 will not have equal likelihood of occurrence and
can bias probability of error measurements, but this effect is usually presumed small by DSL engineers.
C.2
Period of injected “Gaussian” noise
ANSI T1E1.4 studies note that most commercial line simulators make use of pseudorandom noise in
generating Gaussian noise measuring DSL performance. An unfortunate consequence is that the peak
noise samples generated do not accurately follow the Gaussian distribution tails, thus biasing probability of
error measurements in an optimistic direction. Typically, line simulators generate noise by using some
internal analog noise source and adding digitally generated noise to it. If the period of the latter digital
“Gaussian” noise is M, then the peak value of the noise in a set of M samples is also Gaussian with mean
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
 1 
µ = 1 − Q

 Mσ 
and variance
µ2
σ 2peak
2
(
M − 1) ⋅ 2πσ 2 ⋅ e σ
=
M3
.
7
To eliminate an optimistic bias, the tester would need M>10 , which complicates line simulator design.
For the more typical value of M=8192, the bias is optimistic by 2.4 dB (see also [17]), meaning that lab
measurements for M = 8192 are then optimistic by 2.4 dB and should be reduced by such for field
performance.
C.3
dB margin and importance sampling
To avoid long measurement times, importance sampling is a method used by test engineers to test only
the worst-case situations by increasing the occurrence of peak noise samples with respect to Gaussian
noise. Such importance sampling must be very carefully applied for informative results. However, DSL
engineers use a form of importance sampling in the concept of margin. Recalling that DSLs are specified
-7
to have a probability of bit error of 10 with a 6 dB margin. This means that the actual probability of error
-24
would be below 10 , requiring centuries of measurement time. Instead, testing is executed with noise
increased by 6 dB so that reasonable measurement times can be used. The margin concept is one
mechanism for importance sampling.
DSL engineers, however, prefer the supposed practical
interpretation that unforeseen noise disturbances of a temporarily nature will not cause an error with such
a large margin, although justification for such unforeseen noises at a level of 6 dB is difficult (either the
noise change is much smaller for crosstalk changes or much larger for impulse or temporary RF
disturbances).
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Annex D: Additional spectrum management topics currently
under study by the formulating committee of this standard
(informative)
The formulating committee of this standard has considered several additional topics for which specific
requirements or recommendations could not be finalized in the short time available for the development of
the first edition of this standard. These topics include, but are not limited to, the following items which the
formulating committee feels are important and should be addressed in future editions of this standard:
-
Spectrum management guidelines associated with remotely deployed TU-C equipment, such as
ADSL ATU-C implementations that are collocated with a digital loop carrier remote terminal some
distance from, the Central Office.
-
Spectrum management guidelines associated with repeatered DSL applications such as mid-span
ISDN or HDSL repeater implementations.
-
Revision of non-DSL out-of-band metallic and longitudinal signal power limits to provide an
adequate level of protection for DSL systems.
-
Addition of VDSL to the basis systems list. When VDSL is standardized, it is expected to be added
to the basis systems list along with information for the analytical method. When it is standardized, it
is to be spectrally compatible.
-
Possible extension of the Spectrum Management Class 5 upstream band to lower frequencies.
-
Methods for optimizing PSDs, maximizing throughput and binder group capacity.
-
Trade-offs between loop length guidelines and spectral characteristics.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex E: Time varying, user data-dependent crosstalk from
T1 and DDS services
(informative)
Both T1 AMI and 56 kbps DDS are well established and growing services in North America. Thus, it is
important that the effects of these services on other services are properly considered. Specifications for
T1 AMI can be found in ANSI T1.403-1992. Specifications for DDS can be found in ANSI T1.410-1992
and AT&T Technical Reference PUB 54075.
Neither T1 nor DDS include scrambling of user data. As a consequence, both the spectral behavior and
the time domain behavior of T1 and DDS line signals are highly dependent on the user data being
transmitted at any moment. This behavior thus manifests itself in the crosstalk interference of T1 and DDS
into other services. Such behavior strongly contrasts with traditional stationary crosstalk models used to
analyze and test subscriber loop technologies.
T1 and DDS services host many types of user data and protocols. A consequence is that user data
content and data patterns cannot be predicted nor controlled and it should be assumed that any pattern
can be transmitted, that the duration of a pattern is indeterminable and that changes from one pattern to
another can occur at any moment. Examples include bursts of “random” user data followed by idle periods
consisting of HDLC flags or ONEs.
One consequence of data dependency is that the transmit power spectral density (PSD) and the signal
energy in a given frequency band can vary greatly as user data patterns change. The time duration of
each PSD variant is caused solely by the time changes of user data content, and thus the time duration of
each PSD variant may vary from less than a millisecond to many hours. Changes from one PSD to any
other may occur at any moment.
An option for scrambling is defined in T1.410. Currently, however, it is not widely used.
Figure E. 1 and Figure E. 2 show examples of stationary PSD variants for T1 and DDS. Figure E. 3 and
Figure E. 4 show examples of how the power in frequency bands can vary with time. (It is cautioned that
these are but examples and are not inclusive of all possible PSD variants. Note also that other DDS data
rates exist.)
Several conclusions may be drawn regarding crosstalk from DDS and T1:
1) Crosstalk can exceed that commonly modeled based on a random data assumption for T1 and
ISDN in certain frequency bands by as much as 20 dB.
2) Crosstalk should be considered to be time varying. The time duration and time change of each
PSD variant is not predictable nor controllable. It is caused by user data content.
3) Crosstalk in a wide band (for example, tens of kilohertz) can change at least 25 dB .
4) Crosstalk in a narrow band (for example 3 kHz) can change at least 45 dB.
5) The above affects all frequency bands from near DC to the highest range of T1.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
1/8 O N E s
A ll Z E R O s
Q R R S “R an d om ” D ata
-2 5
-3 5
dB
-4 5
-5 5
T 1 P ow er Spectral D en sity V ariations
A ll O N E s
-9 5 dB m /H z
20
kH z
12 0
kH z
22 0
kH z
32 0
kH z
42 0
kH z
52 0
kH z
62 0
kH z
72 0
kH z
82 0
kH z
92 0
kH z
10 20
kH z
Figure E. 1 – Examples of T1 power spectral density variations
D D S P ow er Spec tral D e nsity V ariations
-25 dB m /H z
-35 dB m /H z
-45 dB m /H z
20 47 R a ndom ,
all O N E s,
H D L C flag patterns
-55 dB m /H z
-65 dB m /H z
-75 dB m /H z
20
kHz
120
kHz
220
kHz
32 0
kHz
42 0
kHz
52 0
kHz
62 0
kHz
72 0
kHz
82 0
kHz
92 0
kHz
Figure E. 2 – Examples of DDS power spectral density variations
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Data Dependent Power Change in the 20 - 420 kHz Band
5
Random Data Level
0
-5
-10
dB
-15
-20
Any
Duration
Any
Duration
All ONEs Level
-25
-30
Time
Figure E. 3 – Data dependent power changes in a wide band due to T1 data patterns
Data Dependent Power Change in a Narrow Band at 193 kHz
-30
ZEROs Data Level
-40
-50
dBm/Hz
-60
Random Data Level
-70
ONEs Data Level
-80
-90
Time
Figure E. 4 – Data dependent power changes in a narrow band due to T1 data patterns
126
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex F: Non-continuous CO signaling events
(informative)
F.1 Ringing
Ringing in North America is an AC voltage superimposed on a DC bias. Many installations in the US use
non-sinusoidal 20 Hz ringing with a nominal rms. 90 volts at the ringing source. Other frequencies in use
14)
range from 16 2/3 to 66 2/3 with voltages from 85 to 135 . One ANSI standard sets the maximum
15)
voltage limit to 150V rms and notes cases where it can attain 175V rms.
Ringing is a non-continuous disturber. At the beginning of each ringing burst there is a transition from -48Volt battery feed to -48-Volt with superimposed AC ringing. Nominal interrupts are 2 seconds on and 4
seconds off. Custom ringing cadences with multiple ringing, such as triple cadences, are common. The
ringing waveform is ideally a sine wave with its axis of symmetry shifted -48-Volts from zero. The ringing
burst can be characterized in terms of 100's of milliseconds as shown in Figure F. 1. In this depiction, the
sine wave starts and stops in unity with the DC bias and represents the best case relative to instantaneous
power changes as a result of ring application and trip.
Elements of synchronization are related to the application of ringing in many applications, such as the use
of a common ringing bus serving hundreds of lines. Central office implementations, in many cases,
simultaneously ring multiple lines with concurrent cadence. As such, the application and withdraw of
ringing is generally without regard to the phase angle of AC energy. The peak voltage when ringing is
tripped can be the sum of the DC and greatest AC or approximately 170 volts as shown in Figure F. 2.
In its worst case, a generated ringing waveform is a trapezoidal shape, which means it has higher
frequency components occurring at 25 mS intervals. Transient energies often result from gap switching in
the ringing generator as shown in Figure F. 3.
Various forms of ringing cadence exist as noted above such as "triple," "double," "long/short," "coded,"
16)
and "teen ringing. For example, triple ringing bursts three times within 1800 mS as shown in Figure F. 4.
These have the effect of increasing random, ring application and removal impulse effects as shown
above.
Telephone Switching systems typically have the capability of ringing as many as one-fourth of the
connected lines. Accordingly, in the worst case, an average of 6 of the 25 pairs in a binder group could be
in some phase of ringing application or removal.
F.2 Supervision (hook flash)
As shown in Figure F. 5, the DC potential is applied to the customer loop through a battery-feed device
consisting of two inductive coils in series with tip and ring. An idle circuit is nominally 48 Volts with no
current flowing.
During service initiation, the customer closes the loop and a transient voltage migration occurs within the
cable pair of greater than 40 volts, that is, it drops to 6 volts across the telephone set.
–––––––
14)
GTE Customer Handbook - 500, Issue 1, 1972
15)
T1.401-1993, "Interface Between Carriers and Customer Installations--Analog Voicegrade Switched Access Lines
Using Loop-Start and Ground-Start Signaling."
16)
ANSI T1.401.02-1995, "Interface between Carriers and Customer Installations--Analog Voicegrade Switched
Access Lines with Distinctive Alerting Features."
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
A sudden voltage change in the presence of distributed capacitance can couple as not all of it gets
cancelled out. A wave front of the sudden change in loop voltage is unbounded and currently unrestricted.
POTS filters for DSL are only on the pair connected to and adjacent pairs are susceptible to the type of
inductive kick as described above. This exists throughout the network today.
F.3 Dial Pulse
These are periodic transitions from on-hook to off-hook in order to convey numeric values typically at 10
pulses per second in North America. Usually, 40 ms make (close) versus 60 ms break (open) as there is
less time required to build the magnetic flux versus lose it. As soon as the dial on the phone is turned, all
of the resistance in the circuit (all the handset circuitry) is shunted. There is a solid short in the circuit in
order to get ready to go to maximum current.
The shorter the loop the higher the current but the less the cross talk potential. This is just the opposite of
longer loops. These phenomena exist on short and longer loops.
128
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
-138 V
nominal peak
90 Vac rms.
4200 mS
1800 mS
Figure F. 1 – Standard ringing potential with best case start/end
20Hz or 50 mS Peak to Peak
~170V worst case
- 90Vrms ac
2
- 48Vdc
0
+ 48Vdc
Figure F. 2 – Standard ringing potential worst case start/end
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
The phase at the transition edge of ringing can be > 500Hz
infinity
Time i+2
Time i
< .5 mS
25 mS
1 mS
Time i+1
Figure F. 3 – Ringing waveforms (worst case generalization)
1800 mS
Figure F. 4 – Triple ringing interval
48Vdc
Figure F. 5 – Simple battery feed arrangement
130
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex G: ADSL Calculated Capacities
(informative)
The following assumptions were used in the analysis:
•
DMT performance computations were performed using following ADSL parameters.
−
Effective Coding Gain 3 dB
−
Required Margin 6 dB
−
Downstream: tone 33 to 255, including pilot-tone carrying data
−
Upstream: tone 6 to 31, including pilot-tone carrying data
−
Minimum number of bits per carrier: 2
−
Maximum number of bits per carrier: 14
•
The DMT capacity calculated was adjusted to remove cyclic prefix and sync-symbol by multiplying
with a factor 512/(512+32) * 68/69). No rounding-down to 32 kbps multiples.
•
Both NEXT and FEXT was included in the evaluations.
•
A white noise of -140 dBm/Hz was added to all cases to model the line background noise.
•
ATU-C and disturbers were assumed to be co-located.
•
ATU-R and disturbers were assumed to be co-located.
•
All evaluations were performed using 26awg loops at varying lengths.
•
HDSL, HDSL2, and T1 were assumed to be repeatered. Only TU-R NEXT and TU-C FEXT is
considered for ATU-R downstream evaluations and repeatered NEXT / FEXT affects are not
considered.
•
T1 disturber was assumed to be in adjacent binder with 15.5 dB reduction.
•
DDS 64K PSD defined by ANSI T1.410-1992
•
ISDN PSD defined by ANSI T1.413-1998
•
HDSL PSD defined by Spectrum Management Class 3
•
HDSL2 PSD defined by Spectrum Management Class 4
•
ADSL PSD defined by Spectrum Management Class 5
•
EC ADSL PSD defined by ANSI T1.413-1998
•
T1 PSD defined by ANSI T1.413-1998
The above assumptions reflect a worst-case engineering model of conditions that approximately represent
less than or equal to 1% of the anticipated real-life loop plant. Due to the numerous unknowns including
actual deployment distributions, RFI interference, loop plant parameters, and inter-loop interference, the
exact percentage of lines seeing conditions equal to or worse than this model are not predicted. ADSL
calculated capacities shown in Table G. 1 are not intended to be performance target rates for systems in
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
the real world.
Table G. 1 – ADSL Calculated Capacity
Basis
System
ADSL Down
ADSL Down
ADSL Down
ADSL Down
ADSL Down
ADSL Down
ADSL Up
ADSL Up
ADSL Up
ADSL Up
ADSL Up
ADSL Up
Loop
Length # Dist.
DDS 64K
26 AWG
9
20
6868
12
24
2461
13.5
24
1354
15.6
10
614
16
10
504
17.7
10
233
9
20
1127
12
20
832
13.5
20
677
15.6
10
523
16
10
483
17.7
10
285
Calculated Capacity
ISDN
HDSL
HDSL2
ADSL
EC ADSL
T1
8558
3660
2110
992
796
227
1026
725
571
421
383
155
5292
360
0
0
0
0
651
317
186
104
88
48
6760
2493
1269
346
196
0
743
422
259
160
140
72
6429
3101
1945
927
766
226
1302
1170
1086
971
944
801
6429
3114
1945
927
766
226
736
416
254
157
137
71
2444
1021
626
350
298
122
1435
1382
1305
1190
1157
1007
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex H: Technology Effects Of and On Legacy Systems
(informative)
H.1
T1 Carrier
In order to assure compatibility with T1, spectrum management class 5 DSL transmission systems and T1
systems should be assigned to pairs that are in different binder groups whenever possible.
H.1.1 Margin computations for linear equalization systems (e.g., T1)
To be added later.
H.1.2 Compatibility with AMI T1
The test for compatibility with repeatered AMI T1 assumes the following:
-
The repeater section is operating with 3 dB of margin,
-
The margin, after taking the proposed DSL system into account, must be at least 2.0 dB
-
The loss of the first repeater section out of an office is assumed to be 22.5 dB at 772 kHz.
-
The loss of subsequent sections is assumed to be 32 dB at 772 kHz.
Power summing margins, to obtain the 2.0 dB of margin, yields a required minimum margin, due to the
DSL system(s) alone, of 9.0 dB.
It has been found empirically that, on a repeater span having a loss of 22.5 dB at 772 kHz, the maximum
-7
noise that can be tolerated at the repeater input, while maintaining a BER of 10 , is -27.5 dBm. The
maximum noise due to DSL system(s) on the first repeater section out of an office, then, shall be equal to
or less than –36.5 dBm (-27.5 – 9.0). Similarly, the maximum noise for a 32 dB span is -40.5 dBm. The
maximum noise due to DSL system(s) for subsequent repeater sections, then, shall be equal to or less
than –49.5 dBm (-40.5 – 9.0).
When evaluating the noise coupled into the repeater, the following equation, developed via curve-fitting,
shall be used to model the repeater input filtering.
Gain(dB ) = a5 f 5 + a 4 f 4 + a 3 f 3 + a 2 f 2 + a1f + a0
where f is in MHz and the coefficients for both the 22.5 and 32 dB sections are shown in Table H. 1.
Defining C(f) as the 1% Unger two-piece model (see Figure A. 1 and Table A. 5) and Gain(f) as given
above, the following conditions for compatibility with T1 carrier must be met:
1.544MHz
End Section (22.5dB ) :
∫ PSDDisturber ∗ C(f ) ∗ Gain(f )df ≤ −36.5dBm
0
1.544MHz
Mid − Span Section (32dB ) :
∫ PSDDisturber ∗ C(f ) ∗ Gain(f )df ≤ −49.5dBm
0
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
H.1.2.1 Evaluation Loops
To be added
H.1.2.2 Reference crosstalk environments
To be added
H.1.2.3 Margin computation
To be added
Table H. 1: Coefficients for T1 repeater input filtering gain equation
Coefficient
Value for 22.5 dB section
Value for 32 dB section
A0
-12.91476008173899
-21.84038057235726
A1
15.74168401196194
40.22938541210919
A2
20.75952294972729
-2.99965401635352
A3
-36.60781681972960
-31.38386179570797
A4
13.09484055899603
18.63736172126514
A5
-0.91231176505002
-3.26384215013252
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex I: C-code
Table I. 1: C-code for DMT margin computation
float dmtmrgn(
float
*signal, /* array of received signal psd samples (resolution =
FDELTA Hz)*/
float *noise,
/* array of received noise psd samples (resolution =
FDELTA Hz) */
int
rate, /* desired bit rate, expressed in units of bits per second
per FDELTA ) */
int
start, /* start of DMT bandwidth (sample number) */
int
end, /* end of DMT bandwidth (sample number) */
int
in_dB) /* FLAG: 1 = PSDs given in deciBels, 0 = PSDs given in
linear units */
{
int
float
float
float
j, firstpass;
snr;
snr_margin ;
delcap, totcap ;
snr_margin = MAXIMUM_VALUE;
firstpass = 1;
snr_margin += MARGIN_STEP;
do {
snr_margin -= MARGIN_STEP;
/* Compute capacity */
totcap = 0.;
for (j = start; j < end; j++) {
if (in_dB) snr = sig[j]-noise[j];
else snr = 10.*log10(sig[j]/noise[j]);
delcap = log(1. + pow(10.,
.1*(snr -snr_margin-SNRGAP))) / log(2);
if (delcap > MAXBITS) delcap = MAXBITS;
if (delcap < MINBITS) delcap = 0;
totcap += delcap;
}
if (totcap > rate && firstpass) {
snr_margin +=10.; totcap=0.; }
else firstpass = 0;
} while (totcap < rate);
return (snr_margin);
}
SNRGAP, MAXBITS, MINBITS, are all adjusted based on the DMT system being evaluated.
MAXIMUM_VALUE and MARGIN_STEP are control how fast and how accurately the routine
computes margin.
MAXIMUM_VALUE is the maximum margin of interest, the integration
begins there. MARGIN_STEP defines the accuracy of the result.
This is a draft document and thus, is still dynamic in nature.
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T1E1.4/2000-002R2 DRAFT ANSI T1.XXX-2000
Annex J: Informative references
(informative)
[1] B. J. Dunbar, et. al., “Dataport – Channel Units for Digital Data System 56-kb/s Rate”, BSTJ, vol 61 no.
9, November 1982.
[2] T. Berger & D. W. Tufts, “Optimum Pulse Amplitude Modulation Part I: Transmitter – Receiver Design
and Bounds from Information Theory,” IEEE Transactions on Information Theory, vol. IT-13, no. 2, April
1967.
[3] Committee T1 Technical Report No. 28, High Bit Rate Digital Subscriber Lines (HDSL)
[4] GTE Customer Handbook - 500, Issue 1, 1972
[5] Transmission Systems for Communications, Bell Telephone Laboratories, Fifth Edition, 1982.
[6] ASTM D 4566, Standard Test Methods for Electrical Performance of Insulations and Jackets for
Telecommunications Wire and Cable, 1994.
136
This is a draft document and thus, is still dynamic in nature.