The 7th International Conference on Power Electronics
PD21
October 22-26, 2007 / EXCO, Daegu, Korea
Analysis and proposition of a PV module integrated converter with high voltage gain
capability in a non-isolated topology
René P. Torrico Bascopé, Fernando L. M. Antunes
Federal University of Ceará
Department of Electrical Engineering
Energy Processing and Control Group
Fortaleza-Ceará-Brazil
rene@dee.ufc.br, fantunes@dee.ufc.br
Samuel Vasconcelos Araújo, Peter Zacharias,
Benjamin Sahan
ISET e.V. Universität Kassel
Power Electronics Division
Königstör 59, 34119 Kassel, Germany
Email: sva@ieee.org
Abstract— Usually considered as one of the future solutions for
grid connection of photovoltaic systems, module integrated
converters were already the focus of several researches and
projects. Most of the proposed approaches relied so far on the
use of high frequency step-up transformers either in isolated
operation or integrated in isolated dc-dc topologies. This paper
analyses the possibility of using non-isolated topologies to
achieve the necessary high-voltage gain for grid connection.
Several circuits were analyzed and the best suited one for the
current application was evaluated and optimized. Experimental
results are presented in the final section.
I.
INTRODUCTION
The first generation of grid connected photovoltaic systems
was composed of several strings of panels associated in
parallel and connected to a single inverter. Such centralized
approach had as disadvantages [1] the necessity of string
diodes (with their inherent power losses) and high voltage DC
cabling. Furthermore, since operation was limited to only one
maximum power point (MPP) for the whole array, mismatch
losses reduced the system efficiency. Finally, due to the high
power level of the inverter, there was little flexibility on
system expansion.
At present, most of the systems are composed of a single or
multiple strings of modules connected to an inverter, the
so-called string and multi-string inverters. This way, in
contrast with their predecessors, losses due to maximum
power point tracking (MPPT) mismatch were reduced, but not
totally eliminated, and string diodes are not necessary
anymore.
The next expected evolution on grid connected
photovoltaic systems is considered as the integration of the
converter in the module and is usually named AC Module,
since the output of the panel can now be directly connected to
the mains. A major highlight of such an approach is the
elimination of MPPT mismatches, allowing optimal coupling
between panel and inverter and therefore increasing generated
power per module. In addition, the small level of power and
modularity allows flexibility in system expansion and low
purchase investment, being considered as the best option for
end-user applications [2]. Since the output of the panels can
be directly connected to the grid, DC cabling and installation
expertise are not necessary, allowing considerable reduction
in installation expenses. Though a higher production cost per
produced Watt is expected for this approach, the mass
production of such small units may in the end increase
competitiveness due to economy of scale. A disadvantage of
d
the module integrated solution is the strict requirement of a
design with long lifetime under harsh ambient conditions that
needs to be tackled by a highly robust power electronic
design, since maintenance is much more complex than the
ones of traditional string inverters.
OVERVIEW AND REQUIREMENTS OF MODULE
INTEGRATED CONVERTERS (MIC)
II.
Prior to the comparison of the circuits, it is necessary to
discuss some important issues regarding the proposed
application.
The trinity efficiency, cost and lifetime gives in general the
orientation on the topology choice and on most of the project
development. It is mainly affected by the amount and rating of
components in such a way that simple topologies are
preferable with the condition that components are not under
severe current or voltage stress.
Cost itself has been mainly one of the critical obstacles for
the further expansion of module integrated solutions. Aside
from the specification of the components, it is also strongly
influenced by the fact that the lower the power rating is, the
higher is the cost per produced kWh [3]. In order to reduce
such disparity, mass production is a mandatory condition and
may only be achieved by flexible solutions capable of
operating with most of the available panels in the market,
what leads to the necessity of high voltage gain capability as
PV panels usually have output voltages around 30 and 50V.
Still regarding costs, another issue often forgotten is that,
as previously explained in the introduction, module integrated
converters are capable of promoting significant reduction on
the installation costs. When one analyses the cost components
of photovoltaic power (as depicted in Fig. 1) [4], it becomes
clear that the higher expected price paid for the converter will
Silicon
Ingots
8%
8%
Module Total
60%
Wafers
8%
Cells
16%
Installation
32%
Inverters
Modules
20%
8%
Fig. 1: Composition of the photovoltaic power costs [4].
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be probably justified by the superior potential reduction on
the installation costs.
Lifetime is closely related to reliability and expected
degree of failure of components and the converter shall be
projected to endure at least the expected useful lifetime of the
photovoltaic module (generally considered as 20 years). As
an aggravating fact, harsh ambient conditions are expected
along with a wide range of temperature variation during a
single day; fact that impress mechanical stress on components
and insulation materials. Among all passive or active
elements, electrolytic capacitors are the ones with the shortest
lifetime [5] and with the further drawback that aging increases
the values of the ESR (equivalent series resistance) and
consequently also the losses. As a conclusion, the chosen
topology shall not require high capacitance values in order to
allow the use of the film technology, especially from the
polypropylene type, which has by far a longer lifetime, good
thermal and electrical stability and finally low ESR.
Power density is also considered as an important factor in
order to reduce installation complexity and costs and is often
achieved by increasing the switching frequency. Nevertheless
a trade-off shall be analyzed in order to avoid reduction of the
efficiency by increasing switching losses.
A. MIC Topologies
In the last seven years, the photovoltaic modules industry
introduced several new cell technologies alongside with
further improvements on already existing ones. As expected,
the parameters of the panels in general also changed, as can be
observed in the comparison between the years 2001 and 2007.
Voltage under Standard Test Conditions [V]
60
Monocyistalline Silicon
Polycrytalline Silicon
Others
50
40
30
20
10
Year 2001
0
0
50
100
150
200
250
300
Output Power under Standard Test Conditions [W]
Voltage under Standard Test Conditions [V]
60
Monocyistalline Silicon
Polycrytalline Silicon
Others
50
40
30
20
10
Year 2007
0
0
50
100
150
200
250
Output Power under Standard Test Conditions [W]
Fig. 2: Evolution of photovoltaic modules parameters [6,7].
300
From Fig. 2, it can be noticed an average increase in the
generated power and voltage under STC (Standard Test
Conditions).
Nevertheless, the output voltage level of most panels is still
not high enough to directly feed an inverter for connection to
the grid. Because of this fact, two approaches have been
observed concerning most module integrated converters.
A. Single-stage converters coupled with photovoltaic modules
with high voltage output
The first approach is rather new and consists of using
specially designed modules with a high output voltage in the
range of 200 to 400V, depending on the local grid voltage [8],
like depicted in Fig. 3 a). This way, only a single stage is
required, reducing costs and control complexity and
increasing overall efficiency. While this approach looks
promising it still remains that such solutions are restricted in
the application spectrum to only few panels with compatible
output parameters; what may not allow a large scale of
production and necessary cost reduction in order to increase
competitiveness.
400V
Module
a)
40V
Module
b)
Fig. 3: Principle configurations of MIC with a) 3-phase, single-stage, b)
Single-phase, dc–dc converter, dc-link capacitor and inverter.
B. Isolated Multi-stage converters coupled with modules with
low voltage output
The second and most common approach is a single-phase
approach using standard modules with lower voltage levels by
employing multi-stage topologies (Fig. 3 b)). A high variety
of different topologies were used in the past and described in
the literature [1]. In most cases the topologies were
characterized by high frequency step-up transformers to
perform the necessary high voltage gain.
Regarding the dc-dc converter, in order to reduce the
number of stages and increase efficiency and reliability,
several approaches used transformers belonging to topologies
derived from isolated dc-dc converters like push-pull [9, 10]
and flyback [11-14]. A common drawback of such
approaches in comparison to others employing a more
complex switching scheme is the higher voltage stress across
semiconductors. Therefore, the smaller amount of switching
devices will not necessarily lead to much efficiency
enhancement, since parameters like the drain-source
on-resistance of MOSFETs are much affected by the voltage
rating (increasing factor of 70% or more for best-in-class
devices rated at 150 and 200V, for example). A further
disadvantage inherent to push-pull derived topologies is the
necessity of balanced magnetizing mechanisms in order to
avoid transformer saturation.
In general, a clear advantage of all isolated approaches is
the obtained galvanic insulation and power decoupling.
Drawbacks are the lower levels of efficiency due to high
frequency switching to reduce transformer size, losses on the
transformer itself and increasing complexity.
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C. New approach: Non-isolated dc-dc converter with high
voltage gain capability.
An alternative to the previous categories that was not yet
analyzed in the literature is the use of non-isolated dc-dc
converters that are capable of reaching the required voltage
gain with high levels of efficiency. Since galvanic insulation
in photovoltaic applications is not a requirement in USA (only
for this AC modules [15]) and in most of the European
countries (except Spain [16]), no further limitations for the
use of non-isolated converters on most photovoltaic panels
are expected. However, [17] revealed that there might be
lifetime issues with some thin-film technologies (e.g. CdTe)
when PV panels are not grounded.
Several high voltage gain topologies were already
proposed and will be evaluated on the following item.
At last but not least, the operation of a non-isolated
photovoltaic system prompts attendance to further safety and
regulatory requirements like limitation of the dc component
on the output current, islanding detection and handling of
leakage currents; though such requirements are out of the
scope of this paper.
III.
NON-ISOLATED DC-DC CONVERTERS WITH
HIGH-VOLTAGE GAIN CAPABILITY
A. Comparison of available topologies
Taking into consideration that most of the mono- and
polycrystalline modules available nowadays in the market
have a MPP voltage level for STC under 40V, the minimum
required voltage amplification to reach 400V for feeding the
input of a grid-connected inverter is at least 10 and values of
20 or more may be necessary under lower irradiation levels.
Such high voltage gain can be theoretically obtained with
the classical boost converter (see Fig. 4); though a serious
limitation for practical operation lies on the fact that very high
values of duty cycle are necessary. The operation at such
conditions needs very precise control and drives for the
switches in order to avoid instability, since small changes in
the value of the duty will lead to great variations in the output
voltage, what makes such implementation problematic and
expensive.
Furthermore, the theoretical gain is normally not
achievable for very high duty cycle values due to operational
limitations, namely, losses at inductances and capacitors,
reverse recovery and turn-on-off time of diodes, switching
transients and core susceptibility for fast changing flux
intensity (what may increase losses) [18]. Therefore, for
obtaining a high voltage gain it is advisable to employ special
topologies that do not require very high duty cycles.
Fig. 4: Boost converter and full-bridge inverter topology.
The first analyzed circuit is depicted in Fig. 5 a) and was
proposed in [19, 20] by the modification of a flyback
converter with the objective of improving the coupling
coefficient. Since the leakage of the coupled inductors lead to
high voltage spikes across the active switch, degrading
efficiency, it was necessary to employ a clamping technique.
Drawbacks of this topology are the input pulsating current
(what may increase the MPPT mismatch), inverted polarity of
the load voltage (likewise the buck-boost converter) and
finally necessity of high turns ratio in order to achieve the
desired voltage gain.
Similarly to the interleaved boost, the circuit introduced in
[21] and depicted in Fig. 5 b) is composed of two boost
converters coupled via an autotransformer with 1:1 turns-ratio
and inverted polarity in order to allow equal current share
between the active switches. The output was configured as a
voltage doubler rectifier. An advantage of this approach in
comparison to the previous is that the input current is
non-pulsating and with low ripple. As negative aspects, the
voltage stress across active switches can be higher than half of
the output voltage due to layout inductances.
Another topology also similar to the interleaved boost
converter was proposed in [22], Fig. 5 c) with the possibility
of using several stages with multiplier capacitors in order to
increase the voltage gain. The voltage stress is limited to half
of the output voltage for the configuration with just one
multiplier stage. A snubber circuit may be required due to the
sum of the reverse recovery currents of the output and
multiplier diodes that increase turn-on losses on the active
switches.
Fig. 5: Some of the analyzed topologies with high voltage gain capability.
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Fig. 6: Other analyzed topologies with high voltage gain capability.
A circuit composed of three coupled inductors was
introduced in [23] as illustrated in Fig. 6 d), with high
utilization grade of the magnetic core. Depending on the
turns-ratio between the inductors, the voltage stress across the
active switch can be as low as 15% of the output voltage. Like
other topologies requiring coupled inductors, the high
turns-ratio between windings increases system complexity
and losses. A simplified variation was also proposed in [24]
with just two coupled inductors, though their leakage
inductance may lead to a ringing phenomenon in the voltage
across diode D2.
In Fig. 6 e), [25], a hybrid boost-flyback converter was
proposed with the advantage of low voltage stress across the
active switch, which is equal or lower than half of the output
voltage; depending on the turns-ratio between the windings.
In addition, such voltage is naturally clamped by output
capacitor that additionally recycles the leakage energy to the
output. As a disadvantage, the input current is pulsating and
requires an additional input filter.
Another topology similar to the previous one and
consisting of a boost converter with coupled inductors was
introduced in [26] and is illustrated in Fig. 6 f). The difference
now is that a voltage doubler rectifier is applied to the
secondary winding. Drawbacks of this approach are the
necessity of a filter to treat the input pulsating current since
the inductor is totally discharged during normal operation and
the critical charge of capacitor C2 in order to achieve the
required voltage gain.
The topology proposed in [27] and depicted in Fig. 6 g)
also used coupled magnetic structures, though in a different
way than previous approaches. Here, a three-state switching
cell composed of two active and two passive switches
employs an autotransformer with equal number of turns and
inversed polarity in order to provide balanced current share
proposed in [28] was used. Coupled with such transformer is a
secondary winding that, with variation of turns-ratio, allows
simultaneously high voltage gain and reduction of voltage
stress across active switches. The use of two active switches
in parallel with reduced voltage rating allows the
minimization of conduction losses. A further advantage is that
frequency across the inductor is the double of the switching
frequency; allowing reduction of the size without the
drawback of higher switching losses. Finally, the input current
is non-pulsated and with little ripple, what leads to low MPPT
mismatches. The voltage level across the active switches is
naturally clamped by the capacitor C1.
Rather than employing coupled magnetic components like
most of the previous topologies, [29, 30] used the switching
capacitor technique, where capacitors are connected in series
and in parallel in each switching cycle transition. Since the
magnetic switches are not available, a higher number of active
switches are necessary in [29], increasing losses and control
complexity. In addition, the converter presented in [30], as
illustrated in Fig. 6 h) does not have a high enough voltage
gain for the proposed application.
B. Chosen topology analysis
Taking in consideration the factors presented in the item C,
the circuit proposed in [27] was chosen as the best suited for
the proposed application.
The operation of the converter can be divided in four stages
Fig. 7: Operation stages of the chosen converter.
as depicted in Fig. 7.
During the first stage, the input inductance is charged with
energy provided by the source, while both switches S1 and S2
are turned-on and share equal amounts of current. The second
stage begins as the switch S1 is turned-off. Diode D1 and D2
are forward biased and the capacitors C1 and C2 are charged
(the last one by the current flowing through the secondary
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the influence of ambient temperature on module voltage was
taken into consideration and hence, only the panels that have
an output voltage lower than 40 V under STC are compatible
with the projected converter.
The output voltage was chosen as 400V, in order to feed a
full-bridge inverter connected to a grid with a RMS voltage
level of 230V. From the selected parameters, the minimum
and maximum voltage gains under nominal power are
respectively 8.8 and 11.5.
For the sake of simplifying the circuit and reducing the
amount of components, only one multiplier stage will be used.
The voltage gain is given by (1) and depicted in Fig. 9 for
some values of turns-ratio, which is represented by the letter
“a” and defined as the ratio between N1 and N2.
30
Voltage Gain
winding). During this stage, the inductance is being
discharged. The third stage is equal to the first one, with both
switches turned-on. Finally, the fourth stage starts when the
switch S2 is turned-off, forward biasing diodes Dp and D3. The
current flowing through the first diode charges once again the
capacitor C1 (for this reason its size is the half of the others).
Due to the relation of the polarities, the current in the
secondary winding has the opposite direction of the one
during the second stage and charges diode C3.
By observing the simplified theoretical waveforms in Fig.
8, it is possible to conclude that the converter needs to operate
at the overlapping mode; what means that the duty cycle of
each active switch must be higher than 50%. Such
requirement comes from the fact that the input inductance is
only charged during the period when both switches are
simultaneously turned-on. In addition, values too near to 50%
may increase the ripple at the input current, requiring larger
input inductances.
a = 2.5
20
a=2
a = 1.5
10
0.5
0.6
0.7
0.8
0.9
Duty Cycle
Fig. 9: Voltage gain as a function of the duty cycle and turns-ratio (a).
Fig. 8: Theoretical waveforms.
As a remark, the current in both primary windings is
equally divided only during the first and third stage. During
the second stage, in order to keep the resultant magnetic flux
at zero in the autotransformer core, the current in the winding
P1 is reduced by the half of relative ampere-turns of the
current in the secondary, while the current on the winding P2
is increased by the same quantity. During the fourth stage, the
inverse situation occurs.
Keeping in mind that such increments/decrements at the
current in both primary windings are directly proportional to
the relative number of turns of the secondary winding; large
values of turns-ratio shall be avoided in order to improve the
stability of the topology and reduce the possibility of reaching
discontinuous conduction mode under low values of input
power.
IV.
DESIGN AND OPTIMIZATION OF THE CHOSEN
TOPOLOGY
With the objective of allowing compatibility with most of
the mono- and polycrystalline modules available in the
market, the converter output power was chosen as 250W
while maximum and minimum input voltages under nominal
power are respectively 45V and 35V. As an important remark,
V
( a + 1)
GV = o =
(1)
Vi (1 − D)
Taking into consideration that the minimum required gain
will be 8.8 and that the converter shall operate on overlapping
mode with a certain margin of security (a minimum duty cycle
of 0.6 was chosen), the maximum allowable turns-ratio under
such requirements is approximately 2.5.
The maximum possible turns-ratio value was chosen for the
project since the higher it is, the lower is the voltage stress
across the active switches and diodes Dp and D1, as
calculated by (2). However, the voltage stress across diodes
D2 and D3 increases for higher values of a, since it follows the
same relation, but multiplied by “a”.
V
VDS S 1 = o
(2)
1+ a
Using a 20% margin of safety, it is possible to employ
150V rated MOSFETs. In order to choose the best-of-class
model, the concept of Figure of Merit (FOM) [31], was used.
It takes into consideration that conduction and switching
losses in a MOSFET are respectively affected by the value of
the drain-source on-state resistance (Rdson) and the gate
charge (Qc). As a remark, Rdson and Qc are inversely
proportional, so that models with very low resistances will not
have an optimal switching behavior and vice versa.
MOSFETs of several manufacturers were therefore analyzed,
and the results are presented in the Fig. 10. Due to availability
during the prototype assembly, MOSFETs FDP2535 from
Fairchild were the chosen one.
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voltages across the active switches S1 and S2 and the currents
through the secondary and primary windings of the
autotransformer. The obtained waveforms were a bitdifferent from the theoretical ones presented before because
of the leakage inductance and coupling factor between the
windings. Since no ringing phenomenon during turn-on of the
active switches was observed, being within the expected
limits, no snubber was necessary and in addition the gate
resistance value was reduced in order to minimize switching
losses.
0,04
FDP42AN15A0
0,035
IRF6643TR1
IRFB52N15D
2SK3590-01
0,03
R dson [ohm]
FDA79N15
0,025
SUP80N15-20L
0,02
FQH90N15
SUM85N15-19
0,015
FDB2532
IXFK180N15P
IRFS4321
0,01
0,005
0
0
50
100
150
200
250
300
T
1->
Gate Charge [nC]
Fig. 10: 150V Rated MOSFETs characteristics.
For D1 and Dp Schottky diodes were specified rated at
150V with low forward voltage drop. The level of voltage
across D2 and D3 required the use of ultra-fast diodes rated at
400 V, though 600V SiC diodes were also tested.
The calculated inductance for a maximum ripple of 20% of
the input current was 274 µH. A ferrite core with the ETD
geometry was chosen instead of an EE since it provides a
reduction of more than 11% on the mean length per turn for
the same cross-sectional area, decreasing cooper losses. In
order to increase the cooper fill factor, litz wire with a
rectangular profile was used.
The autotransformer was also built using an ETD core.
Interleaved winding was employed, as depicted in Fig. 11, in
order to reduce leakage inductance and proximity effect
between the layers.
2->
T
3->
T
T
4->
1 ) [Y TS he e
2 ) Y[ TS he e
3 ) Y[ TS eh e
4 ) Y[ TS eh e
1(t ]) .C h1 2 0 V
1 0u
1(t ]) .C h2 2 0 V
1 0u
1(t ]) .C h3 2 A
1 0u
1(t ]) .C h3 2 A
1 0u
Fig. 12: Drain-source voltage across switch S1 and S2; current across
secondary and primary autotransformer windings (200V/div; 2A/div;
10µs/div).
The input and output voltages and currents are presented in
Fig. 13, with the remark that the desired voltage gain was
reached using the specified duty cycle. The input current has
low ripple content, what increases the efficiency of the MPPT
and reduce size of the input capacitor.
T
3->
T
4->
Fig. 11: Cross-section of the magnetic core depicting the interleaved
winding used in the autotransformer.
The output capacitors were specified to limit the high
frequency ripple to 2.5% at the output voltage. The low value
of required capacitances allowed the use of polypropylene
film capacitors. The low voltage rating (150V for C1 and
200V for C2 and C3) permitted further volume reduction of
the specified capacitors.
The low frequency ripple due to single-phase connection
will require a larger dc-link capacitor [1], but this
specification is out of the scope of this paper.
V.
EXPERIMENTAL RESULTS
In order to evaluate the performance of the topology for the
proposed application, a 250W prototype was built. The
specified switching frequency was 20 kHz.
In Fig. 12 is depicted the measured drain-to-source
T
1->
T
2->
Fig. 13: Output voltage, input voltage, input current, output current,
(200V/div; 50V/div; 5A/div; 1A/div;10µs/div).
In Fig. 14 are presented the efficiency curves for different
levels of output power and two input voltage levels. The use
of smaller gate resistances allowed the reduction of the
switching losses and further enhancement of the efficiency.
SiC diodes were also tested, though for the proposed
configuration their use actually reduced the efficiency a bit;
possibly because of the higher forward-voltage drop value in
comparison to the ultra-fast diode. In Fig. 15 is depicted the
experimental set-up.
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98
[7]
97
[8]
Efficiency [%]
96
95
94
93
92
fs= 20 kHz
Vi = 45 V
ηmax = 97,68%
ηeur (250W) = 96,29%
[9]
fs= 20 kHz
Vi = 35 V
ηmax = 97,29%
ηeur (250W) = 96,34%
[10]
91
0
50
100
150
200
250
Input power [W]
[11]
Fig. 14: Efficiency as a function of the output power.
[12]
Lb
autotransformer
PWM Control
C3
S1
[13]
C2
S2
[14]
C1
Gate driver
[15]
Fig. 15: Experimental set-up.
VI.
CONCLUSIONS
This paper proposed the use of non-isolated dc-dc
converters with high voltage gain integrated to a photovoltaic
module. Several topologies with such capability were
compared and the best suited one was projected following
optimizing procedures. A 250W laboratory prototype was
built and experimental results were evaluated. The converter
reached a higher degree of efficiency in comparison to the
original application, mainly due to the use of optimized
semiconductors and other project parameters. The efficiency
can be further enhanced by optimizing the circuit layout and
reducing the current density in the magnetic components.
As next steps, the inverter can be designed and the
interaction of the system with the photovoltaic module and
grid can be examined.
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