Hybrid Multiconverter Conditioner Topology for
High-Power Applications
María Isabel Milanés-Montero, Member, IEEE, Enrique Romero-Cadaval, Senior Member, IEEE, and
Fermín Barrero-González, Senior Member, IEEE
Abstract—A novel multiconverter conditioner topology and its
control stage are proposed in this paper. It is formed by an active
conditioner in parallel with a hybrid conditioner composed of
an active filter in series with one or more passive filters. This
topology allows the reduction of the inverter ratings, constituting
an effective solution at high-power levels. Collaborative control
strategies are developed for the new topology, which share the
compensation objectives between the two converters. These control strategies and the tracking techniques are based on estimating
the load current, achieving new algorithms with a reduction in the
number of meters in the control stage. The conditioner operates
properly in three-phase four-wire systems reducing the harmonic
distortion and/or imbalance and attaining the unity displacement
power factor. Experimental results are included for the testing of
the topology and its control.
Index Terms—Active filter (AF), hybrid filter, multiconverter
topology, power conditioner.
I. I NTRODUCTION
T
HE use of nonlinear loads injecting harmonic and reactive
current components into the electrical power system has
undergone rapid growth in recent years. The reduction or
elimination of these components can be achieved by using compensation equipment installed at the point of common coupling
(PCC). Conventional topologies, such as passive filters, active
or hybrid conditioners, or universal conditioners, have been
developed for this purpose [1]–[11]. However, these topologies
have restrictions to be used in high-power applications due
to the semiconductor technology limits regarding the inverter
rating [12], [13].
A solution to this drawback is the employment of two
or more devices to collaborate in the compensation. Seriesconnected active multiconverter conditioners [14]–[16], a
thyristor binary compensator in parallel with an active conditioner [17], and parallel-connected active multiconverter conditioners sharing the dc link [13], [17] or with the independent
dc bus [14], [19]–[22], have been developed using different
collaborative strategies. In this paper, a new parallel-connected
multiconverter topology, in which one of the equipment is a
hybrid conditioner, is proposed. We call it the parallel active
Fig. 1. Block diagram of the parallel-connected multiconverter topology: two
power stages (active filter in parallel with a hybrid filter, sharing the dc bus) and
one collaborative control stage.
with parallel Hybrid conditioner (pApH). It is formed by two
converters connected in parallel, sharing the dc bus (see Fig. 1).
One of the converters, termed slow because it operates with a
low switching frequency, is responsible for the fundamental and
dominant harmonic components in the load current. This device
is aided by the fast conditioner, which operates at a higher
frequency, compensating the higher harmonic components.
The main advantage of this topology is the separation of the
correction which leads to a reduction in the inverter ratings,
making this equipment viable at high-power levels. Another
advantage, whose demonstration is out of the scope of this
paper, is the reduction in volume, losses, and price of the
conditioner due to the possibility of using ferromagnetic core
filter inductors for the slow converter [18].
The sharing strategy between the two conditioners and their
respective principles of operation are explained in Section II.
The hybrid multiconverter topology is described in Section III.
The novel control stage, based on the load current estimation
by means of the source current measurement, is explained
in Section IV. Experimental results for a 1.2-kVA laboratory
prototype are presented in Section V. Finally, in Section VI, a
comparison between a hybrid multiconverter conditioner and
a classical shunt active monoconverter conditioner is carried
out by a simulation to calculate the design rate power of the
inverters in each topology. It is demonstrated that the novel
topology achieves a decrease in the inverter ratings, making it
useful for high-power applications.
II. S HARING S TRATEGY AND P RINCIPLES OF O PERATION
Manuscript received November 15, 2009; revised April 16, 2010 and
July 1, 2010; accepted July 8, 2010. Date of publication July 29, 2010; date
of current version May 13, 2011. This work was supported by the Junta de
Extremadura (Regional Government), Spain, under project PDT08A046.
The authors are with the Power Electrical and Electronic Systems Research
Group, School of Industrial Engineering, University of Extremadura, 06006
Badajoz, Spain (e-mail: milanes@unex.es).
The hybrid filter topology chosen for the multiconverter
pApH conditioner is the Active filter in series to Passive filter
(AsP) because for high-power loads it constitutes the most
economic solution due to the reduction attained in the inverter
power [12], [23]. If this hybrid filter is designed and controlled
TABLE I
F UNCTIONS OF THE ACTIVE E QUIPMENT OF E ACH C ONDITIONER
Fig. 2. Correction using a hybrid multiconverter pApH conditioner.
to behave as a dominant harmonic filter (DHF) [24] it will
be formed by an active filter (AF) in series with one or more
parallel passive filters (PFs) tuned to the dominant harmonic
frequencies in the load current, that are usually of low order.
These components would be mainly filtered passively, while the
AF, which will have a low rating, will collaborate in overcoming the drawbacks of passive filters working alone. Also, the
hybrid filter could compensate dynamically the displacement
power factor, dPF. As the harmonics in the reference current
for this filter are of low order, a low switching frequency could
be used for the inverter of this device, which is called the slow
conditioner.
The other parallel active (pA) conditioner will then be responsible for the higher order harmonics, which usually have
lower amplitudes, so the inverter rating will be smaller. The
tuned harmonic components left uncompensated by the slow
equipment to avoid passive filter overload and the switching
harmonics of the hybrid conditioner are also potential tasks for
this equipment. As high harmonic components will be entrusted
to this conditioner, a higher switching frequency would be
needed, and it would act as the fast device.
This sharing strategy demonstrates the advantage of this
topology compared to a hybrid monoconverter AsP conditioner
[9]. If reactive power is proposed as a compensation objective,
the passive filters have to be designed with a narrow bandwidth.
It means that the hybrid equipment could filter out only the
lower order frequencies in the load current if an admissible
inverter rating for high-power applications is looked for. The
hybrid filter attains a selective but not global compensation. The
proposed multiconverter topology avoids this drawback, since
the fast conditioner compensates the higher harmonic components in the load current, achieving an overall compensation.
If both filters operate as current-controlled sources, the
single-phase equivalent circuit of the multiconverter topology
is the one shown in Fig. 2, where iAFs and iAFf are the currents
injected by the inverters of the slow and fast conditioners,
respectively.
The reference current for the hybrid filter, iHF,ref , behaving
as a DHF is as follows:
lim
iHF,ref = iAFs,ref = iL1 − i+
L1d + iLhPF
(1)
where iL1 is the fundamental load current; i+
L1d is the active positive-sequence fundamental load current component;
and ilim
LhPF is the limit tuned load current allowed by the
conditioner to avoid passive filter overload, calculated as
follows:
ilim
LhPF
=
lim
if ILhPF ≤ IFPh
PF
iLhPF ,
iLhPF
ILhPF
lim
· IPFh
,
PF
lim
if ILhPF > IPFh
PF
(2)
with ILhPF being the rms value of a tuned harmonic component
lim
being the maximum rms value
in the load current and IPFh
PF
of each tuned harmonic current in the passive filter(s).
One can notice that with this reference current, the hybrid
filter avoids reactive power overcompensation in case of loads
with lower lagging reactive power than the fixed installed
passive filter reactive power.
The reference current for the fast conditioner will be needed,
collaborating with the slow device, to attain the desired source
current, iS,ref , which depends on the compensation objectives.
This current will be calculated as follows:
iAFf,ref = iL − iS,ref − iHF,meas
(3)
where iHF,meas is the measured hybrid filter current.
If a perfect harmonic cancellation (PHC) global control
strategy [25], [26] is selected for the multiconverter equipment,
aiming for the source current to be in phase with the positivesequence fundamental component of the voltage at the PCC,
the functions of the active equipment of each conditioner are
the ones summarized in Table I.
There is no type of interaction between both converters
because the hybrid filter operates as a DHF with the precise reference current proposed in (1). Possible interaction in terms of
zero-sequence components between both converters is avoided
since the slow converter is responsible for fundamental and
dominant harmonic frequencies. Only zero-sequence components at these frequencies due to unbalanced load current would
flow through the slow converter. Other possible zero-sequence
components (triplen harmonics or imbalance at other frequencies) will only circulate through the fast converter. For the same
reason, the reference current of the hybrid filter assures that
the current at the fast converter switching frequency does not
circulate through the slow one. On the other hand, the current
at the slow converter switching frequency is proposed as one of
the compensation goals of the fast converter (see Table I).
Fig. 3.
Three-phase four-wire source with nonlinear load and shunt hybrid multiconverter conditioner.
III. D ESCRIPTION OF THE T OPOLOGY
In this paper, a particular topology of the one presented in
Section II is used, which only has one passive branch (Fig. 3).
The active part of this topology is formed by a neutral-pointedclamped voltage-source inverter with six branches. The three
fast conditioner branches are connected to each phase of the
utility by a filter inductor, while the slow conditioner branches
are connected in series to the passive filter impedances. The
inductor of the passive filter will also filter the slow inverter
switching harmonics.
To tune the passive filters, a controlled rectifier supplying
a resistive load was considered as a nonlinear load since in
high-power applications, such as adjustable speed ac drives of
1000 hp and larger, or for dc drive applications to supply the
dc directly to the dc motor, phase-controlled rectifiers instead
of diode rectifiers, are typically used [27]. This load requires
fundamental reactive power compensation to maintain unity
dPF, and its dominant harmonics are the fifth and seventh
orders. As one passive branch has been selected for the hybrid
conditioner, it is possible to tune the passive filter to a lower,
higher, or intermediate order. After studying these options, the
tuning to the lowest harmonic (fifth order) was finally selected
because of the following:
–
–
–
The filter impedance at the higher dominant harmonic
will be inductive, eliminating the possibilities of resonances, as opposed to tuning to the 6th or 7th orders,
which could produce resonances between the filter and
the source impedance near the 5th harmonic frequency.
As the amplitude of the harmonics in the load current
decreases as the harmonic order increases, the voltage
that the active filter has to generate to make the 7th order
harmonic component to circulate across the hybrid filter
will be reduced.
The passive filter will provide a higher impedance at
higher and non-dominant harmonic orders (h > 7), reducing the voltage provided by the active filter to avoid
the derivation of these harmonics through the hybrid
branch.
IV. C ONTROL S TAGE
The global compensation objectives of the conditioner are as
follows:
– dPF correction for every firing angle in the controlled
rectifier.
– Harmonic reduction in the source current.
– Imbalance elimination in the source current.
The control stage is formed by collaborative control strategies to achieve these global compensation objectives and tracking techniques. Collaborative control strategies have been developed with capabilities to do the following:
– Split the compensation objectives between the two converters (Table I),
– Guarantee an efficient overall performance, avoiding
interferences between the two converters,
– Allow that one converter could help the other one in
anomalous situations, as overload.
As the resulting collaborative strategies are complex, special
attention has been paid to develop algorithms with a number of
measurements as minimum as possible.
A. Collaborative Control Strategies
If control strategies based on the load current detection are
employed, as was previously proposed in Section II, in threephase four-wire systems, three load currents (iLa , iLb , and iLc ),
six compensating currents (iHFa , iHFb , iHFc , iAFfa , iAFfb , and
iAFfc ), three phase-to-neutral PCC voltages (uPCCa , uPCCb ,
and uPCCc ), and the dc bus voltage (Udc ) need to be sensed.
This implies thirteen measurements to control the multiconverter conditioner. An improvement is proposed so that the
control strategies will be based on the load current estimation
to minimize the number of sensors.
Fig. 4. Block diagram of the hybrid multiconverter conditioner control strategies.
The multiconverter conditioner uses a sinusoidal source current (SSC) global control strategy. The collaboration between
the two converters is based on a frequency sharing operation:
the slow converter employs a selective harmonic compensation
control algorithm using harmonic load current extractors, while
the fast converter cooperates with it to comply with the global
compensation objectives. Also, the control is implemented so
that the converter responsible for the dc bus control loop can be
chosen. These collaborative control strategies assure that there
is no type of interaction between the two conditioners.
The control block diagram of the hybrid multiconverter conditioner is shown in Fig. 4. It is composed of two parts which
are explained in the following sections.
Extraction of the Reference Source Current: The block diagram for the extraction of the reference source current is
displayed in the upper signal path in Fig. 4. The reference
source current using an SSC strategy operating as PHC [21],
but based on the measurement of the source current in 0–d–q
coordinates, can be calculated as follows:
⎤
⎡
⎤
0
iS0,ref
⎦
⎣ iSd,ref ⎦ = K ⎣ u+
PCC1d
+
iSq,ref
uPCC1q
⎡
=
+
p+
S1 + Δp1(Udc)
2
u+
PCC1d
+
2
u+
PCC1q
⎡
0
⎤
⎣ u+
⎦
PCC1d
u+
PCC1q
(4)
where the term Δp+
1(Udc) is the positive-sequence fundamental
active power absorbed from the grid for controlling the dc bus
voltage. This term is obtained from the output of a proportionalintegral (PI) controller whose input is the error between the
reference dc bus voltage and its measurement. The positivesequence fundamental PCC voltage is obtained by an autoadjustable synchronous reference frame (ASRF) [28].
Calculation of the Reference Hybrid Conditioner Current:
The calculation of the reference current for the hybrid filter
proposed in (1) has to be changed because the load currents
are not measured. A new control method which estimates these
variables from the measurement of the source currents will be
used, resulting in the following reference current for the slow
converter:
∗
∗lim
+ i∗lim
(5)
iHF,ref = iL1 − i+
L5 + iL7
L1d
where the superfix “∗” means that this variable is not measured,
but estimated.
The harmonic terms in (5) are obtained using two blocks
called the hth harmonic load current extractor (hLCE), one for
the fifth and another for the seventh harmonic orders (see lower
signal path in Fig. 4). In Fig. 5, the block diagram of an hLCE
is displayed, where the inputs are the source current and the
positive-sequence harmonic angle (calculated by multiplying
the output angle of the ASRF, θ1+ , by the harmonic order), and
the output is the estimated harmonic load current. The principle
of operation of this module is the following: If a dominant
harmonic load current iLh is to be absorbed by the hybrid filter,
this component has to be null in the source current. Two synchronous reference frames (positive- and negative-sequence)
and one one-phase ASRF [28] (zero-sequence) are used to
Fig. 5. hLCE using control strategies based on the source current measurement.
Fig. 6. Fundamental compensating load current extraction (1LCE) using control strategies based on the source current measurement, with the addition of a dc
bus control loop.
calculate the source current components in 0–d–q coordinates.
A PI controller for each component is used to estimate this
component in the load current.
Also, to avoid the fifth passive filter overload, (2) is particularized as follows:
i∗lim
Lh =
i∗Lh ,
i∗
Lh
∗
ILh
h lim
· IPF5
,
∗
h lim
if ILh
IPF5
∗
h lim
if ILh
> IPF5
(6)
∗
where it must be highlighted that ILh
is the rms value of
the tuned harmonic component in the estimated load current;
h lim
is the maximum rms value of each tuned harmonic
and IFP5
current in the fifth passive filter.
It is necessary to add an additional block for the dc bus
voltage control in the fundamental harmonic term in (5). For
this voltage to be constant and near its reference, the hybrid
filter has to absorb fundamental positive-sequence active power
from the utility. Hence, a new term, Δi+
1d(Udc) , has to be
included, resulting finally in the following:
∗
∗lim
∗lim
+ Δi+
iHF,ref = iL1 − i+
L1d
1d(Udc) + iL5 + iL7 . (7)
This term is obtained from the output of a PI controller
whose input is the error between the reference dc bus voltage
and its measurement. The PI controller has been designed with
an enable signal to choose whether the slow conditioner is in
charge of the dc bus control (enabling the PI) or whether this
function is left to the fast conditioner (disabling the PI or not
turning on the slow converter). It also improves the reliability
of the multiconverter conditioner, since in case of failure of
one converter, the other one could continue operating. The
control block diagram for the fundamental component is shown
in Fig. 6.
In Fig. 4, one observes that seven measurements only are
needed for the control strategies (three source currents, the dc
bus voltage, and three PCC voltages).
Fig. 7. Duty cycle determination for the hybrid conditioner (slow).
Fig. 9. Experimental prototype: 1: Slow (hybrid) conditioner, 2: Fast conditioner, 3: Ferromagnetic core inductors (passive filter), 4: Capacitors (passive
filter), 5: Air core inductors, 6: DC bus capacitors.
TABLE II
PASSIVE F ILTER PARAMETERS
Fig. 8. Duty cycle determination for the active conditioner (fast).
B. Tracking Technique
The tracking technique determines the duty cycle for the conditioner aimed at eliminating the error between the reference
and the measured currents e in a switching period TS , namely:
D = 0.5 − e
uAF
LAF
−
TS Udc
Udc
(8)
where uAF is the voltage on the ac side of the active filter.
In the slow conditioner, the hybrid filter current is measured,
so e will be determined from the difference between the reference and the measured currents. The voltage uAF can be
calculated as follows:
uAF = uPCC + iHF,meas · ZCPF5
(9)
where ZCPF5 is the impedance of the passive filter without
inductance.
Fig. 7 shows the block diagram of the dead-beat technique
for the slow conditioner, in which TSs is its switching period.
As the fast conditioner current is not measured, a strategy
which allows the estimation of the error between the reference
and the measured currents is needed. If the hybrid filter is turned
on from the fast converter point of view, the set formed by the
load plus the slow conditioner behaves as a new load without
first, fifth, and seventh order components to correct. In a fastswitching period TSf , the new load current can be considered
approximately constant, so:
ΔiS + ΔiAFf = (ΔiL − ΔiHF ) ≈ 0 → ΔiS = −ΔiAFf
(10)
which means that the error can be calculated as the difference
between the reference and the measured source currents changing the sign. Fig. 8 shows the dead-beat technique for the fast
conditioner. In this converter, uAF is equal to the PCC voltage.
The tracking technique requires the measurement of three
additional signals, the hybrid filter currents, resulting finally
in ten variables necessary for the control stage of the novel
topology.
V. E XPERIMENTAL R ESULTS
The novel topology and control were tested on a 1.2-kVA laboratory prototype (Fig. 3). A three-phase four-wire system
√was
used, 50 Hz, and nominal base parameters UBL−L = 100 3 V
3φ
and SB
= 1200 VA, from which the base values of current,
IB = 4 A, and impedance, ZB = 25 Ω, can be obtained.
The nonlinear load is formed by a three-phase controlled
rectifier with resistive load, RL = 48 Ω, selected so that the
maximum power demanded, namely, the nominal power of the
load (SL max = 1170 VA), is lower than the basis power of
the system.
Fig. 9 shows a photograph of the experimental multiconverter
conditioner prototype, highlighting its components. The capacitance and inductance of the passive filter have been calculated
so that at the fundamental frequency (equivalent capacitive behavior) the reactive power equals the maximum reactive power
demand, QL1 max = 561.21 VAr (when the firing angle is
45.26◦ ), and at the fifth harmonic, the inductive and capacitive
reactances are equal. Afterwards, as the inductor of the passive
filter has to operate as a filter inductor in the active equipment, it
has been checked that the value of LPF5 was between the design
limits of LAFs , taking into account the double criteria, namely:
the ripple and maximum current derivatives. The quality factor
of the filter was selected so that the resistance of the filter was
lower than the source impedance at the tuned frequencies. The
values of the passive filter parameters are indicated in Table II,
while Table III gives the parameters used in the experimental
conditioner. The values of the capacitances C1 and C2 are
calculated assigning a maximum ripple in the dc voltage of 3%.
The control algorithms are implemented using a real-time
control system DS1104 (dSPACE), composed of a Power
TABLE III
PARAMETERS OF THE E XPERIMENTAL C ONDITIONER
Fig. 11. Operation of the multiconverter conditioner. From top to bottom:
waveforms of iL , iS , iAF , iHF (5 A/div), and frequency spectra of these
currents (400 mA/div; 125 Hz/div) in the same order. (a) α = 0◦ ; (b) α = 45◦ .
Fig. 10. Operation of the multiconverter conditioner. From top to bottom:
waveforms of iL , iS , iAF , iHF (5 A/div), and frequency spectra of these
currents (400 mA/div; 125 Hz/div) in the same order. (a) α = 0◦ ; (b) α = 45◦ .
PC603e/250 MHz processor and a Texas Instruments DSP
TMS320F240. This platform has four multiplexed A/D inputs
of 16 bits (2 μs sampling time) and four A/D inputs of 12 bits
(8 ns sampling time). With this control platform, we can carry
out experimental tests which only need eight inputs, (iSa , iSb ,
iSc , iHFa , iHFb , uPCCab , uPCCcb , and Udc ).
The control strategies and tracking techniques in detail are
provided in the previous two sections. For the generation of the
switching signals of the inverters, two symmetric pulse width
modulations were employed. These are generated by the slave
DSP of the DS1104 platform from the duty cycles calculated
in the tracking techniques. The switching frequency of the slow
and fast conditioners was fixed to 4 and 10 kHz, respectively.
The experimental results when the fast conditioner is turned
on first, so that it is responsible for the dc bus control (the
PI is disabled), are shown in Fig. 10 for different firing angle
values. In Fig. 10(a), there is no fundamental reactive power,
so the fundamental hybrid current component is null appearing
only the fifth and seventh compensating currents which fully
eliminate these components in the load current. The active
conditioner demands fundamental current for the dc bus control
and reduces the high-order components in the load current.
However, in Fig. 10(b), the firing angle of the rectifier is not
zero, so fundamental reactive power is demanded by the load.
This power term is delivered by the slow conditioner, so that the
hybrid filter current contains a fundamental component in this
situation.
In Fig. 11, similar experiments have been conducted, but
turning on the slow conditioner first, so that the dc bus control is
the task of this equipment (the PI is enabled). This is the reason
why the fundamental component appears in the hybrid conditioner although no fundamental reactive power is demanded by
the load [see Fig. 11(a)].
Total harmonic distortion (THD) values of the load and
source currents obtained in these experimental tests are summarized in Table IV. The results fulfill the standard IEEE519 (SCR < 50) except in Fig. 10(b) due to the ripple at
TABLE IV
THD VALUES OF C URRENTS IN THE E XPERIMENTAL T ESTS
Fig. 13. Single-phase equivalent circuits of the systems analyzed by simulation: (a) active monoconverter conditioning and (b) proposed hybrid multiconverter conditioning.
VI. D ISCUSSION
The hybrid multiconverter topology proposed in this paper
involves an increase in the number of active and passive elements and control algorithms with more complexity, so the
employment of this conditioner makes sense only if lower
inverter ratings are attained, allowing its use for high-power
applications. The design rate power of the inverter can be
calculated as
Sinv = 3Uinv Iinv
Fig. 12. Operation of the multiconverter conditioner under unbalanced PCC
voltages. From top to bottom: waveforms of the currents for phases a, b and c
(5 A/div) and frequency spectra of these currents (400 mA/div; 125 Hz/div) in
the same order. (a) iL and (b) iS .
4 kHz (switching frequency of the slow converter), which
cannot be compensated by the fast converter since its switching
frequency has had to be reduced to 10 kHz due to computational
limitations of the control platform.
To test the operation of the conditioner under unbalanced PCC voltage conditions, the phase c conductor is
connected to the neutral conductor of the utility, causing
+
−
/UPCC1
=
inverse- and zero-sequence components (UPCC1
+
0
UPCC1 /UPCC1 = 50%). The load currents for each phase are
shown in Fig. 12(a), while the three source currents are presented using the same axis in Fig. 12(b) so that balanced
and sinusoidal source currents can be appreciated. The source
current THD is improved to 2%, fulfilling the IEEE 519 limits.
(11)
where Uinv and Iinv are the rms values of the voltage and the
current in the ac side of the inverter, respectively. It has to be
noted that Uinv is not the voltage in the output of the active
equipment (after the filter inductor), but the voltage in the ac
terminals of the inverter.
A simulation analysis was carried out to compare the inverter
ratings of a conventional active monoconverter conditioner
injecting the global reference compensation current,
iAF,ref = iL − iS,ref
(12)
and a hybrid multiconverter conditioner with the reference
currents proposed in (1) for the slow conditioner and in (3) and
(4) for the fast one.
The single-phase equivalent circuits of the two systems under
comparison are shown in Fig. 13. One observes that the active
filters are displayed as voltage sources in phase with an inductor, which is equivalent to the current sources in Fig. 2. Also,
the inductor in the hybrid conditioner [see Fig. 13(b)] takes part
in both the active and the passive filters, since it was designed
for that purpose.
TABLE V
PARAMETERS U SED IN THE S IMULATION T EST
voltage in the ac side of the inverter (since the semiconductor
devices depend on this value), and this voltage is higher than
the PCC voltage in the active equipment (active monoconverter
and active multiconverter) due to the voltage drop in the filter
inductor.
TABLE VI
S IMULATION R ESULTS : M AXIMUM A PPARENT P OWER OF THE I NVERTER
IN THE L EAST FAVORABLE F IRING A NGLE S ITUATION R ELATIVE
TO THE M AXIMUM L OAD P OWER
VII. C ONCLUSION
Since at high-power levels phase-controlled rectifiers are
typically used in large adjustable speed drives, we selected
the same load proposed in the experimental tests, a controlled
rectifier supplying a resistive load, varying the firing angle of
the rectifier, so that variable fundamental reactive power compensation is needed. The passive filter in the hybrid conditioner
was also tuned to the fifth order. The values of the parameters
used in the simulation analysis, listed in Table V, are the same
as the ones collected in Tables II and III for the experimental
tests, since the system and load are equal. The filter inductance
of the monoconverter conditioner has been selected with the
same criterion explained for the multiconverter conditioner.
The study was performed under sinusoidal and balanced
source voltages. For each firing angle value between 0◦ and
60◦ , the inverter ratings were calculated using (11) and drawn
as a curve versus the firing angle. Since a variable load has
been selected, the inverter ratings depend on the specific load
conditions for each firing angle. Then, we determined the ratio
between the maximum apparent power of the inverter in the
least favorable firing angle situation and the maximum load
power, Sinv /SL . The load power is calculated as
SL = 3UPCC IL
(13)
being UPCC the rms value of the PCC voltage and IL the rms
value of the load current.
The results are summarized in Table VI. For a conventional
active conditioner, this ratio is greater than 90%. With a hybrid
multiconverter topology, the fast conditioner inverter ratio is
below 16% and approximately 20% for the slow conditioner
inverter. These results validate the usefulness of the topology
for high-power levels due to the decrease in the inverter rating.
The size of a classical stand-alone active filter is typically
considered about the value of the THD of the load current.
For example, for a load current of 40% THD, the active filter
is rated 40% of the power size of the load. This low value
of power ratio is due to the fact that the conditioner does not
compensate variable fundamental reactive power (for example,
in case of diode rectifier loads), and the rating of the active filter
is calculated with the voltage, not in the ac side of the inverter,
Uinv , but after the filter inductor (so the voltage equals UPCC ).
However, the rating of the inverter should be calculated with the
A novel hybrid multiconverter topology composed of a
hybrid conditioner in parallel with an active one has been
proposed in this paper. The main advantage of the topology is the decrease attained in the rating of the inverters. A
simulation analysis demonstrated the proper performance of
this topology at high-power levels due to the lower inverter
ratings obtained as compared with the classical shunt active
monoconverter topology. Collaborative control strategies were
developed to split the compensation objectives between the
two converters, to guarantee an efficient overall performance
avoiding interferences between the two converters and to allow
that one converter could help the other one in anomalous
situations. Control strategies and tracking techniques for the
global conditioner without the measurement of the load current
were presented. Some experiments have been conducted with
a 1.2-kVA laboratory prototype under different load conditions
and under unbalanced voltages.
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María Isabel Milanés-Montero (S’03–M’06) was
born in Badajoz, Spain, in 1974. She received the
M.Sc. degree in industrial engineering and the Ph.D.
degree from Universidad de Extremadura, Badajoz,
Spain, in 1997 and 2005, respectively.
In November 1998, she joined Universidad de Extremadura as an Assistant Professor. She is currently
with the Power Electrical and Electronic Systems
Research Group. Her major fields of interest include
power quality, active and hybrid power filters, renewable energy sources control, and electrical machine
drives.
Enrique Romero-Cadaval (S’03–M’05–SM’10)
was born in Villafranca de los Barros, Badajoz,
Spain, in 1968. He received the M.Sc. degree in
electronic industrial engineering from Escuela Técnica Superior de Ingeniería, Universidad Pontificia
de Comillas, Madrid, Spain, in 1992 and the Ph.D.
degree from Universidad de Extremadura, Badajoz,
Spain, in 2004.
He is a full Professor in power electronics at
Universidad de Extremadura. He is currently with the
Power Electrical and Electronic Systems Research
Group. His research interests are power electronics, power quality, electromagnetic inference, active power filters, and renewable energy sources control.
Fermín Barrero-González (M’95–SM’10) was
born in Puebla de la Reina, Badajoz, Spain, in 1959.
He received the M.Sc. degree in electrical engineering from Universidad Politécnica de Madrid, Madrid,
Spain, in 1984 and the Ph.D. degree from The Universidad Nacional de Educación a Distancia, Madrid,
in 1995.
He is a full Professor in electrical engineering at
Universidad de Extremadura, Badajoz. He is currently coordinating the Power Electrical and Electronic Systems Research Group. His research interest
areas are power electronics in the power system, flexible ac transmission
systems, active power filters, and electrical machine drives.
Dr. Barrero-González is a member of the IEEE IAS Industrial Static Converters Committee, European Working Group.