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C 2014 Wiley Periodicals, Inc.
V
AN EFFECTIVE THIRD HARMONIC
GENERATOR USING A LEFT-HANDED
NONLINEAR TRANSMISSON LINE
In Bok Kim,1 Kang Wook Kim,1 Hyoungsuk Yoo,2
Jonghoo Park,3 and Hongjoon Kim3
1
School of Electronics Engineering, Kyungpook National University
2
Department of Biomedical Engineering, School of Electrical
Engineering, University of Ulsan, Ulsan, Republic of Korea
3
Department of Electrical Engineering Kyungpook National
University, 1370 Sankyuk-dong, Buk-Gu, Daegu 702-701, Korea;
Corresponding author: hongjoon@gmail.com
1. INTRODUCTION
The modern communication systems require high-frequency signal sources with high stability and low noise. High-frequency
signal sources can be obtained by multiplying the low frequency
signal. Generally, the frequency multiplier uses either transistors
or diodes as a nonlinear element. In particular, varactor diodebased frequency multipliers, in general, generate very little noise
(phase- as well as amplitude-noise). The only noise source in
this type of multiplier is thermal noise, which comes from the
series resistance of the varactor.
In this article, we present an effective third harmonic generator using a left-handed nonlinear transmission line (LH NLTL),
which is composed of a series of varactors and shunt inductors.
LH media, first postulated by Veselago [1], are becoming an
exciting reality, particularly as they are being demonstrated in
resonant radio frequency and microwave circuits [2–4]. In [5],
the authors presented a theoretical investigation of the basic
nonlinear wave propagation phenomena in LH media, which is
based on the duality of the conventional NLTL, a LH NLTL
with anomalous dispersion. The authors in [5], theoretically
proved that the third harmonic generation can be very effective
in LH NLTL, but they did not show the actual experimental
results. In [6–8], the authors presented the experimental results
for the second harmonic generation in a short LH NLTL. However, they could not show the effective third harmonic generation result predicted by [5] due to the limited bandwidth of the
LH NLTL they fabricated. Compared to the previous work [5–
8], our work shows that the bandwidth of the LH NLTL can be
extended significantly with careful fabrication, and demonstrates
that the third harmonic generation is very effective with LH
NLTL. Compared to the earlier literature regarding the third
harmonic generation, our work shows a comparable performance
with a smaller size, wider bandwidth, and a simpler structure.
2. EXPERIMENTAL RESULT AND DISCUSSION
Figure 1(a) shows the proposed third harmonic generator using
six-section LH NLTLs. The circuit was realized on a Rogers
RO4003 board with er 5 3.8 and thickness h 5 0.5 mm. The fabricated harmonic generator circuit was 12 3 7 mm2 in size, except
for the connectors and extra space. A row of 0.25 3 0.25 mm2
copper pads separated by 0.2-mm gaps was formed on the surface
Received 24 July 2013
ABSTRACT: In this article, an effective third harmonic generator
using a left-handed nonlinear transmission line (LH NLTL) is presented. The proposed harmonic generator using the LH NLTL is
composed of a series of varactor diodes and shunt inductances, which
were implemented with a 0.2-mm wide microstrip line. The harmonic
generator was built on a Rogers RO4003 substrate and has dimensions
of 12 3 7 mm2. The fabricated LH NLTL provides a maximum insertion loss of 5.2 dB, and a minimum insertion loss of 2.2 dB for a wide
frequency band from 2.3 to 18.7 GHz. The measured minimum third
harmonic conversion loss was 14.2 dB at an input frequency of 2.5
GHz. When compared to other works, a better third harmonic conversion efficiency is achieved with much smaller size in lower frequencies.
C 2014 Wiley Periodicals, Inc. Microwave Opt Technol Lett 56:568–
V
570, 2014; View this article online at wileyonlinelibrary.com. DOI
10.1002/mop.28146
Key words: third harmonic generator; left-handed nonlinear transmission line; varactor diodes
568
Figure 1 (a) The fabricated third harmonic generator using LH NLTL.
(b) The equivalent circuit
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 56, No. 3, March 2014
DOI 10.1002/mop
Figure 2 The measured insertion loss of the fabricated LH NLTL
Figure 3 Output power (first, second, and third harmonics) versus
input frequency. Each input frequency had 25 dBm of input power
of the Rogers board. The 12 varactor diodes (M/A-COM
MA46H120 hyperabrupt junction GaAs flip-chip) were attached
between these pads. The Cj0 of the diodes, from the manufacturer’s data sheet, is 1.1 pF, and they show a capacitance ratio of
C(0V)/C(10V) 5 7.5. The nonlinear capacitance in each section is
formed by two antiseries diodes. Shunt inductances were implemented with a 0.2-mm wide microstrip line connecting the pads
to the ground plane on the back of the board. The length of these
inductance lines was 4.5 mm. The pads on the board surface,
together with an inherent parasitic effect, introduce unavoidable
series inductance and shunt capacitance, making the whole circuit
a composite right-handed/LH transmission line having the equivalent circuit shown in Figure 1(b).
Figure 2 shows the insertion loss of the fabricated LH
NLTL. The high-pass type frequency response with a Bragg cutoff frequency indicates the LH nature of the device. The frequency range from 2.3 to 18.7 GHz corresponds to the LH
passband. Parameters of the circuit model in Figure 1(b) were
extracted from the S parameters simulated with an Ansoft circuit
simulator: CL 5 1 pF, LL 5 2.1 nH, CR 5 0.4 pF, LR 5 0.9 nH,
Rd 5 2 ohm. To enhance the bandwidth of the LH region and
minimize the parasitic effect, various simulations and experiments were conducted. Also, the number of LH NLTL sections
were carefully selected through simulations to maximize device
nonlinearity.
Figure 3 shows the output power versus the input frequency
when 25 dBm of input power for the frequency range from 2.1
to 3.4 GHz. In the frequency range, the third harmonic conversion loss ranges from 14.2 to 21.3 dB at 25 dBm of input
power. The minimum third harmonic conversion loss was 14.2
dB when the input frequency was 2.5 GHz. At this frequency,
the third harmonic output power was 8 dB higher than the second harmonic and 10 dB higher than the first harmonic. This
means the input frequency component attenuates significantly
due to the high-pass-filter response of the LH NLTL, and the
third harmonic is more dominant than the second harmonic in
the LH NLTL structure. Also, as seen by the magnitude
response as shown in Figure 2, good conversion efficiency was
achieved when the input frequency was close to the Bragg cutoff frequency. Table 1 compares the performance of the proposed third harmonic generator with those in other papers. It
can be seen that this LH NLTL-based third harmonic generator
has wider frequency bandwidth and is smaller in size than other
third harmonic generators reported in the literature. If the harmonic generator is designed with a high dielectric-constant or
fabricated in monolithic microwave integrated circuit (MMIC),
the size can be smaller. And, if the harmonic generator is added
filter in order to suppress the other harmonic except third harmonic, it is used for effective tripler.
3. CONCLUSION
In this article, we present an effective third harmonic generator
using a LH NLTL. The proposed harmonic generator shows a
broader frequency bandwidth in a smaller size with comparable
performance compared with other reported research results. If
the varactor diode is optimized and the parasitic elements are
minimized by using an MMIC fabrication process, a better result
can be obtained. Also, by scaling down the suggested circuit, it
is possible to achieve a harmonic generator at a much higher
TABLE 1 A Performance Comparison of Harmonic Generator
Title
Frequency
Band (GHz)
Structure
[9]
LPF 1 input matching 1 HBV 1 output matching 1 BPF
[10]
APDP (antiparallel pair diode) 1 BPF
[11]
LPF 1 phase shifters 1 Schottky diode 1 BPF
This work
Varactor diode 1 stub
DOI 10.1002/mop
13.1–13.9 (fo)
Pin 5 22 dBm
5.71–6.28 (fo)
Pin 5 22 dBm
7–9.5 (fo)
Pin 5 22 dBm
2.1–3.4 (fo)
Pin 5 25 dBm
Conversion
Loss (dB)
Size (cm)
11.2–17
2 3 1 Alumina
16.6–18.5
5 3 2 RT5880
17–20
5 3 2 RO4003
14.2–21.3
1.2 3 0.7 RO4003
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 56, No. 3, March 2014
569
frequency. Because of its compactness and simple structure,
there may be many applications may exist in microwave (even
terahertz) systems.
ACKNOWLEDGMENT
This research was supported by Basic Science Research Program through the National Research Foundation of Korea (NRF)
funded by the Ministry of Education, Science and Technology
(2012R1A1A1041888).
notched function is achieved to reject worldwide interoperability for
microwave access, WLAN, and X-band signals. The multimode reconfigurable band-notched characteristic is realized by integrating four
switches onto the TS-TSIR and the PSLR. Simulated and measured
results show that this antenna, with multimode reconfigurable characteristics and dual band-notched function, can operate from 2.7 to 12 GHz
(VSWR<2), which includes the entire UWB band. This antenna maintains omnidirectional radiation patterns that is suitable for UWB appliC 2014 Wiley Periodicals, Inc. Microwave Opt Technol Lett
cations. V
56:570–574, 2014; View this article online at wileyonlinelibrary.com.
DOI 10.1002/mop.28152
REFERENCES
1. V.G. Veselago, The electrodynamics of substances with simultaneously negative values of e and l, Sov Phys Usp 10 (1968), 509–514.
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10. J. Min, S. Cho, H. Kim, Y. Kim, K. Lee, D. Kim, H. Yune, U.
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based on microstrip line circuits and schottky diodes, In: 2010 AsiaPacific Microwave Conference Proceedings (APMC), Yokohama,
Japan, December 7–10, 2010, pp.164–167.
C 2014 Wiley Periodicals, Inc.
V
A COMPACT CIRCULAR SLOT UWB
ANTENNA WITH MULTIMODE
RECONFIGURABLE BAND-NOTCHED
CHARACTERISTICS USING RESONATOR
AND SWITCH TECHNIQUES
Yingsong Li,1 Wenxing Li,1 and Qiubo Ye2
1
Department of Information and Communications Engineering,
Harbin Engineering University, Harbin, Heilongjiang 150001, China;
Corresponding author: liwenxing@hrbeu.edu.cn
2
Communications Research Centre, 3701 Carling Avenue, Ottawa,
Canada K2H 8S2
Received 21 June 2013
ABSTRACT: In this article, we propose a circular slot ultrawideband
(UWB) antenna with multimode reconfigurable dual band-notched characteristics. By using a two-stage T-shaped stepped impedance resonator
(TS-TSIR) and a parallel stubs loaded resonator (PSLR), dual band-
570
Key words: ultrawideband antenna; dual band notch antenna; reconfigurable antenna; T-shaped stepped impedance resonator; parallel stub
loaded resonator
1. INTRODUCTION
Since the Federal Communication Commission (FCC) allocated
the frequency band 3.1–10.6 GHz for commercial ultrawideband
(UWB) systems, UWB radio techniques have been widely studied both in industry and academia [1–4]. As one of the key
components of the UWB systems, antennas with wide bandwidth, compact size, and omnidirectional radiations, have been a
hot topic and have attracted much attention in recent years. A
number of UWB antennas have been reported and investigated
for meeting these purposes mentioned above [2–4]. However,
there are some existing narrow-band systems over the designated UWB frequency band, such as worldwide interoperability
for microwave access (WiMAX) operating at 3.4–3.69 GHz,
WLAN operating at 5.15–5.35 GHz, and X-band satellite communication systems working at 7.7–8.5 GHz, which may interfere with the UWB systems. To mitigate the potential
interferences, a lot of UWB antennas integrated with single or
dual band-notched characteristics have been reported and are
well designed [5–11]. Most of these band-notched characteristics
are realized by etching various slots either in the radiation patch
or in the ground plane. The etched slots will leak electromagnetic wave, which deteriorates the radiation patterns. To circumvent above shortcomings, some band-notched UWB antennas
are proposed by using integration with parasitic strips [9] and
stubs [10]. Although these UWB antennas can achieve good performance, they can only be used as band-notched antennas. To
remedy this defect, designing a band-notched UWB antenna
with reconfigurable function is desirable and attractive.
Recently, several reconfigurable band-notched UWB antennas
are studied [11, 12], but most of them have either one reconfigurable notch band or one reconfigurable mode [11].
In this article, an effective method is utilized to design a
miniaturized UWB antenna with multimode reconfigurable dual
band-notched characteristics, which has six reconfigurable
modes and can have two notch bands either both at WiMAX
band and X-band or both at WLAN band and X-band. By integrating a two-stage T-shaped stepped impedance resonator (TSTSIR) inside the circular ring radiation patch and by etching a
parallel stubs loaded resonator (PSLR) into the CPW transmission line, two notch bands are achieved. To realize the multimode reconfigurability, four ideal switches are integrated into
the TS-TSIR and the PSLR. By controlling the states of the
switches, the proposed antenna can be used either as a dual
band-notched UWB antenna, or as a single band-notched UWB
antenna, even as a UWB antenna. To verify the design, two
antennas with all the switches ON or OFF are fabricated and
measured. In this study, the presence of a metal bridge
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 56, No. 3, March 2014
DOI 10.1002/mop