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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006
Analysis and Design of a Low-Stress Buck-Boost
Converter in Universal-Input PFC Applications
Jingquan Chen, Member, IEEE, Dragan Maksimović, Member, IEEE, and Robert W. Erickson, Fellow, IEEE
Abstract—In converters for power-factor-correction (PFC), the
universal-input capability, i.e., the ability to operate from any
ac line voltage world-wide, comes with a heavy penalty in terms
of component stresses and losses, and with restrictions on the
dc output voltage. In this paper, we propose a new two-switch
topology, boost-interleaved buck-boost (BoIBB) converter, which
can offer significant performance improvements over single-switch
buck-boost converters (including flyback, SEPIC, or Cuk topologies) or other two-switch buck-boost converters in universal-input
PFC applications. The paper presents an analysis of the converter
operation and component stresses, as well as design guidelines.
High efficiency (over 93%) throughout the universal-input ac line
voltage range is demonstrated on an experimental 100-W, 200-V
dc output, universal-input BOIBB PFC rectifier.
Index Terms—Boost-interleaved buck-boost (BoIBB) converter,
power-factor-correction (PFC), root-mean-square (RMS).
I. INTRODUCTION
I
T is well known that the boost converter topology is highly
effective in power factor correction (PFC) rectifier applications, provided that the dc output voltage is close to, but slightly
greater than, the peak ac line voltage [1]. In universal-input applications, with the root-mean-square (RMS) input line voltage
in the 90–260 V range, the output voltage of the boost converter
has to be set to about 400 V. At low line (90 V ), the switch
conduction losses are high because the input RMS current has
the largest value, and because the largest step-up conversion is
required. The inductor has to be oversized for the large RMS
current at low line input, and for the highest volt–seconds applied throughout the input-line range. As a result, a boost converter designed for universal-input PFC applications is heavily
oversized compared to a converter designed for a narrow range
of input ac line voltages. Furthermore, because of the large energy storage filter capacitor at the output, the boost converter
has the inrush current problem that can only be mitigated using
additional components.
In universal-input PFC applications, the capability of providing both step-up and step-down conversion is attractive because the output dc voltage can be set to any value. However,
conventional single-switch buck-boost topologies, including the
buck-boost, flyback, SEPIC, and Cuk converters, have greatly
Manuscript received July 25, 2002; revised October 10, 2005. This paper was
presented in part at IEEE APEC’01. Recommended for publishing by Associate
Editor Y.-F. Liu.
J. Chen is with the FyreStorm Inc., Sunnyvale, CA 94086 USA (e-mail:
jingquan.chen@ieee.org).
D. Maksimović and R. W. Erickson are with the Colorado Power Electronics
Center, Department of Electrical and Computer Engineering, University of Colorado, Boulder, CO 80309-0425 USA.
Digital Object Identifier 10.1109/TPEL.2005.869744
increased component stresses and component sizes compared
to the boost converter [2]–[6].
In general, if their conversion characteristics meet the
input/output specifications, the boost converter (for voltage
step-up) or the buck converter (for voltage step-down) feature
the smallest component stresses. This is a result of the direct
energy transfer path from the input to the output in one of the
switching subintervals in these two converter topologies. The
boost and the buck converters require the minimum indirect
energy delivery and therefore have the minimum component
stresses for a given voltage conversion ratio. Based on this
observation, it is of interest to investigate buck-boost converter
topologies with two independently controllabe switches that
can operate as boost (for voltage step-up) or as buck (for voltage
step-down) converters during portions of an ac line cycle.
Fig. 1 shows (a) the standard buck-boost converter and the
two well-known two-switch buck-boost configurations, (b)
the buck-cascaded buck-boost (BuCBB) converter, and (c) the
boost-cascaded buck-boost (BoCBB) converter. When the two
active switches are operated independently, the two-switch
converters can achieve buck or boost operation with minimum
indirect energy processing. As a result, at the expense of
additional switches and controls, the two-switch buck-boost
topologies can offer reduced component stresses.
It has been shown that a number of other two-switch
buck-boost configurations can be constructed. In [7], complete families of two-switch buck-boost converters that can
achieve minimum indirect energy delivery have been generated
through the synthesis method based on the equivalent ac and
dc circuits [8]. A comparison of the two-switch buck-boost
converter topologies in terms of the switch and the inductor
stresses can be found in [7]. In particular, the boost-interleaved
buck-boost (BoIBB) converter shown in Fig. 2 has been identified as a configuration with potentials for significantly smaller
switch stresses compared to the more conventional cascaded
buck-boost converters of Fig. 1(b) and (c), and with lower
conduction losses and reduced inductor stresses compared to
the boost converter. In power factor correction applications,
further advantages of this configuration include the ability to
choose the output dc voltage arbitrarily, and the absence of the
inrush current problem.
The purpose of this paper is to present a detailed analysis of
the operation and the component stresses in the BoIBB converter, as well as design guidelines and practical implementation techniques in universal-input PFC applications. Operating
modes and basic steady-state characteristics of this converter
are described in Section II. Operation of the BoIBB converter
as a PFC rectifier together with an analysis of the switch and
0885-8993/$20.00 © 2006 IEEE
CHEN et al.: ANALYSIS AND DESIGN OF A LOW-STRESS BUCK-BOOST CONVERTER
Fig. 1.
(a) Standard buck-boost converter, (b) the buck-cascaded buck-boost (BuCBB) converter, and (c) the boost-cascaded buck-boost (BoCBB) converter.
Fig. 2.
Boost-interleaved buck-boost (BoIBB) converter.
321
inductor stresses and conduction losses are discussed in Section III, in comparison with the converters shown in Fig. 1.
Section IV describes an experimental 100-W, 200-V dc output,
universal-input BoIBB PFC rectifier, with experimental results
shown over the universal-input ac line voltage range.
II. OPERATING MODES AND STEADY-STATE CHARACTERISTICS
OF THE BOOST-INTERLEAVED BUCK-BOOST CONVERTER
The proposed boost-interleaved buck-boost (BoIBB) conis shown in
verter with two controllable switches
Fig. 2. In contrast to the cascaded topologies, such as the
converters of Fig. 1(b) and (c), where the buck and the boost
converter are simply connected in series, in the BoIBB converter
and
) is effectively interleaved
the boost switch cell (
and
).
with the buck switch cell (
Let
and
be the duty ratios of the switches
and
, respectively. In continuous conduction mode (CCM), the
volt–second balance relations for the two inductors yield the
for the BoIBB
overall dc voltage conversion ratio
converter
(1)
Fig. 3. Operating modes of the BoIBB converter: (a) boost and (b) buck.
If
is always on,
1,
1 1
, and the converter
operates in the boost mode, which is shown in Fig. 3(a). The
is zero. In this mode, the input current
average voltage across
and .
is divided through
If
is always off,
0,
, the converter operates
and
form a lowin the buck mode, as shown in Fig. 3(b).
and
is zero
frequency filter. The average current through
is equal to the difference between the
and the voltage across
input and the output voltage. The inductor
in the buck mode
has the same role as the inductor in the simple buck converter.
The basic steady-state results for the two modes of operation are
summarized in Table I.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006
TABLE I
STEADY-STATE RESULTS FOR THE BOIBB CONVERTER
IN THE BOOST AND THE BUCK MODES OF OPERATION
III. OPERATION OF THE BOIBB CONVERTER
AS AN IDEAL RECTIFIER
In this section, we analyze operation of the BoIBB converter
as an ideal PFC rectifier. Expressions for the switch and the inductor RMS currents, and the volt-seconds applied to the inductors are derived, so that conduction losses and magnetic sizes
can be evaluated. We also compare the total switch and inductor
RMS currents of the BoIBB converter with the three converters
shown in Fig. 1.
In PFC rectifier applications, the rectified input voltage is
(2)
In an ideal PFC rectifier, the output voltage is regulated at a
is proportional
constant value , and that the input current
to the input voltage
Fig. 4. (a) Waveforms of the rectified input voltage v (t) and the dc output
voltage V and (b) duty ratios of the boost and the buck cells in the BoIBB
converter operated as an ideal PFC rectifier.
Therefore, we can assume quasi steady-state operation, which
means that the switch duty ratios as functions of time can be
found from the steady-state dc conversion results in Section II
(3)
(4)
where the emulated resistance
is constant for a given output
power.
Fig. 4(a) shows the waveforms of the input and the output
voltage in one half of a line period, for the case when the dc
output voltage is chosen to be lower than the peak of the input
voltage. The converter operates in the boost or the buck modes
according to the condition of the input voltage and the output dc
voltage, as shown in Fig. 4(b). In Section V, we show that the
switchover between the boost and buck modes can be accomplished automatically using a relatively simple PWM controller,
without the need to compare the input and the output voltage to
facilitate the mode switching.
Since the input voltage waveform is periodic with the period
2 (half line cycle) and symmetric with respect to
4,
of
4]. Opthe analysis can be restricted to the time interval [0
eration in continuous conduction mode (CCM) is assumed.
The average inductor currents in a switching period are
(5)
When
is conducting, its current is the sum of the two inconducts the same current as .
ductor currents, while
is always off, and the current through
In the buck mode,
equals a small current ripple. Therefore, the total RMS curand
can be found from (4) and (5), and (6) and
rents of
(7), shown at the bottom of the next page. The volt–seconds apand
during a switching period are the same as the
plied to
volt-seconds applied to the inductor in a simple boost converter,
and are given by
A. Analysis of Stresses
1) Boost Mode: In the time interval 0
, as shown in
Fig. 4, the input voltage is lower than the output voltage, and
the converter operates in the boost mode: the boost switch cell
is active, while the buck cell
is inactive (
is always on). In the case when the output voltage is higher
than the peak of the input voltage, the converter operates in the
boost mode always. In this case, the results of this section still
with
4. The converter
apply. One only needs to replace
switching frequency is much higher than the ac line frequency.
(8)
where
is the switching period.
4], where
2) Buck Mode: In the time interval [
is the line period, the instantaneous input voltage is greater than
the output voltage, and the converter operates in the buck mode:
the buck cell is active and the boost cell is inactive ( is always
and
form a low frequency filter between the input
off).
CHEN et al.: ANALYSIS AND DESIGN OF A LOW-STRESS BUCK-BOOST CONVERTER
and the output, and have insignificant effects in quasi steadystate operation.
and
can be expressed as:
The duty ratios of
323
TABLE II
COMPARISON OF COMPONENT RMS CURRENTS AT LOW LINE (120 VRMS) AND
HIGH LINE (240 VRMS) FOR A PFC RECTIFIER WITH THE DC OUTPUT
200 V AND THE OUTPUT POWER P
100 W
VOLTAGE V
=
=
(9)
The average inductor currents in a switching period are
(10)
Both
and
are conducting currents in both boost and buck
modes, and the RMS currents are found from (4), (5), and (10)
as (11) and (12), shown at the bottom of the page.
during a switching period are
The volt–seconds applied to
the same as the volt-seconds applied to the inductor in the buck
converter
(13)
The volt–seconds applied to
buck mode.
are approximately zero in the
B. Comparison of Stresses
It is of interest to compare the switch and inductor RMS currents of the BoIBB converter to the three converters of Fig. 1, in
PFC applications. We assume that the two-switch BuCBB and
BoCBB converters are operated in the same way as the BoIBB
and
duty cycles shown in
converter, with the switch
Fig. 4. In the BoCBB converter of Fig. 1(c), the energy storage is
at the output. Equations (14)–(16), shown at the bottom of the
next page, summarize the results for the standard buck-boost
converter of Fig. 1(a), the BuCBB converter of Fig. 1(b), and
the BoCBB converter of Fig. 1(c), respectively.
As an example, Table II compares the component RMS currents in a 100-W BoIBB PFC rectifier with the dc output voltage
200 V, to the RMS currents in the three converters of Fig. 1
for the low ac line input (120 Vrms) and the high ac line input
(240 Vrms).
At the low-line input, in this example, the two-switch buckboost converters always operate in the boost mode. According to
(6)
(7)
(11)
(12)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006
(6), (15) and (16), the RMS current of the switch
in this mode
is the same for all three two-switch configurations. The RMS
, which always stays on in the boost
currents for the switch
mode, are different. In the BoIBB and the BoCBB, the switch
conducts the inductor current
. In the BuCBB,
conducts the inductor current , which is significantly higher at the
rms currents in the BoCBB,
low-line input. At low line, the
which is equal to the dc load current, is somewhat lower than in
the BoIBB due to the filtering action of .
For the high-line input, the three two-switch buck-boost converters operate in either the boost mode or the buck mode during
different portions of a line cycle. In the buck mode, according
has the same RMS current
to (11), (15), and (16), the switch
currents in
in all three two-switch converters. The switch
the boost mode are different, with the BoIBB having the lowest
RMS current stress at high line.
In the boost mode, the two inductors in the BoIBB share
the input current, which is not the case in other two-switch
conbuck-boost converters. Furthermore, in the buck mode,
ducts zero dc current in the BoIBB converter. As a result, in the
example of Table II, the BoIBB has the lowest total RMS inductor current.
In the example of Table II, it can be observed that the component RMS current stresses in the two-switch converters are
significantly smaller than in the standard buck-boost converter.
However, the standard buck-boost converter has only one inductor and only one transistor switch. Let us assume that the
on-resistance of the transistor switch in the standard buck-boost
and
converter is two times lower that the on-resistance of
in the two-switch buck-boost converters. Similarly, let us
assume that the inductor series resistance in the standard buckboost and the BuCBB is two times smaller than the series reand
in the BoCBB or BoIBB. Under these
sistances of
assumptions, Fig. 5 shows the switch and inductor conduction
losses normalized to the losses in the standard buck-boost, as
. At high line, i.e., at low
functions of the conversion ratio
, the BoIBB switch conduction losses are the lowest of the
three two-switch configurations. At low line, i.e., for
, the switch conduction losses in the BoIBB are somewhat
higher than in the BoCBB, but lower than in the standard buckboost. It should be noted that the switch voltage stresses in the
two-switch converters are lower than in the standard buck-boost,
resulting in an even more favorable comparison in terms of the
transistor switch utilization in the BoIBB and the BoCBB compared to the standard buck-boost converter. The BuCBB converter has higher RMS current stresses because the buck switch
has to conduct the input current in both buck and boost
modes.
The inductor conduction losses in the BoIBB and the BuCBB
are substantially smaller than in the standard buck-boost both at
low line and at high line. Furthermore, volt–seconds applied to
the inductors are significantly smaller than in the conventional
buck-boost converter, leading to reduced size of the magnetics.
Notice, however, that the BuCBB advantages in terms of the inductor utilization are offset in part by the penalties in the switch
conduction losses.
In conclusion, the BoIBB converter features the most favorable results for the switch conduction losses and the inductor
RMS current stresses over the universal-input voltage range.
IV. DESIGN CONSIDERATIONS
In this section, we address some of the design considerations
related to the selection of , , , and the dc output voltage
in the BoIBB converter.
(14)
(15)
(16)
CHEN et al.: ANALYSIS AND DESIGN OF A LOW-STRESS BUCK-BOOST CONVERTER
325
Combining (17)–(19) and using the results for
yields the CCM condition
from (4)
(20)
Defining
tion in the boost mode becomes
, the CCM condi-
(21)
In order to operate in CCM throughout the line cycle, and under
all input voltages and output power levels, from (21) we have
(22)
is the emulated input resistance for the largest ac
where
input voltage, and the lowest load power.
inductor cur2) Buck Mode: In the buck mode, only the
rent ripple is relevant for operation in CCM. The CCM condition
can be derived from
(23)
where
(24)
which yields
(25)
The right-hand side expression in (25) has a maximum that depends on the ratio of the peak input voltage
and the output
1.5 V, we have
voltage . For
Fig. 5. Comparison of (a) the total switch conduction losses in the
+
two-switch converters normalized to the buck-boost converter, (I
I
)=(I
=2) and (b) the inductor conduction losses in the
two-switch converters normalized to the buck-boost converter, (I
+
)=(I
=2).
I
(26)
and for
1.5 V, we have
(27)
A. Conditions for CCM Operation
The inductor current ripples are affected by the choice of
and
. In this section we derive conditions for operation in
continuous conduction mode (CCM).
is
1) Boost Mode: In the boost mode, when the diode
and
. Therefore, the CCM
on, it conducts the sum of
condition can be written as
(17)
where
ples
and
are the peak inductor current rip-
From (25)–(27), the conditions for CCM operation throughout
the line cycle are
if
if
.
(28)
The results of this section can be used to select the inductance
values to achieve CCM operation over the entire line cycle, or
over a desired portion of the line cycle, and over desired ranges
of input voltages and loads.
B. Selection of
(18)
(19)
We select the capacitance
so that the input voltage variations do not affect the output voltage or the input current through
path when the converter operates in the buck mode,
the ,
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006
i.e., so that the natural resonant frequency of
above twice the line frequency
and
is well
(29)
also affects the capacitor voltage ripple in the
The value of
boost mode. The peak-to-peak capacitor voltage ripple can be
found as
(30)
where is the output power. The ripple in (30) has the maximum value
when
when
.
(31)
The worst case is at the maximum power and the minimum input
line voltage. Finally, it should be noted that there is no need
to act as a low-frequency energy storage element. As a
for
result, the value selected according to (29) or (30), or according
to the capacitor RMS current stress, is typically much smaller
compared to the energy-storage capacitance .
Fig. 6. Experimental BoIBB rectifier (L
2.25 F, C
150 F, f
100 KHz, V
=
=
= 4 mH, L = 4 mH, C =
= 200 V).
C. Selection of the dc Output Voltage
Compared to a boost PFC rectifier, an advantage of the BoIBB
converter is that the output dc voltage
does not have to be
. For universal-input opgreater than the peak ac line voltage
is in the range from approximately 125 V to approxeration,
imately 370 V. If, for example, we select the dc output voltage
200 V, the ratio
varies from 0.54 at high line to 1.6
at low line. In this range, according to the results of Fig. 5, the
component stresses in the BoIBB are significantly lower compared to the standard buck-boost converter. However, for PFC
0.5)
applications where a low dc output voltage (e.g.,
is required, the BoIBB converter may not be the best solution,
as shown by the stress analysis and the comparison in Fig. 5.
V. EXPERIMENTAL RESULTS
An experimental prototype (Fig. 6) has been built to verify
feasibility and performance of the BoIBB converter. In our ex4 mH,
4 mH,
2.25 F,
perimental setup,
150 F, and the switching frequency is
100 KHz.
200 V.
and
are
The dc output voltage is set to
International Rectifier IRF840 [500-V, 8-A metal-oxide semiand
conductor field-effect transistors (MOSFETs)] and
are Philips Semiconductor BYM26C (600-V, 2.4-A ultra-fast
diodes).
A. Controller Implementation
In the experimental prototype shown in Fig. 6, the input
current shaping and the output voltage regulation are achieved
using a standard average current-mode PFC controller chip
(UC3854 [9]). The input voltage, the input current and the
output voltage are sensed and scaled to the proper levels following the usual practices with the UC3854 average current
mode controller. To achieve proper operation of the switches
and
in the boost and the buck modes, the pulse-width
Fig. 7. Dual PWM generator using TL1451.
modulator (PWM) in UC3854 is not used. Instead, the output
of the average current-mode compensator on the
UC3854 chip is fed to a dual PWM chip (TL1451 [10]) to
produce the switch control signals with the duty ratios
and
. These signals are then sent to a high and low side
gate driver (IR2110 [11]) to control the high-side buck switch
and the low-side boost switch
.
Fig. 7 shows a diagram of the TL1451 dual PWM generator
, which is generated by the avcontrolled by the voltage
erage current-loop compensator. TL1451 uses two dead-time
comparators, such that it is possible to generate two independently controlled PWM signals. The two comparators compare
and
to the same triangle modulation
the control inputs
signal
. As a result, the two PWM signals are synchrohas a valley of
nized. The triangle waveform
1.45 V and a peak of
2.05 V. The duty cycle of the
output PWM signal can be expressed as
(32)
where
600 mV, and
is
or
. The two amplifiers, which are included in the TL1451 chip
CHEN et al.: ANALYSIS AND DESIGN OF A LOW-STRESS BUCK-BOOST CONVERTER
Fig. 8.
Output duty ratios as functions of v
327
(t).
for the purpose of constructing voltage-loop compensators, are
to achieve
used for level-shifting of the control signal
the desired duty-cycle modulation according to Fig. 4
Fig. 9. v
a line cycle.
(t) (solid curve) and v (t) (dashed curve) during one half of
(33)
We assign PWM channel 1 to control the boost switch
and
. The buck/boost mode transichannel 2 to the buck switch
tion occurs when the boost switch is completely off while the
and
buck switch is fully on, or when
. In order to have a smooth mode transition, these two
conditions should occur at the same time, which can be accomplished by designing the circuit according to the following:
(34)
(35)
Another consideration in the selection of the two offset voltfalls into the proper output range
ages is to ensure that
of UC3854. If we define
(36)
or equivalently
(37)
and
as functions of the control
the output duty cycles
can be expressed as
voltage
(38)
and
(39)
The results in (38) and (39) are plotted in Fig. 8.
(solid curve) as a
In quasi steady-state operation,
function of time is shown in Fig. 9 for one half of an ac line
cycle. Notice that the BoIBB converter operates in the boost
and in the buck mode when
mode when
. The transition between the boost mode and
the buck mode occurs at
. The experimental
Fig. 10. Rectified input voltage v (t) (top) and the ac portion of the control
(t) (bottom): (a) low-line input (120 Vrms) and (b) high-line
voltage v
input (240 Vrms). Ch2: 100 V/div, Ch3: 500 mV/div.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006
Fig. 12. Efficiency of the experimental BoIBB PFC rectifier as a function of
the input line voltage.
The efficiency is 93.8% and the total current harmonic distortion
is 4.6%.
Fig. 12 shows the rectifier efficiency as a function of the input
line RMS voltage. In contrast to conventional PFC rectifiers
where the efficiency is significantly lower at low line, the efficiency remains greater than 93% throughout the line voltage
range (90 Vrms–264 Vrms).
VI. CONCLUSION
A new two-switch topology, named boost-interleaved
buck-boost (BoIBB) converter, is proposed for universal-input
PFC applications. A comparison with conventional buck-boost
or two-switch buck-boost converters shows that the BoIBB
converter has advantages of lower switch voltage stresses,
potentials for lower switch and inductor conduction losses, and
reduced size of the magnetics.
Experimental results are provided to verify the validity of the
new topology. High efficiency (over 93% throughout the whole
ac line voltage range), and low current harmonic distortion at
both high and low line inputs are demonstrated.
REFERENCES
Fig. 11. Rectified input voltage v (t) (top) and the input ac line current i (t):
(a) 120-V low-line input and (b) 240-V high-line input. Ch2: 100 V/div, Ch4:
0.5 A/div.
results of
are reported in Fig. 10, which correspond
well to the ideal waveforms in Fig. 9.
B. Experimental Results
Experimental waveforms are shown in Fig. 11. The output
power is 100 W. In Fig. 11(a), the input line voltage has a lowline RMS value of 120 Vrms and the converter operates in the
boost mode always. The efficiency is 93.7% and the total current
harmonic distortion is 1.9%. The waveforms of Fig. 11(b) are
for the high-line input (240 Vrms). The converter operates in
the boost or the buck modes in different parts of the line period.
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CHEN et al.: ANALYSIS AND DESIGN OF A LOW-STRESS BUCK-BOOST CONVERTER
Jingquan Chen (M’02) received the B.S. degree in
electrical engineering from Tsinghua University, Beijing, China, in 1995, the M.S. degree in power electronics from the Chinese Academy of Sciences, Beijing, in 1998, and the Ph.D. degree in electrical engineering from the University of Colorado, Boulder, in
2002.
From 2002 to 2003, he was a Senior Member of
Research Staff at Philips Research, Briarcliff Manor,
NY. Since 2003, he has been with FyreStorm, Inc.,
Sunnyvale, CA, where he is currently a Principal Engineer in the Advanced Research and Development Group. His current research
interests include synthesis, modeling, and the digital control of switching power
converters.
Dragan Maksimović (M’89) received the B.S. and
M.S. degrees in electrical engineering from the University of Belgrade, Belgrade, Yugoslavia, in 1984
and 1986, respectively, and the Ph.D. degree from
the California Institute of Technology, Pasadena, in
1989.
From 1989 to 1992, he was with the University
of Belgrade. Since 1992, he has been with the Department of Electrical and Computer Engineering,
University of Colorado, where he is currently an
Associate Professor and Co-Director of the Colorado
Power Electronics Center (CoPEC). His current research interests include
power electronics for low-power, portable systems, digital control techniques,
and mixed-signal integrated circuit design for power management applications.
Dr. Maksimović received the NSF CAREER Award and a Power Electronics
Society TRANSACTIONS Prize Paper Award in 1997.
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Robert W. Erickson (F’00) received the B.S., M.S.,
and Ph.D. degrees in electrical engineering from
the California Institute of Technology, Pasadena, in
1978, 1980, and 1982, respectively.
Since 1982, he has been a member of the Faculty
of Electrical and Computer Engineering, University
of Colorado, Boulder, where he is currently Professor and Chairman. He established and co-directs
the Colorado Power Electronics Center, which is
helping the industry in low-harmonic rectifiers,
dc–dc converters for battery-powered systems, and
magnetics modeling for multiple-output converters. He is the author of the
textbook Fundamentals of Power Electronics, now in its second edition. He is
the author of approximately 60 journal and conference papers in the area of
power electronics.
Dr. Erickson received the IEEE Power Electronics Society Transactions Prize
Paper Award, for the paper “Nonlinear Carrier Control for High-Power-Factor
Boost Rectifier” in 1996.