Research Report
2005-27
Double Side Control of Wound Rotor Induction Machine for
Wind Energy Application Employing Half Controlled
Converters
D. Panda, T. A. Lipo*
Rockwell Automation Advanced Technology Lab,
Richmond Heights, Ohio 44143, USA, dpanda@ieee.
org
*Department of Electrical and Computer Engineering
University of Wisconsin, Madison, WI 53706, USA,
lipo@engr.wisc.edu
isconsin
lectric
achines &
ower
lectronics
onsortium
University of Wisconsin-Madison
College of Engineering
Wisconsin Power Electronics Research
Center
2559D Engineering Hall
1415 Engineering Drive
Madison WI 53706-1691
© 2005 Confidential
Double side control of wound rotor induction machine for wind energy application
employing half controlled converters
Debiprasad Panda 1, Thomas .A.Lipo 2
Rockwell Automation Advanced Technology Lab,
Richmond Heights, Ohio 44143, USA, dpanda@ieee.org
2
Department of Electrical and Computer Engineering
University of Wisconsin, Madison, WI 53706, USA, lipo@engr.wisc.edu
1
Abstract – A double side converter fed wound rotor induction
machine control for a wind energy application is proposed in
this paper. In order to reduce the cost of a wind generator
system, a new configuration using half controlled converters
for both the stator and rotor circuit as well as for the line side is
proposed. The proposed controller reduces the required KVA
rating of both machine side and line side converters, improves
the efficiency of the wind generator, helps operating over a
wide speed range and supports near unity power factor
interface with the grid. It is shown that the combined KVA
rating of both the machine side converters are even less than
that of the machine side converter for a conventional rotor side
control configuration. Also, both stator and rotor side being
connected to the grid through two power stages, the system will
be least affected by the fault or disturbances in the gird. The
proposed configuration is simulated for a 30kW wound rotor
machine in SABER and the simulation results are presented.
Keywords: wound rotor induction machine, wind generator,
Super-synchronous generating mode, half controlled converter,
reduced switch configuration.
List of symbols:
V
: stator terminal voltage
s
: rotor terminal voltage
Vr
i s = i sd + ji sq
i
i
r
=i
ms
ψs
ψr
rd
+ ji
rq
: stator current
: rotor current
: stator flux magnetizing current
: stator flux
: rotor flux
ψ m : air gap flux
Tq
: electromagnetic torque
L0
: magnetizing inductance
σs
: stator leakage factor
σr
: rotor leakage factor
es
: stator induced voltage
er
: rotor induced voltage
ωs
: stator angular frequency
i sr
:stator current in rotor reference frame
s
r
:rotor current in stator reference frame
i
µ
ε
:angular position of stator magnetzing flux
:angular position of the shaft
I. INTRODUCTION
The state-of-the-art technology for wind power employs
a doubly-fed machine with only rotor side control where the
stator is directly connected to the grid [1-4]; which leads to
significant reduction in cost. However, such systems operate
only over a limited speed range; typically field weakening is
not possible and circulating power flows between the stator
and rotor circuit in sub-synchronous generating mode. Also,
the stator of these systems being directly connected to the
grid, suffers from the disturbances and faults occurring in
the grid. By employing power converters both in the stator
and the rotor side, some of these problems can be
eliminated. In addition, the excitation current can be shared
amicably between the stator and the rotor of the machine.
By doing so, the winding design (copper volume) and losses
in the machine as well as in the power converter can be
optimized and the machine size may be reduced. A control
algorithm employing expensive full bridge converters for
both the stator and the rotor was reported in [5, 6] for high
power motoring application. In a recent publication [7] by
the present authors, less expensive half controlled converters
are proposed for machine side control; but, in line side still
an expensive full bridge converter was used. In this paper,
less expensive half controlled converters are proposed for
both the machine and line side control without any major
sacrifice in the performance of a wind generator.
In the organization of this paper, first the operating
principle of a doubly fed induction motor is explained, the
different operating modes for this machine are given and the
best operating regime for wind generator is chosen. Then a
cost effective power converter is proposed for this operating
regime; and the advantages and disadvantages of the
proposed converter are discussed. After that a control
algorithm for the proposed wind generation system is
formulated. The algorithm is validated through SABER
simulation results and finally conclusions are made.
II. OPERATING PRINCIPLE
The operating principle of a doubly-fed wound rotor
induction machine is well documented in the literature [1-6].
In a doubly-fed wound rotor machine, the control can be
realized either from the stator side or the rotor side or from
both the sides. The machine can be controlled as a generator
or a motor in both the sub- and super-synchronous operating
mode. Depending on the operating modes, the power flow
varies in the stator and the rotor winding of the machine and
the detail power flow diagram for different operating modes
are given in Fig 1.
In conventional rotor side control [1-4], with two backto-back IGBT based power converters, all four operating
modes can be realized. However, during such operation, it
may be seen that in sub-synchronous generating (i) and subsynchronous motoring (ii) modes the power flow direction
through the stator and the rotor side are of opposite
direction. Hence, there essentially occurs a condition where
circulating power flows around the stator and the rotor and
which reduces the efficiency at speeds below synchronous
speed. In order to avoid such possibilities, the machine
should be operated only in the super-synchronous mode.
However, in conventional rotor side control scheme, the
super-synchronous speed mode is achieved only above the
rated synchronous speed and thus the operating speed range
becomes limited.
The proposed method in this paper explores a new
control scheme where the machine can be theoretically
operated at any speed (speed is limited only due to
mechanical restrictions) and at the same time it will operate
always in super-synchronous mode. Hence, all the analysis
and control of the machine for this paper will be restricted to
super-synchronous generating mode (mode (iii) in Fig. 1).
latter, the stator circuit is represented as a controllable
current source. The steady-state phasor diagram for the
super-synchronous generating
q-axis
isσsL0ωs
Vs
es = er
irq
ir
ψm
isd ird
isq
ims
ψs
is
Fig. 3 Steady-state phasor diagram (stator reference frame) of a doublyfed wound rotor induction machine for the super-synchronous generating
mode
mode in stator reference frame is given in Fig. 3. In the
diagram, counterclockwise direction of rotation is assumed
as the positive direction. Neglecting the stator resistance, it
may be assumed that the stator flux ψ s has two
(i)
components; the stator leakage component and the
magnetizing component (see Fig. 2a). The former is due to
the stator current alone, while the latter is due to both the
stator and rotor currents. An equivalent current ims can be
(ii)
(iii)
defined in the stator reference frame, which is responsible
for the stator flux. This is termed as the stator flux
magnetizing current. The direction of ψ s (which is in phase
(iv)
Fig. 1 Power flow diagram of a doubly-fed wound rotor induction
machine for: (i) sub-synchronous generating mode, (ii) sub-synchronous
motoring mode, (iii) super-synchronous generating mode and (iv) supersynchronous motoring mode
with
ims ) is defined as the d-axis and, the direction of the
stator voltage, which is in quadrature with ψ s , is termed as
the q-axis. It is possible to resolve
i s and i rs along and
perpendicular to ims . The components of the currents along
the d-axis are represented with subscript ‘d’, and those along
the q-axis with subscript ‘q’. The mathematical relations
between the currents in this stator flux reference and the
expression for the torque are given below.
(a)
(b)
Fig. 2 Equivalent circuit diagrams of a doubly-fed wound rotor induction
machine: (a) for rotor side control (LHS) and (b) for stator side control
(RHS)
i sq = −(n r / n s ) × i rq
(1)
ims = (n r / n s )ird + i sd
(2)
A. SUPER-SYNCHRONOUS GENERATING MODE
Simplified equivalent circuits of a doubly-fed wound
rotor induction machine controlled from the rotor side and
the stator side are given in Figs 2a and 2b respectively. In
Fig. 2a, it is assumed that the rotor currents can be injected
at any desired phase, frequency and magnitude. Therefore,
the rotor circuit can be represented by a controllable current
source. The equivalent circuit is drawn in the stator
reference frame; hence the rotor current is represented as
irs . Similarly, when the control is exerted from the stator
side the equivalent circuit can be drawn as in Fig. 2b. In the
Tq =
3 P
2 2
i ms i rq
L0
1+σ s
(3)
From the above equations, it may be seen that the
magnetizing current of the machine ims is a summation of
the d-axis components of the stator and the rotor currents.
Also, the q-axis components of both sides are simply related
by their turns ratio and are of opposite direction. The same
relationship can be explained with the phasor diagram in
Fig. 3. The torque of the machine is a product of
magnetizing component and q-component of the rotor
current. Hence, torque can be controlled either by
controlling flux component, the torque component or by
controlling both simultaneously.
induced voltage
es and the lagging angle is dependent on
the leakage inductance and the stator current. If load current
is varied then stator current angle with respect to the stator
induced voltage will also vary proportionately. Similarly,
for any positive value of i sd , the phase angle between
es and is will be reduced and the power factor will be
approaching unity power factor. Thus beyond a certain
value of i sd , the current will start leading the induced
voltage.
Fig. 4 Proposed wind generator system
B. PROPOSED POWER CONVERTER CONFIGURATION AND
CONTROL STRATEGY
The proposed wind generator system is shown in Fig. 4.
It may be seen that both the stator and the rotor side are
connected to the grid through two half controlled power
stages in each side. In this operating mode as mentioned
earlier, the power is extracted from both the stator and the
rotor of the machine and thus the converters connected to
the machine windings will always function as rectifiers and
the total power of the machine can be shared arbitrarily
between the stator and the rotor of the machine.
Similarly, two new half controlled converters are
employed for the line side. Each half controlled converter
employs only three active switches and each switch is rated
only for half the rated power of the wind generator. Thus,
combining both the line side half controlled converters, total
six active switches are used and all are rated for only half
the rated power. In addition, twelve diodes and six thyristors
are also used in the line side and all the thyristors and diodes
are rated for only half the rated power. All six thyristors, and
six out of twelve diodes (the diodes antiparallel to the active
switches) are line commutated type, which are much less
expensive. Thus, the line side converter employing two half
controlled circuits will be much more cost effective
compared to a full bridge converter (in full bridge all six
switches and diodes are of inverter grade and rated for full
power).
In an earlier paper [8], it was shown that the distortion in
the current and voltage waveforms of a half controlled
converter is somewhat less for a lagging power factor
(typically 15 to 30 degree lagging current) type current
commands compared to that with unity power factor and
leading power factor type current commands. From Fig. 3, it
may be seen the stator side terminal voltage, Vs is lagging
the stator induced voltage
operating as a generator. If
es so that the machine is
i sd is assumed to be zero, then
o
the stator current will be exactly 180 out of phase with the
stator terminal voltage. Hence, it may be inferred that the
stator current will be lagging in nature with respect to stator
Similar operation may be explained for the rotor side,
with the help of Fig. 2b and a similar phasor diagram in
rotor reference frame. In order to operate both the stator and
the rotor side in the lagging power factor operating mode or
reduced leading power factor condition, the excitation
current through the individual stator and rotor side should be
much less than the torque component of the current. The
best means to achieve this result is to distribute the flux
equally between the stator and the rotor windings. In such
case, each individual half-controlled converter will handle
only 50% of the excitation current. On the other hand, both
the converters will carry the rated active component of the
current. Thus, each converter will carry 1 pu active
component of current and 0.5 pu reactive component
current. With such an arrangement power factor in each
converter can be controlled to a certain extent. For smaller
load or zero load, the flux may be reduced proportional to
the torque demand. Thus, by controlling the field current as
a function of load, the phase currents through both the stator
and the rotor windings can be achieved with less distortion.
C. COMPARISON OF KVA RATING OF THE MACHINE SIDE
CONVERTER BETWEEN THE PROPOSED AND CONVENTIONAL ROTOR
SIDE CONFIGURATION
In this proposed doubly-fed wound rotor induction
generator system, the air gap flux producing component of
the currents will be distributed equally between the stator
and the rotor windings. In such case, each individual halfcontrolled converter will handle only 50% of the excitation
current. On the other hand, both the converters will carry the
rated active component of the current. Thus, each converter
will carry 1 pu active component of current and 0.5 pu
reactive component current.
In the conventional rotor side control scheme, the
converter has to supply the full excitation current as well as
the active torque component of the current through the rotor
winding. If one assumes both the rated magnetizing current
and the torque component of currents are equal (i.e. 1 pu
each) and orthogonal to each other, the total current rating of
the rotor side converter as well as the current rating of the
2 2
rotor winding goes up to 1.414 pu ( (1 + 1 ) = 1.414 ).
Hence, both the rotor side converter and the rotor windings
need to be over designed by 41.4% to accommodate the
magnetizing current. However, by splitting up the excitation
current equally (0.5 pu) between the stator and the rotor
windings, the KVA rating of each side power converter will
2
2
be reduced to only 1.118 pu ( (1 + 0.5 ) = 1.118) of the
rated power. Since each half controlled circuit is having only
3 switches instead of 6 switches for its fully controlled
counter part, the KVA rating of the active switches
combining both the stator and the rotor side machine end
power converter reduces to only 1.118 pu of the rated power
of the machine compared to 1.414 pu for its fully controlled
counterpart. Thus, with the proposed power converter
configuration the KVA rating of the combined machine side
power converter is reduced by 30%. Using similar
arguments it may be shown that with the proposed
configuration, the efficiency of the combined machine side
power converter will be higher than the conventional rotor
side control scheme.
Following a similar analysis as above, it may be shown
that by reducing the burden on rotor winding the total
copper loss of the machine is expected to improve. With
proposed control scheme the total copper loss of the
machine will improve by 16.67% [7]. Again, in
conventional scheme the losses in the rotor windings being
larger the cooling of the machine will become difficult
compared to the proposed method. Most important point to
note here is that by running the system always in super
synchronous generating mode and by employing two
separate sets of power converter in both the stator and the
rotor side the flow of circulating currents between the stator
and the rotor side converters is avoided.
D. STARTING METHOD
It can be seen from Fig. 4 that an extra IGBT (Sw_st)
switch in series with a diode are added between the DC bus
and phase A of the stator winding for providing starting
excitation to the machine. With this extra switch DC
excitation can be provided to the stator though phase A and
either one of phase B or phase C. In this paper, phase B is
chosen for that purpose. Thus, controlling Sw_st and Sw16
simultaneously, a DC current can be passed through phase A
and phase B, while phase C remains open circuited. This DC
excitation will generate a DC flux in the stator magnetic
circuit. Since, the rotor is assumed to be rotating freely by
wind thrust, the rotating phases in the rotor will pass through
this DC flux and voltages will be induced in the rotor
windings. At this time, power can be extracted from the
rotor winding to the DC bus by actively controlling the rotor
side half controlled converter. After few cycles of such
operation, when sufficient current builds up through the
rotor winding, Sw-st will be switched off and a regular halfcontrolled converter for the stator side will be activated and
the control will be shifted from starting mode to normal
running mode. It should be mentioned here that the extra
starting switch needed for this circuit is rated for only 1520% of the rated current. Thus, this starting switch does not
add significant extra cost to the system. Another alternative
starting method is proposed in [7], where the special starting
switch (Sw_st) is not required; instead a three phase
contactor is employed between the stator winding and the
grid.
E. LINE SIDE CONVERTER WITH TWO HALF CONTROLLED
CIRCUITS
In the present application, the line side converter needs
to convert the power from DC to AC for which the inverter
or regenerative operation of the converter is essential. None
of the previously investigated half controlled converters [8,
9] have the above feature. In this paper, by having a hybrid
configuration employing thyristors and IGBTs, a 3-phase
half controlled power converter is realized which is capable
of acting as a rectifier as well as an inverter in half
controlled manner. Then by combining the two
complementary half controlled configuration a similar
operation of a fully controlled 4-quadrant converter is
achieved (see Fig. 4).
F. COMMUTATION OF THE THYRISTORS FOR LINE SIDE
CONVERTER
With the proposed line side converter each leg of
individual half controlled converter consists of one IGBT,
one line commutated thyristor and two anti-parallel diodes.
Thyristors are enabled only during inverter (regeneration)
operation. During inversion, the upper half thyristors (T31,
T33, T35 in Fig. 4) conduct when the respective phase
voltages are positive and currents are negative (i.e. flows
from the converter to the line). Similarly, the lower half
thryistors (T44, T46, T42) conduct when the respective
phase voltages are negative and currents are positive.
The commutation of the lower half thyristors (T44, T46,
T42) can be explained with the help of Fig. 5. The symbols
t d , t µ and t q in Fig. 5 represents the commutation delay
time, commutation time and minimum
turn-off time respectively. Normally, the lower half
thyristors (T44, T46, T42) which are connected to the
negative DC bus voltage, can be made conducting one at a
time even when their respective phase voltages are negative
given the fact that the DC bus voltage is always higher than
peak line voltage. For example in Fig. 5, when
Fig. 5 Commutation methodology for the line commutated thyristors
phase A voltage is negative and phase B and Phase C
voltages are positive; and also if phase A current is positive,
then thyristor T44 will be conducting till either T46 or T42
are enabled. If either of them is enabled, then T44 will start
commutating. In this example T44 must be commutated off
between O and P. After point P, the phase A voltage
becomes positive with respect to phase B and if T44 still
conducts at this point, then it can not be commutated off for
another cycle and similarly T46 can not be turned on.
Hence, the commutation between phase A and phase B
should be completed between O and P. Again, a thyristor
needs to be kept off for a minimum period of t q before the
voltage across it again becomes positive. Thus, the
commutation process has to be completed at least
t q (70 µ s for this example) time prior to point P.
Thus, the maximum conduction period of each lower
half thyristors will be from 1800 to 3300 while the
commutation period spans from 3000 to 3300. The thyristors
do not get turned off instantaneously. The commutation
overlap time t µ is decided by the line inductance and the
current magnitude at the commence-ment of commutation.
As mentioned earlier, if the commutation is delayed beyond
3300 then commutation failure may be observed. Similarly if
commutation is advanced much before 3300, then a line side
short circuit situation will arise with the incumbent phase
thyristor T46 and the outgoing phase diode D44 conducting
until it reaches 3300. However, the voltage difference
between the incumbent phase and outgoing phase close to
3300 being less and at the same time two phase inductance
being in series with the line voltage, the current flowing
between the phases will not be significant for a small
duration (about 100 micro sec). However, it will cause
somewhat distortion in the current waveform since, negative
current (flowing into the line) will flow through the
outgoing phase when positive current is desired. Hence, it is
desirable for the commutation to be completed at the close
vicinity of 3300 (not later than 3300 and at the same time not
much earlier). For a lesser current it will be wise to delay the
commutation process in order to have less current distortion.
Thus, a commutation delay t d , inversely proportional to the
phase current, is introduced here. Similarly, the operation of
upper half thyristors can be explained for the positive half
cycles.
III. CONTROL SCHEME
The proposed control scheme can be divided into
machine side control and line side control.
A. MACHINE SIDE CONTROL SCHEME
A detailed machine side control block diagram is given
in Fig. 6. The proposed control algorithm employs a
conventional PI controller for its outer speed loop. The
speed loop controller generates torque reference as its
output. In order to facilitate implementation of the
controller, the relation between the stator and rotor q-axis
component of current, the expression of torque and
expression of magnetizing currents given in Eq. (1) to Eq.
(3) can be used. From the given equations, it is clear that the
quadrature axis rotor current irq generates the machine
torque. The stator q-axis current i sq is automatically
developed as the reflection of irq (see Fig.6). Thus in the
control block diagram it is shown that the torque reference
*
( Tq ) is directly proportional to
irq* . The stator component
q-axis current reference is generated in the block diagram
following Eq. (1). The magnetizing current ims can be
supplied from both the half controlled converters by
arbitrary current sharing as given by Eq. (3). However, in
the proposed method, both the converter always shares the
magnetizing current equally.
In the block diagram it is shown that the total
magnetizing current is multiplied with a gain of 0.5 to
generate the individual magnetizing reference currents
*
*
( i sd , ird )
for
the
two
half
controlled
converters.
Interestingly, the total magnetizing flux current reference
*
ims
is not constant for the proposed method. Thus, the iron
losses using this type of controller will be reduced by the
square of the magnetizing current in addition to the reduced
copper loss mentioned previously. In Fig. 6 it is shown that
the same as a function of the required torque
Tq* of the
machine. From Eq. (3) it may be seen that the torque is a
function of ims and irq so that the magnetizing current is
varied as a function of load. The same is exercised in order
to reduce the THD in the stator and the rotor winding
currents for all load conditions.
Since, half controlled circuits drive the machine and
since the excitation current capability of the same is a
function of load, current is varied with load in this proposed
method of excitation. Once the d-axis and q-axis current
references for both the stator and the rotor windings are
found, a dq to abc transformation is applied so that the three
phase reference currents for stator and rotor currents are
generated. A hysteresis controller is employed in each side
for controlling the stator and rotor currents. The speed of the
machine is sensed through a shaft sensor and by integrating
this signal the position of the rotor ε is found. In the
proposed control the stator and rotor windings are intended
to share the power equally. In order to achieve this result,
the stator and rotor frequencies are always maintained at
half of the shaft frequency and their phase sequence is
opposite to each other. The stator frequency sequence is
maintained in the same direction as the shaft and the rotor
frequency is of opposite sign. By passing the speed signal
through a gain block of 0.5 and then integrating, the position
of the stator flux µ is obtained. These ε and µ values are
required for dq to abc transformation as shown in Fig. 6.
B. LINE SIDE CONTROL SCHEME
The proposed control block diagram for the line side
converter is given in Fig. 7. In the proposed control
Fig. 6 Block diagram of the machine side control algorithm
Fig. 7 Block diagram of the proposed line side control algorithm
scheme, the outer loop employs a PI controller for
regulating the DC bus voltage. The output of a PI controller
also determines the amplitude of current through the phases.
This amplitude, when multiplied with sinusoidal unit vectors
derived in phase with the phase voltage, gives the combined
current references for each phase. However, the combined
reference currents multiplied by 0.5 gives the reference
currents for the phases of individual half control rectifiers.
In the half controlled circuits, current of each phase can
follow the references only through one-half cycle when the
IGBTs are in operation; whereas in the other half cycle
when the thyristors are conducting the current remains
unregulated and therefore can not always follow the
reference. In such a case, the complementary side converter
must compensate for the resulting error due to its
counterpart. Hence, an active filter type control
configuration has been added with the current references
generated from the voltage controller. It can be seen from
Fig. 7 that the individual rectifier reference currents are
added to the error of the complimentary half to generate the
final current references of each half controlled rectifiers.
However, the current references through each phase of both
the rectifiers are limited to half of the peak rated current.
The current controller chosen was a conventional hysteresis
current controller.
IV. SIMULATION RESULTS AND DISCUSSION
The proposed control scheme with the half controlled
power converters was investigated on a 30kW wound rotor
induction machine using SABER. The simulation results
with full excitation current and 25% rated load is shown in
Fig. 8(i). The results at 25% load with field excitation as the
function of load are shown in Fig. 8(ii). It may be seen that
the current waveforms are markedly improved in case of
variable field excitation. In contrast the waveforms with
rated excitation and at lower load are highly distorted and
the effect of the distortion is visible in the torque waveform.
Hence, the proposal to vary the field excitation as function
of load is validated. The steady-state simulation results at
two typical operating speed (30Hz and 90Hz of shaft speed)
and the FFT of the current waveforms are given in Figs 9(i)
and 9(ii) respectively. It may be seen that that both the stator
and rotor current waveforms are of same frequency and their
magnitude differs slightly due to non-unity turns ratio
between them. It is also apparent that the current waveforms
are quite smooth and the corresponding FFT shows that the
currents contain typical even order harmonics but their
magnitude is not appreciably high.
The simulation results of starting performance with both
contactor-start method [7] and the switch-start method are
given in Figs 10(i) and 10(ii) respectively. In the contactorstart method (Fig. 10(i)) it may be seen that initially the
stator draws the excitation current from the grid and the
rotor voltages are induced due to the stator excitation. When
the rotor currents build up sufficiently, the contactor is
switched off and the stator side half controlled takes over the
control. Similarly for switch-start method (Fig. 10(ii)),
initially the DC excitation is provided through phase A and
phase B and due to this excitation rotor voltage builds up
and when rotor starts generating, the DC excitation is
removed and the half controlled converter is switched ON
with its regular control scheme. Thus, it is demonstrated that
the machine can start successfully without incurring much
extra cost to the system.
Similarly, the proposed line side half controlled
converters were simulated with 480V line voltage and 3mH
line inductances and the simulated waveforms are given in
Fig. 11(i) to (iii). In Fig. 11(i), the current waveforms of
individual half controlled converters (ia1, ib1, ic1 and ia2,
ib2, ic2) as well as their combined waveforms (ia, ib, ic) are
shown. It may be seen that individual waveforms are
sinusoidal and regulated in one half cycle and unregulated
nature in the other half cycle. These results are taken when
the active filter algorithm was not enabled. With active filter
enabled, the similar waveforms are plotted in Figs 8 (ii) and
(iii) for full load and 1/3 rd load respectively. It may be seen
the peak currents through individual half controlled
converters (ia1 and ia2) are one-half (15A) of their
combined (30A) current waveform (ia). Hence, it can be
inferred that the diodes, thyristors and IGBTS of each half
controlled converter is only 50% of the total power. The
FFTs of the combined current waveforms are also given in
the upper traces of each figure. The calculated THD at full
load is found as 6.63%. At 1/3rd load, the THD is less than
3%. Also, the combined current waveform is maintained at
unity displacement power factor.
V. CONCLUSION
A new double side control algorithm for a doubly fed
wound rotor induction machine employing half controlled 3phase converters is explored in this paper. Using half
controlled converters the system cost is reduced and at the
same time wide operating speed range and unity power
factor interface with the grid is achieved. By splitting the
excitation current equally between the stator and rotor
windings the efficiency of the machine as well as the power
converter are improved; and also it further reduces the KVA
rating of the machine side converters. Even though the
motoring mode of operation is sacrificed in the proposed
method, for wind energy application that is not a serious
problem since wind itself can be used for accelerating the
machine. The machine side power converters are shoot
through safe, do not require dead time delay or isolated
power supply for their gate drives. Since both the stator and
the rotor side frequency can be controlled at will, the zero
frequency condition can be avoided so that the proposed
algorithm is suitable for sensorless operation. Since, both
side of the machine is connected to the grid through power
converters; the system can be isolated from the grid in a
controlled fashion during fault condition.
Turns ratio (ns / nr)
1.21
VI. REFERENCES
[1] R. Pena, J.C.Clare, G.M.Asher, “Doubly fed induction generator using
back-to-back PWM converters and its application to variable-speed
wind-energy generation”, IEE Proc., Vol. 143, Pt.B, No.3, pp. 231-241,
May 1996.
[2] Y. Tang and L. Xu, “A flexible active and reactive power control
strategy for a variable speed constant frequency generating system”,
Proceedings of the IEEE IAS Annual Meeting, 1993, pp 568-573.
(i)
[3] D. Panda, E. Benedict, G. Venkataramanan and Thomas A. Lipo,”A
novel control strategy for the rotor side control of a doubly-fed
induction machine”, Proceedings of IEEE IAS Annual Meeting,
Volume:3,2001 pp 1695-1702.
[4] R. Datta, V. T. Ranganathan, “Direct power control of grid-connected
wound rotor induction machine without rotor position sensors”, IEEE
Transtactions on Power Electronics, Vol. 16, Issue 3, May 2001, pp
390-399.
[5] Y. Kawabata, E. C. Ejiogu, T. Kawabata, “Vector-controlled doubleinverter-fed wound-rotor induction motor suitable for high-power
drives", IEEE Transactions on Industry Applications, Vol. 35, No. 5,
Sept/Oct 1999. pp 1058-1066.
[6] Poddar G., Ranganathan V. T., “Direct torque and frequency control of
double-inverter-fed slip-ring induction motor drive”, IEEE Transaction
on Industrial Electronics, Vol. 51, Issue 6, Dec. 2004 pp 1329-1337.
[7] D. Panda and T. A Lipo, "Reduced switch count double converter fed
wound rotor induction machine drive for wind energy application",
IEEE International Electrical Machines and Drives Conference,
IEMDC 2003, Vol. 3, 1-4 June, 2003, pp. 1924-1931.
[8] Jun Kikuchi, Madhav D. Manjrekar, Thomas A Lipo, "Performance
improvement of half controlled three phase pwm boost rectifier",
Proceedings of IEEE Power Electronic Specialists Conference, 1999,
Vol. 1, January 1999, pp. 319-324.
[9] D. Panda and T. A Lipo, "A half-power rating improved 3-phase sixswitch boost rectifier using two half controlled configurations with a
common DC bus", Proceedings of IEEE Power Electronic Specialists
Conference, 2003, Vol. 3, 15-19 June, 2003, pp. 1069-1074.
Appendix 1: Machine parameters
Power
30kW
Voltage (Vs)
480V
Frequency (fs)
60Hz
Poles (P)
6
Inertia (J)
5 kgm^2
Frictional coefficient (B)
0.0519247
Magnetizing inductance (Lm)
24.1mH
1.326mH
Stator leakage inductance ( σ L )
s
0
Rotor leakage inductance ( σ r L0 )
1.2467mH
Stator resistance (Rs)
Rotor resistance (Rr)
0.107 ohm
0.062 ohm
(ii)
Fig. 8 Simulation results at (i) 25% load with full field excitation and (ii)
at 25% load with filed excitation as a function of load
Fig. 9 Typical simulation results of a 30kW doubly-fed wound rotor induction machine: (i) with rotor shaft frequency at 30Hz (LHS) and (ii) with rotor
shaft frequency at 90Hz (RHS)
Fig. 10 Simulation results during starting of a 30kW doubly-fed wound rotor induction machine: (i) with contactor start method (LHS) and (ii) with
extra switching start method (RHS)
Fig. 11 Typical simulation results of a 30kW line side converter : (i) with full load and without active filter algorithm, (LEFT ), (ii) with full load
and with active filter algorithm (CENTER) and (ii) with 30% load and with active filter algorithm (RIGHT)