A Current-Source Active Power Filter with a New DC Filter Structure
Mika Salo
Department of Electrical Engineering, Institute of Power Electronics
Tampere University of Technology
P.O.Box 692, FIN-33101 Tampere, Finland
Abstract- The main drawback of the current-source active
power filter is the heavy and bulky dc side filter. The large dc filter is needed to store the energy of the compensated harmonic
components. In this paper a new smaller dc filter structure is
proposed for the current-source active power filter. In the presented dc filter structure the energy of the most important harmonics are stored in resonant circuit which makes it possible to
decrease the overall size of the filter. The function of the proposed dc filter structure is examined with both simulations and
experimental tests.
Load
Rectifier
bridge
Lf
Power i ldA
supply i
tA
isA
us
Ls
T2
T4
T6
Cs
(a)
Load
Rectifier
bridge
Lf
Power i ldA
supply i
tA
T1
Supply
filter i
rA
isA
T3
T5
dc filter
idc
Ldc
icr
us
Lr
Ls
T2
T4
Cr
T6
Cs
(b)
Fig. 1. The current source active power filter (a) with conventional dc filter
structure and (b) with modified dc filter structure.
13th load current harmonics. Other components in the dc circuit
have order of 18, 24, 30 etc. However, the 6th harmonic components makes the biggest ripple in the dc current because of it’s
large magnitude and low frequency.
The proposed dc filter structure shown in Fig. 1(b) is designed
to damp the 6th harmonic component effectively. The proposed
dc filter structure uses a parallel resonant circuit which is tuned
for 6th harmonic component. Other harmonics of the dc circuit
are filtered with the inductor L dc connected in series with the resonant circuit. With the proposed modified dc filter structure the
amount of the total filter inductance can be significantly reduced.
Impedance of the conventional dc filter structure is
II. PROPOSED DC FILTER STRUCTURE
The most important harmonics in typical nonlinear loads are
5th and 7th harmonic components which produce 6th harmonic
component in the dc circuit of the active power filter. Next important harmonic component in the dc circuit of the CSAF is the
12th harmonic which is caused by the compensation of 11th and
idc
T5
Ldc
I. INTRODUCTION
In recent years, active power filters have been widely investigated for the compensation of harmonic currents in electrical
power systems. These active power filters are divided into two
types: voltage-source active filter (VSAF) and current-source
active filter (CSAF). CSAF has advantages of excellent current
control capability, easy protection and high reliability over
VSAF [1]. The main drawbacks of the CSAF has been so far the
lag of proper switching devices and large dc side filter. The new
IGBTs with reverse blocking capability are being launched on
the markets which are suitable for CSAF [2]. However, the
bulky and heavy dc side filter is still a problem.
Fig. 1(a) shows the most common main circuit structure of
the current-source active power filter (e.g. [3]-[5]). The line
current characteristics are improved by injecting the current
components opposite to the harmonics of the load current.
The energy of the injected harmonic components is stored in
and restored from the dc circuit which makes ripple in the dc
current. In order to keep this ripple in an acceptable level relatively large dc filter inductor is needed.
In this paper a new smaller dc filter structure is presented
for the current-source active power filter. The proposed dc filter structure is shown in Fig. 1(b). In the presented dc filter the
energy of the most important harmonics are stored in resonant circuit which makes it possible to decrease the overall
size of the filter.
T3
T1
Supply
filter i
rA
dc-filter
Z(s)
= R dc + sL dc
(1)
where R dc is the resistance of the dc filter. For modified dc filter
structure can be written as
Z(s)
Lr s + Rr
= R dc + sL dc + --------------------------------------2
Lr Cr s + Rr Cr s + 1
(2)
III. CONTROL OF CSAF
The proposed dc filter topology can be used with any control system of CSAF. Fig. 3 shows a control system [6]
which is practical for current-source topology. It is realized
in the synchronously rotating reference frame where the active power of the active filter can be simply controlled with
real axis component isx and the reactive power with imaginary axis component isy of the filter current. The superscript
s in space vector variables and x/y in space vector components
refers to a synchronously rotating coordinate system. The harmonic compensation is based on the feedforward control of
the load currents. The active filter currents isxy are controlled
in an open-loop manner.
The reference values for active filter current vector are calculated as follows:
i *sx
= i *ffx + i *dcx
(3)
80
Impedance[dB]
where L r , C r and R r are the inductance, capacitance and resistance of the parallel resonant circuit respectively.
Impedances of the conventional (--) and modified (-) dc filter
structures are plotted in Fig. 2 as a function of the angular frequency. With conventional dc filter L dc =170 mH and R dc =8Ω.
The modified dc filter have parameter values: L dc =30 mH,
R dc =3Ω, L r =15 mH, Rr = 2Ω and Cr =18.7 µF. The resistances are approximate values at 300 Hz. Fig. 2 shows also the impedance curve with 45 mH dc filter inductor (.-) which is the
total inductance of the modified dc filter structure. These parameter values are used with the active power filter of which nominal
power is 5 kW.
Fig. 2 shows that impedance of the modified dc filter is increased rapidly at 1900 rad/s (300 Hz). This is caused by the parallel resonant circuit which is tuned for this frequency (6 th
harmonic component). At this frequency the impedance of the
modified dc filter is higher than the impedance of the conventional filter which indicates that the proposed filter structure can
store effectively the energy of the filtered 5th and 7th load current harmonics without large ripple in dc current. Fig. 2 shows
also that the impedance of the modified filter is very low at 2700
rad/s (420 Hz). This is caused by the series resonance of the proposed dc filter structure. The frequency of the series resonance is
determined by Cr and the parallel connection of L dc and L r .
When selecting the parameters for the modified dc filter structure it should be made sure that the frequency of the series resonance is not near to 12th harmonic component which is the next
important harmonic component in the dc circuit of the active filter. Otherwise, the series resonance should neither be near 6th
harmonic component. This can be avoided when the ratio of L r
and Ldc is between 1/2-2.
60
40
20
0 2
10
3
10
4
10
Angular frequency [rad/s]
Fig. 2. Impedances of the convential dc filter when Ldc =170 mH (--) and
L dc =45 mH (.-) and impedance of the modified dc filter (-) as function of the
angular frequency.
and
i *sy
= i *ffy + i *qy
(4)
* (both components combined in one expression),
where iffxy
*
i
and i * are outputs of the feedforward, dc current and
dcx
qy
the reactive power controls respectively. These two components form the rectifier current reference vector i rs * which is
transformed to the stationary reference frame and fed to the
modulator. Due to the open-loop control of the active filter
currents the currents references are not realized accurately
because the supply filter takes capacitive currents. Also, oscillations may occur in supply currents due to the LC-filter resonance if the active filter current references are rapidly
changed. Detailed describtion of the control methods to solve
both of these problems can be found in [6].
A. Dc current control
The task of the active filter is to compensate the harmonics of
the non-linear load. The magnitude of the dc current is changed
as the energy of the harmonic components is stored in and restored from the dc circuit. This ripple in the dc current is the basic feature in the active power filter and for that reason the dc
current control should not try to remove it. However, the dc current control should work effectively when the reference value of
the dc current is changed. For that reason, a non-linear PID controller, where the input of the controller is the square of the error
signal, is proposed for the control system shown in Fig. 3. With
small error values the controller acts slowly and when the error
value is increased faster control dynamics is achieved. Fig. 4
shows the block diagram of the non-linear PI controller where
k+1
the modified proportional gain P̃dc
depends on the error value. In practice, it is reasonable to limit P̃dck+1 between 0 and
max
P̃dc , which is done with Saturation block.
To understand how i dcx is constructed in the dc-current
control we can first consider that
3
--- u sx i dcx
2
= u dcbr i dc
(5)
Load
Lf
ildA
Power
supply
itA
CSAF
irA
isA
idc
Ldc
udcbr
3->2
Cs
ild
θs
e jθ s
s
ild
ildy
HPF
-1
CDC
Cr
idc
Modulator
us
*
idc
i*r
θs
e jθ s
+
FG
PID(e2)
Reactive
power control
ildx
^
ildx
Lr
Ls
HPF
^
ildy
s*
*
iqy
-1
CDC
*
isy
+
+
*
udcbr
s*
ir
=
+
*
isx
is
c1
+
*
*
iffy
*
iffx
+
*
idcx
+
Feedforward
control
Dc current
control
Fig. 3. Control system of CSAF with modified dc filter topology.
i.e. that the ac and dc active powers of the converter are
equal in steady state if the converter losses are ignored.
u dcbr is the dc-voltage of the rectifier bridge. By solving (5)
for i dcx and by using the reference values of i dcx and u dcbr
we have
*
i dcx
2
= ----------- u *dcbr i dc = cu *dcbr i dc
3u sx
(6)
which is used in Fig. 3 to transform the dc-voltage reference
of the rectifier bridge to vector variable.
Delay
, k+1 k+1
idc
*
idc
*
abs
k+1
Ts
Tdc
+
+
Saturation
Pdc
+
Pdc
Saturation
+
Delay
+
IV. SIMULATION RESULTS
The proposed control methods are tested with the simulation model. The simulation model is built in discrete form to
have close analogy with the microcontroller implementation.
Simulation is based on the control system shown in Fig. 3.
Sampling time of the feedforward and dc current controllers
is 50 µs and the modulation frequency 10 kHz. The supply filter is realized with parameters: L s =0.6 mH and C s =8 µF.
Figs. 5 and 6 show the simulation results of CSAF with modified and conventional dc filter structure respectively. As a nonlinear load a three-phase diode rectifier with RL-load is used.
The parameters of the modified dc filter are L dc =15 mH,
L r =10 mH and Cr =28.1 µF and conventional L dc =100 mH.
Ddc
Ts
+
, k+1
u*dcbr
Saturation
Fig. 4. Non-linear PID controller for dc current control.
In both simulations the the ripple of the dc current is about
1 A. The total harmonic distortion (THD) of the load current
in both simulations is 26.8%. THDs of the supply current with
modified dc filter is 4.2% and with conventional filter 4.0%.
Fig. 6(d) shows the current icr of the resonant circuit capacitor. It’s amplitude is about 3A and frequency 300 Hz. This ac
current is caused by the 300 Hz ac voltage component across
the resonant circuit. This 300 Hz ac voltage component is
caused by the compensation of the 5th and 7th load current
harmonics as was explained earlier.
Figs. 7 and 8 show the simulation results of CSAF with
10
10
5
5
it[A]
15
ild[A]
(a)
15
(b)
0
0
-5
-5
-10
-10
-15
-15
0
0.02
0.04
0.06
0
0.02
0.04
0.06
0.04
0.06
t[s]
t[s]
15
15
10
5
(d)
icr[A]
(c)
idc[A]
10
0
-5
5
-10
0
0
0.02
0.04
-15
0.06
0
0.02
t[s]
t[s]
15
15
10
10
5
5
0
(b)
it[A]
(a)
ild[A]
Fig. 5. Simulation results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with
RL-load. (a) Load current i ldA , (b) supply current itA , (c) dc current i dc and (d) current of the resonant circuit icr .
0
-5
-5
-10
-10
-15
0
0.02
0.04
-15
0.06
0
0.02
t[s]
0.04
0.06
t[s]
15
(c)
idc[A]
10
5
0
0
0.02
0.04
0.06
t[s]
Fig. 6. Simulation results of CSAF with conventional dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RL-load. (a) Load current i ldA , (b) supply current itA and (c) dc current i dc .
modified and conventional dc filter structure respectively
when as a non-linear load a three-phase diode rectifier with
RC-load is used. In this case the parameters of the modified
dc filter are L dc =30 mH, L r =15 mH and Cr =18.7 µF and conventional L dc =170 mH. The size of L dc is increased in both
modified and conventional dc filter solutions because RC-type
diode load contains more harmonics than RL-type load. Also,
the size of L r is increased in order to keep the series resonance
of the modified filter far enough from the parallel resonance.
Furthermore, smaller Cr decreases the current of the resonant
circuit capacitor. Anyway, Figs. 5(d) and 7(d) shows that icr
is much larger with RC-type load than RL-type load due to
the larger amount of harmonics included in RC-load which
increases also the harmonics of the dc circuit. THD of i ldA is
10
10
5
5
(b)
0
it[A]
15
ild[A]
(a)
15
0
-5
-5
-10
-10
-15
0
0.02
0.04
-15
0.06
0
0.02
0.04
0.06
0.04
0.06
t[s]
t[s]
15
15
10
10
(c)
(d)
icr[A]
idc[A]
5
0
-5
5
-10
0
0
0.02
0.04
-15
0.06
0
0.02
15
15
10
10
5
5
(b)
0
it[A]
(a)
ild[A]
t[s]
t[s]
Fig. 7. Simulation results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with
RC-load. (a) Load current i ldA , (b) supply current itA , (c) dc current i dc and (d) current of the resonant circuit icr .
0
-5
-5
-10
-10
-15
0
0.02
0.04
0.06
-15
0
0.02
t[s]
0.04
0.06
t[s]
15
idc[A]
10
(c)
5
0
0
0.02
0.04
0.06
t[s]
Fig. 8. Simulation results of CSAF with conventionald dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RC-load. (a) Load current i ldA , (b) supply current itA and (c) dc current i dc .
87.9% and THDs of itA with modified and conventional dc
filter 7.3% and 7.6% respectively.
V. EXPERIMENTAL INVESTIGATION
The prototype of CSAF is built using 1200 V, 50 A IGBTs.
The control system realization is based on the Motorola
MPC555 32-bit single-chip microcontroller. The supply filter
parameters and the sampling times of the control system are
the same as used in simulation model.
Figs. 9 and 10 show the experimental results of CSAF with
modified and conventional dc filter structure respectively. As
a non-linear load a three-phase diode rectifier with RL-load is
used. The dc filter parameters are same as used in simulations.
15
15
10
10
5
(b)
0
it[A]
ild[A]
5
(a)
0
-5
-5
-10
-10
-15
0
0.02
0.04
-15
0.06
0
0.02
t[s]
0.04
0.06
0.04
0.06
t[s]
15
15
10
5
(d)
icr[A]
(c)
idc[A]
10
0
-5
5
-10
0
0
0.02
0.04
-15
0.06
0
0.02
15
15
10
10
5
5
(b)
0
it[A]
(a)
ild[A]
t[s]
t[s]
Fig. 9. Experimental results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RL-load. (a) Load current i ldA , (b) supply current itA , (c) dc current i dc and (d) current of the resonant circuit icr .
0
-5
-5
-10
-10
-15
0
0.02
0.04
0.06
-15
t[s]
0
0.02
0.04
0.06
t[s]
15
idc[A]
10
(c)
5
0
0
0.02
0.04
0.06
t[s]
Fig. 10. Experimental results of CSAF with conventional dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RL-load. (a) Load current i ldA , (b) supply current itA and (c) dc current i dc .
By comparing Figs. 5, 6, 9 and 10 it can be seen that the
simulation and experimental results are in good agreement.
THD of i ldA in Figs. 9(a) and 10(a) is 27.1% and THDs of itA
with modified and conventional dc filter 3.3% and 3.2% respectively.
Figs. 11 and 12 show the experimental results of CSAF
with modified dc filter in two cases when RC-type diode rec-
tifier load is used. In the first case shown in Fig. 11 the load
current contains only 1.3% of 3rd harmonic current. In the
case of Fig. 12 the amount of 3rd harmonic is 7.1%. The 3rd
harmonic component in load currents causes 2nd harmonic
voltage component in the dc circuit. However, the impedance
of the modified dc filter for 2nd harmonic component is very
low as can be seen in Fig. 2. As a result, the ripple in dc cur-
10
10
5
5
(b)
0
it[A]
15
ild[A]
(a)
15
0
-5
-5
-10
-10
-15
-15
0
0.02
0.04
0.06
0
0.02
0.04
0.06
0.04
0.06
t[s]
t[s]
15
15
10
10
(d)
icr[A]
5
idc[A]
(c)
0
-5
5
-10
0
0
0.02
0.04
-15
0.06
0
0.02
t[s]
t[s]
Fig. 11. Experimental results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RC-load. Load current contains only small amount of 3rd harmonic component. (a) Load current i ldA , (b) supply current itA , (c) dc current i dc
and (d) current of the resonant circuit icr .
15
15
10
10
5
(b)
0
it[A]
(a)
ild[A]
5
0
-5
-5
-10
-10
-15
0
0.02
0.04
-15
0.06
0
0.02
t[s]
0.04
0.06
0.04
0.06
t[s]
15
15
10
10
5
(d)
icr[A]
idc[A]
(c)
5
0
-5
-10
0
0
0.02
0.04
0.06
-15
0
0.02
t[s]
t[s]
Fig. 12. Experimental results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RC-load. Load current contains large amount of 3rd harmonic component. (a) Load current i ldA , (b) supply current itA , (c) dc current i dc and
(d) current of the resonant circuit icr .
rent is much larger in Fig. 11(c) than in 12(c) due to the significant 100 Hz component (2nd harmonic).
In principle, symmetric three-phase load should not contain
3rd harmonic component which was also confirmed with simulations. The 3rd harmonic components seen in measured
load current is caused propably by the distorted supply voltages.
The experimental results of the conventional dc filter are
shown in Fig. 13. In Figs. 11-13 the THD of i ldA is around
82%. The THDs of itA shown in Figs. 11(b), 12(b) and 13(b) are
5.7%, 7.5% and 6.0%.
According to simulations and experimental investigation it
can be concluded that the proposed dc filter structure works
well if the load currents are symmetrical and do not contain
15
15
10
10
5
(b)
0
it[A]
ild[A]
5
(a)
0
-5
-5
-10
-10
-15
0
0.02
0.04
-15
0.06
0
0.02
t[s]
0.04
0.06
t[s]
15
(c)
idc[A]
10
5
0
0
0.02
0.04
0.06
t[s]
Fig. 13. Experimental results of CSAF with conventional dc filter structure when the current harmonics are produced using a three-phase diode rectifier
with RC-load. (a) Load current i ldA , (b) supply current itA and (c) dc current i dc .
3rd harmonic component. It seems that in practice threephase diode rectifier with RC-type load generates 3rd harmonic in load currents and is not practical for proposed filter
structure. However, the modified filter can be used if the 3rd
harmonic component is not compensated. In the case of RLtype diode rectifier load the amount of 3rd harmonic is minimal and the proposed filter structure can be used. In this case
the amount of total inductance needed in the modified dc filter
is decreased to 1/4 compared to the conventional dc filter.
VI. CONCLUSIONS
In this paper a new smaller dc filter structure is proposed for the current-source active power filter. In the
presented dc filter structure the energy of the most important harmonics are stored in resonant circuit which makes
it possible to decrease the overall size of the filter. After
simulations and experimental tests it was found that the
proposed filter structure works well with symmetrical
loads if the load current doesn’t contain 3rd harmonic
component.
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[1]
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1997.
[2]
A. Lindemann, “Characteristics and applications of a reverse
blocking IGBT”, PCIM Europe, pp.12-16, January-February 2001.
[3]
S. Fukuda and T. Endoh, “Control method and characteristics of active power filters”, 5th European Conference on Power Electronics
and Applications, Vol 8, pp. 139-144, 1993.
[4]
S. Fukuda and T. Endoh, “Control method for a combined active filter
system employing a current source converter and a high pass filter”,
IEEE Trans. Ind. App., Vol. 31, No. 3, pp. 590-597, 1995.
[5]
M.-X W ang and H. Pouliquen, “Performance of an active filter using
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Electronics and Applications, Vol. 8, pp. 218-223, 1993.
[6]
M. Salo and H. Tuusa, H., “A novel open-loop control method for a
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