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DESIGNING w/ THE HMC414MS8G PA UTILIZING
A LOW COST LAMINATED PRINTED CIRCUIT BOARD
General Description
The HMC414MS8G is a high efficiency GaAs InGaP Hetrojunction Bipolar Transistor (HBT) MMIC power
amplifier, which operates between 2.2 - 2.8 GHz. The amplifier is packaged in a low cost, surface mount 8
lead package with an exposed base for improved RF and thermal performance. With a minimum of external
components, the amplifier provides 20 dB of gain, +30 dBm of saturated power at 32 % PAE from a +5.0 V
supply voltage. The amplifier can also operate with a 3.6 V supply. Vpd can be used for full power down or
RF output power/current control.
Application Problem
Figure 1 - HMC414MS8G evaluation board
15
APPLICATION NOTES
An evaluation board from Hittite is provided in order to demonstrate the full performance of the HMC414MS8G
power amplifier that uses Rogers high frequency laminate, RO4350. This laminate is specifically designed
for RF/microwave circuits and displays stable properties over a broad range of environmental conditions.
The board topology, shown in figure 1, is single layer with a ground plane height of 10 mils. The RF input
and output circuitry is comprised of 50 Ω grounded coplanar transmission line with vias running along the
transmission line to provide ground to the top metal. Although this design provides excellent performance
at microwave frequencies, it may not be practical for many applications due to cost and level of integration.
For this reason, board materials such as FR4, BT and GETEK are often used due to their low cost and
multilayer properties. Grounded coplanar transmission lines also present a problem due to the requirement
of ground vias along the line. In many multilayer boards, these vias would occupy valuable real estate,
which is better suited for components. In order to address these issues, Hittite has designed an evaluation
board utilizing a single layer FR4 board with microstrip line used as the transmission line topology. Although
not multilayer, this design procedure can easily be adopted for many board topologies and materials. This
product note will discuss the methodology used in the design.
Evaluation Circuit Composition
The HMC414MS8G evaluation board uses grounded coplanar transmission line for the RF input and output.
The difference between grounded coplanar and microstrip is the distribution of the electric field. Figure
2 shows a comparison of the electrical field distribution between grounded coplanar and microstrip. The
electric field lines in grounded coplanar are confined mainly between the transmission line and top layer
ground although some of the field is distributed to the lower ground plane. In microstrip, the majority of the
field is confined between the transmission line and ground with fringing fields off to the side. It is this field
distribution which affects the impedance of the transmission line. Since the majority of the field in grounded
For price, delivery, and to place orders, please contact Hittite Microwave Corporation:
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15 - 21
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DESIGNING w/ THE HMC414MS8G PA UTILIZING
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coplanar is coupled across the slot, it is the width of the slot that sets the impedance. This allows for the
width of the transmission line to be either wide or thin, depending on the particular application. This is not
the case in microstrip where width of transmission line is dictated by the dielectric constant and substrate
height.
Figure 2 - (a) Grounded coplanar transmission line, (b) Microstrip transmission line
APPLICATION NOTES
15
In order to demonstrate these differences, the line width, impedance of the grounded coplanar, and
microstrip lines are calculated using a transmission line synthesizer tool, TLINE1. The board material is FR4
at a height of 62 mil and a εr=5.4. For a 50 Ω line impedance, a grounded coplanar line could be realized
with a 6-mil gap and a 21-mil transmission line or a 9-mil gap and a 33 mil transmission line. A 50-Ω
microstrip line on the same material requires a 113-mil transmission line width. This line width is excessive
and presents a problem in most board layouts when interfacing to the amplifier I.C. Reducing the line
width, from 113 mil to 35 mil (a more manageable width), increases the impedance from 50 Ω to 82 Ω. This
increased impedance will affect the performance of the amplifier and mismatch the system. If microstrip line
is to be used, the board height of the FR4 must be reduced or external circuitry must be added to transform
the impedance back to 50 Ω. In this application, simple lowpass matching networks are used to transform
the impedance.
Via holes are used in Printed Circuit Boards (PCB) to provide both RF and DC ground. While the ideal via
is a perfect ground, the physical via contains parasitic inductance, resistance and capacitance to ground as
shown in figure 3.
Figure 3 - Physical via model
The parasitic capacitance is the parallel plate capacitance of the via pad and the ground plane and is
approximated by the equation:
A=area of top pad
εο=permittivity of free space (8.85*10-12 Farads/meter)
εr= relative permittivity
D= distance from via pad to ground plane
15 - 22
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A LOW COST LAMINATED PRINTED CIRCUIT BOARD
This capacitance is relatively low (< 0.05 pF) and, coupled with the operating frequency, will not dominate
in the overall reactance of the via. However, the inductance and the resistance are factors in high power
applications. The following equation can be used to estimate the via inductance2:
Where r is the radius of the via and h is the substrate height. From the equation it can be seen that as the
height of the substrate increase so does the via inductance. Using the above equation the inductance of
the 62-mil FR4 board and the 10 mil Rogers 4350 board is calculated. The calculated inductance for the
Rogers 4350 is approximately, 0.016 nH, where the FR4 inductance is 0.43 nH. In addition, the resistance is
higher in the FR4 board via since DC resistance is directly related to the length. DC resistance is calculated
by the equation:
where h is the length of the via, A is the cross sectional area, and σ is the electrical conductivity of the
metal.
The increased parasitics must be compensated for when laying out the circuit for the FR4 board. The
overall inductance and resistance is reduced by using multiple vias in close proximity to all of the shunt
components.
15
Typically, matching networks for power amplifiers are designed using load pull data measured from a
particular amplifier or device. However, many times, this data is either not available or difficult to obtain
due lack of equipment or time. Hittite has tuned the output matching networks to its power amplifiers for
maximum output power and efficiency. Using these known load impedances, it is possible to design an
equivalent matching network using different circuit topologies or board materials. Figure 4 shows the output
load portion of the HMC414MS8G evaluation board. The output connector is terminated in a 50-Ω load and
the device is removed to allow probing of the output-matching network. By placing the probe as close to the
device output pin reference plane and ground plane as possible (see figure 4), the probe will be positioned
to achieve the best ground-signal-ground as possible.
APPLICATION NOTES
Designing The Output Matching Network From A Known Load
Figure 4 - HMC414MS8G evaluation board output load probe.
The probe is calibrated for a one-port measurement using calibration standards provided by the probe
For price, delivery, and to place orders, please contact Hittite Microwave Corporation:
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15 - 23
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DESIGNING w/ THE HMC414MS8G PA UTILIZING
A LOW COST LAMINATED PRINTED CIRCUIT BOARD
1.6
1.4
1.2
0.9
0.8
0.7
1.0
manufacturer. The output load measurement is shown in Figure 5.
1.8
5
0.
2.
0
0.6
0.2
0.4
0.
4
0
3.
0.6
0.3
0.8
4.0
1.0
5.0
Measured Evaluation
Board Output Load
1.
0
0.2
0.8
0.6
10
0.4
0.1
20
20
10
5.0
4.0
3.0
2.0
1.4
1.8
1.6
1.2
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.2
0.1
0
0.3
0.2
50
50
0.2
20
0.4
0.1
10
0.6
0.8
0.2
0
1.
5.0
1.0
4.0
0.3
0.8
3.
0
0.6
4
0.
1.8
0.6
0
2.
0.
5
0.4
APPLICATION NOTES
15 - 24
1.4
1.2
1.0
0.9
0.8
0.7
1.6
0.2
15
Figure 5 - Measured output load of HMC414MS8G evaluation board
Table 1 lists the impedance of the load at three frequency points. Notice that the resistance is less than
10 Ω . This is a typical optimum load match for a power amplifier. Depending on the periphery of the
output device, the optimum load match is anywhere from 5 Ω to 20 Ω.
Frequency (GHz)
Resistance (Real Ω)
Reactance (Imaginary Ω)
2.2
9.3
-10.6
2.4
8.0
-8.9
2.8
5.6
-5.8
Table 1 - Measured output load impedance of HMC414MS8G evaluation board
HMC414MS8G FR4 Evaluation Board
FR4 board material is widely used throughout the industry in many consumer products because of its
low cost and multilayer capabilities. Because of its wide popularity, many companies are manufacturing
FR4 material. As a result, there is a large disparity in electrical properties owing to the wide range of
manufacturing processes between the companies. In addition, to maintain cost, the manufacturing process
is not held to a tight tolerance resulting in variations in loss tangent and dielectric constants. These variations
are noticeable above 3 GHz and are the reason why FR4 is seldom used beyond this frequency. When
performing a design using FR4 material it is essential to know the manufacturer and the electrical properties
for the material being used. If the source of the board material is unknown by the board manufacturing
house, then a specific board manufacturer should be requested. This request will avoid future difficulties
during testing. For this design, the specifications for the board material was requested and received. The
critical parameters are listed in table 2.
For price, delivery, and to place orders, please contact Hittite Microwave Corporation:
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FR4 Material Critical Parameters
Parameter
Specification
Units
Height (h)
62
mil
Permittivity (εr)
5.4
--
Loss Tangent (δ)
0.035
--
Copper Thickness (T)
1.4
mil
Table 2 - Critical FR4 board material specifications
Design of Output Matching Network
1.2
1.6
1.8
0.4
3.0
0.6
0.3
0.8
4.0
1.0
(1)
5.0
1.0
0.2
0.8
10
0.4
0.1
20
5.0
50
20
10
4.0
3.0
1.4
1.8
1.6
2.0
1.2
1.0
0.8
0.7
0.9
0.6
0.5
0.3
0.2
0
0.4
0.2
0.1
(2)
0.6
Ideal
50
0.2
20
0.4
0.1
10
0.6
L11
0.61 nH
C12
2.98 pF
0.8
0.2
1.0
5.0
1.0
4.0
0.8
Measured
0.6
3.0
0.4
1.8
0.6
2.0
0.5
0.4
1.4
1.2
1.0
0.9
0.8
0.7
1.6
0.2
Figure 6 - (a) Measured load data versus ideal, (b) Ideal matching network
Using the topology from the ideal matching network the final matching network is designed using the FR4
material previously described. The layout for the entire FR4 board is shown in Figure 7, which also includes
an input-matching network.
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Order Online at www.hittite.com
APPLICATION NOTES
0.5
2.0
0.6
0.2
0.4
0.3
15
1.4
0.9
0.8
0.7
1.0
Using the data obtained from the probing of the output load of the HMC414MS8G evaluation board, a
matching network for the FR4 board is designed. Initially an ideal matching network will be designed to
determine roughly the required inductance and capacitance. From Figure 5, it can be seen that a low pass
network will be adequate to match the 50 Ω output to the desired load impedance. Starting from 50 Ω,
a shunt capacitor will rotate along the constant admittance line intersecting the constant impedance line
where the output load is located. A series inductance is then required to move up the constant impedance
line. To simplify the process, a matching synthesis program is used to match a single frequency point from
Table 1.4 The initial synthesis yielded a result that has higher inductance than required. Reducing the
series inductor produced the results in Figure 6.
15 - 25
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DESIGNING w/ THE HMC414MS8G PA UTILIZING
A LOW COST LAMINATED PRINTED CIRCUIT BOARD
Figure 7 - The HMC414MS8G FR4 evaluation board
1.6
1.4
1.2
1.0
0.9
0.8
0.7
1.8
2.
0
0.6
5
0.
0.4
0
3.
0.6
0.3
0.8
4.0
1.0
5.0
1.
0
0.2
0.8
Simulated FR4 Output Load
0.6
10
0.4
0.1
20
50
20
10
5.0
4.0
3.0
2.0
1.4
1.8
1.6
1.2
1.0
0.9
0.8
0.7
0.6
0.5
0.4
0.2
0.1
0
0.3
0.2
50
0.2
20
0.4
0.1
10
APPLICATION NOTES
0.2
0.
4
15
Ideally, the RF traces into and out of the amplifier are 50 Ω. However, in this case, a 50 Ω line would be too
wide (113 mil). For this application, a more manageable line width is 35 mil, which has an impedance of 87
Ω. The high impedance line is used as a series inductor on the output-matching network. After tuning, the
value of the shunt capacitor is reduced to 1.5 pF. A 15-pF series blocking capacitor and an 18 nH choke is
added for DC bias. The capacitors used in the matching network are high Q microwave ceramic capacitors.
It is important to insure that the equivalent series resistance (ESR) of the capacitors is low to minimize loss.
The capacitors should always be operated below or at the series resonance. Above series resonance, the
reactance is inductive. In addition, the parallel resonance’s should be calculated to insure that they do not
fall in the band of operation. The above output matching was simulated using Eagleware’s Genesys5 basic
simulation software. S-parameter files from the capacitor and inductor vendors are used in the simulation
file to improve the accuracy of the model. The results of the simulation are compared to the probed output
load of the original evaluation board. As shown in Figure 8, there is good agreement between the simulated
and actual load data.
0.6
0.8
0.2
0
1.
5.0
1.0
4.0
0.3
0.8
3.
0
0.6
Measured R04350 Evalution
Board Output Load
4
0.
1.8
0.6
0
2.
0.
5
0.4
1.4
1.2
0.9
1.0
0.8
0.7
1.6
0.2
Figure 8 - Measured versus simulated output load
15 - 26
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Design of Input Matching Network
Normally, the HMC414MS8G power amplifier input is adequately matched. However, since high impedance
lines are being used, the input match is slightly degraded. To improve the input match, a low pass matching
network is used.
As with the output-matching network, an ideal input matching network is synthesized to establish a baseline
for the component values. The input impedance is determined with the output-matching network in place.
To shift the reference plane from the input of the evaluation board to the reference plane of the amplifier, an
electrical delay is applied to the network analyzer. The amount of delay was determined by placing a short
at the input reference plane of the amplifier (See Figure 9). A measurement was taken using an HP8753D
Network Analyzer. A matching network is then synthesized to match 34.6-j24.9 Ω to 50 Ω.
A matching synthesis program is utilized to determine the optimum input matching network for a frequency
range of 2.2 GHz to 2.8 GHz. The ideal input matching network consists of a shunt 0.9-pF capacitor and
a series 3.3 nH inductor. Figure 10 (a) shows the simulated input match versus the input match with no
matching network.
Figure 9 - Determining the reference plane of the input
No Input Matching Versus Input Matching
0
RETURN LOSS (dB)
-10
No input matching,
measured data.
L1
3.3 nH
(1)
-20
-30
(2)
C1
0.9 pF
APPLICATION NOTES
15
Ideal matching network.
-40
-50
2200
2300
2400
2500
2600
2700
2800
FREQUENCY MHz
Figure 10 - (a) Matched input versus non matched input, (b) Ideal input matching network
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Figure 7 shows the input match as it is implemented on the FR4 board. During test, it was determined that
the optimum value for the shunt capacitor is 0.5 pF instead of 0.9 pF. The value of the inductor remained
the same.
The HMC414MS8G power amplifier input is in line with its output separated by a single pin. Unless the
pin is properly grounded, feedback from the output will cause the amplifier to oscillate. Figure 11 shows
pin 2 well grounded with the ground plane tapering up to the pin. This will provide the necessary isolation
between the input and output to prevent potential oscillations.
Measured Results
The board was measured for gain, return loss, 1 dB compression, saturated output power and efficiency.
These measured results are shown in figures 12,13,14 and 15, respectively. The results are compared
to HMC414MS8G evaluation board, which is constructed on Rogers 4350 high frequency laminate. Bias
voltage is set to 3.6 V resulting in a quiescent current of approximately 240 mA.
In order to accurately assess the performance of the circuit, the losses from the transmission line and
connectors are de-embedded from the initial measured results. These losses are calculated up to, but
not including the matching networks. Figure 11 shows the reference planes from which the losses are
calculated.
APPLICATION NOTES
15
Figure 11 - Reference planes of calculated board loss
A simplified model of the connector is used to determine its losses. The simple model consists of a length
of coaxial line, which is dimensionally equivalent to the connector. The losses due to the transmission
line are calculated using =EMPOWER=6, an electro-magnetic simulator. Table 3 lists the magnitude of the
losses. Notice that the output loss does not increase with higher frequency. This is due to the bias choke,
which has a higher reactance at the upper frequencies.
Board Loss
Frequency (GHz)
Input Loss (dB)
Output Loss (dB)
Total Loss (dB)
2.2
0.53
1.00
1.53
2.4
0.58
1.00
1.59
2.8
0.72
0.92
1.65
Table 3 - Simulated board losses
15 - 28
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HMC414 FR4 vs HMC414 Rogers 4350 Evaluation
Board Gain, Vdd= 3.6V
30
HMC414 FR4 vs HMC414 Rogers 4350 Evaluation
Board Return Loss, Vdd= 3.6V
0
Rogers 4350 Board, H=10mil
FR4 Board, H=62mil
-5
RETURN LOSS (dB)
GAIN (dB)
25
20
Rogers 4350 Board, H=10mil
15
-10
-15
FR4 Board, H=62mil
-20
2.3
2.4
2.5
2.6
2.7
-25
2.2
2.8
2.3
2.4
FREQUENCY (GHz)
2.5
2.6
2.7
2.8
FREQUENCY (GHz)
Figure 12 - Gain, FR4 board versus
HMC414MS8G evaluation board
Figure 13 - Return loss, FR4 versus
HMC414MS8G evaluation board
PSAT and P1dB vs Frequency Vdd= 3.6V
30
45
Psat,FR4, H=62mil
Power Added Efficiency vs Frequency
Vdd= 3.6V, Pout Saturated
40
28
FR4, H= 62mil
15
Psat, Rogers 4350, H=10mil
PAE (%)
Pout (dBm)
35
26
24
30
25
20
Rogers 4350, H=10mil
22
P1dB, Rogers 4350, H=10mil
15
P1dB,FR4, H=62mil
20
2.2
2.3
2.4
2.5
2.6
2.7
2.8
10
2.2
2.3
2.4
FREQUENCY (GHz)
Figure 14 - P1dB and saturated power, FR4
versus HMC414MS8G evaluation board
2.5
2.6
2.7
Figure 15 - Power Added Efficiency (PAE), FR4
versus HMC414MS8G evaluation board
Table 4 lists the specifications for the HMC414MS8G power amplifier as set in the Hittite data sheet.
Comparing the measured results to the table shows that the FR4 board design meets or exceeds the
minimum requirements.
Vdd= 3.6V
Parameter
Min.
Spec / FR4
Frequency Range
Typ.
Max.
Spec / FR4
2.2-2.8
Gain
Units
GHz
17 / 19
20
25 / 26
3 / 31
8
-- / 24
dB
21 / 21.5
25
-- / 25
dBm
Saturated Output Power (Psat)
23 / 23.5
27
-- / 28
dBm
Supply Current (Icc)
240 / 240
--
--
mA
Input Return Loss
Output Power for 1dB Compression (P1dB)
Vpd=3.6V
2.8
FREQUENCY (GHz)
APPLICATION NOTES
10
2.2
dB
Notes: Minimum occurs at 2.7GHz
Table 4 - Electrical specifications, Ta=+25° C, Vpd=3.6V
For price, delivery, and to place orders, please contact Hittite Microwave Corporation:
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Thermal Considerations and Analysis
One parameter that is often overlooked in amplifier design is thermal relief. Many times a part will be
designed with excellent initial performance only to fail later due to excessive temperature. These failures
can be avoided by following a few simple steps. Figure 16(a) shows a transparency of the HMC414MS8G
in its MS8G package. Beneath the package, there are several vias with a diameter of 15 mils. It is best to
have additional smaller vias under the package instead of fewer larger vias. Smaller vias are more easily
filled with solder (less voids) allowing for a more efficient transfer of heat. In addition, thermal resistance
from the flange to ground is inversely proportional to the number of vias.
15
APPLICATION NOTES
Figure-16 - (a) Via hole placement beneath flange, (b) simplified thermal model
Figure 16(b) shows a simple thermal model of a part mounted on a board. It consists mainly of three thermal
resistances, from the junction to the flange (TRjf), the flange to the bottom of the board (TRvia), and from
the bottom of the board to air (Trambient). The thermal resistance from the junction to the flange is typically
supplied by the manufacturer and in the case of the HMC414MS8G, it is 37 °C/W at a maximum flange
temperature of 85 °C. Trambient is a function of copper thickness, surface area, and surface radiation.
Determination of TRambient is a complicated process and therefore beyond the scope of this application
note. However, the thermal resistance of the via is approximated with the following equation:
Where, h=length of the via (cm),
=thermal conductivity of the via metal (W/cm-K), Router=outer radii of
the via (cm), and Rinner=the inner radius of the via (cm). From the equation, two observations are made;
1) longer vias have higher resistance and 2) filled vias have lower resistance than open vias. It is the
latter observation which highlights the benefits of filling the vias with solder. Finally, the number of vias will
decrease the resistance by 1/N, where N is the number of vias. An example is in order to demonstrate the
application of the above equations.
Example:
It was previously mentioned that the thermal resistance of the HMC414MS8G is 37 °C/W. This information
is extracted from the Hittite data sheet for this particular part. However, this information may not be obvious
and, as a result, warrants further explanation. The information supplied in the data sheet is in the form of
absolute ratings. The maximum channel temperature (junction temperature) for this device is 150 °C for a
continuous power dissipation of 1.73 W at a flange temperature of 85 °C. From this information the thermal
15 - 30
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resistance is calculated using the following equation:
Using the information presented in the data sheet results in a thermal resistance of 37 °C/W. The same
information can also be extracted by taking the inverse of the thermal conductance, which is also given in
the absolute maximum ratings section of the data sheet. For the HMC414MS8G the thermal conductance
is 27 mW / °C (Note: the thermal conductance in the data sheet is referred to as a “de-rating factor”).
Using the equation to calculate the via thermal resistance, the thermal resistance is computed to be 152 Ω.
This resistance is calculated for a via that is 0.062 ” long with a diameter of 0.015 ” and is filled with solder.
Figure 16(a), shows that nine vias reside directly under the package flange. These vias will contribute to
the majority of the heat transfer from the bottom of the package to the bottom of the board and are included
in the model. The heat transfer contribution of the remaining vias is complex and will not be included in the
model. The nine vias will reduce the thermal resistance of the via by 1/9 to 17 Ω.
A model for the HMC414MS8G mounted on the FR4 board is shown in Figure 17. The thermal resistance
from the back of the board to the surrounding air has been omitted since the temperature of the board was
allowed to reach steady state. The model is comprised of a constant current source, which is representative
of the dissipated power and the thermal resistance of the junction to flange (TRjf), and from the flange to the
bottom of the board (TRvia).
15
APPLICATION NOTES
In order to establish a reference temperature, a probe is placed on the bottom of the board directly below
the package flange. The amplifier is biased with 5 V with an input power of 15 dBm. The output power is 30
dBm with a DC current of 432 mA. The amplifier is left on until the temperature at the bottom of the board
reaches steady state. The temperature is observed to be 63.8 °C with a surrounding ambient temperature
of 27 °C. The amount of power dissipated as heat is calculated using the following equation:
Figure 17 - The HMC414MS8G amplifier on FR4 board, simplified thermal model
This simplified thermal model is analyzed in the same way a simple electronic circuit is analyzed by using
either Ohm’s or Kirchoff’s laws. For example, to determine the temperature of the flange
The following equation is used:
For price, delivery, and to place orders, please contact Hittite Microwave Corporation:
12 Elizabeth Drive, Chelmsford, MA 01824 Phone: 978-250-3343 Fax: 978-250-3373
Order Online at www.hittite.com
15 - 31
MICROWAVE CORPORATION
v00.0103
DESIGNING w/ THE HMC414MS8G PA UTILIZING
A LOW COST LAMINATED PRINTED CIRCUIT BOARD
In this particular case, the flange temperature is determined to be 84.2 °C. Making the appropriate
substitutions in the above equation yields a junction temperature of 128.6 °C. The analysis shows that the
junction temperature of the device is below the critical threshold of 150 °C. However, this is at an ambient
temperature of 27 °C which suggests that a heatsink would be required if the device is to be operated at
higher ambient temperatures. Using the same procedure the maximum ambient temperature, before the
junction exceeds 150 °C, is calculated to be approximately 49 °C.
The procedure and model used in the thermal analysis of the board should be used to establish baseline
conditions only. Thermal analysis is complex with many variables that will affect the outcome. Upon
establishing a baseline, a thorough thermal analysis should be performed to determine the optimum board
design.
Conclusion
APPLICATION NOTES
15
A power amplifier that originally was mounted and tuned on high frequency laminate is mounted on low
cost FR4 board utilizing high impedance microstrip lines. Using sound RF practices, a low pass outputmatching network is designed that presents the optimum load to the power amplifier. In addition, a low
pass matching network is synthesized that transforms the high impedance line at the input to the input
load impedance of the amplifier. A simple thermal analysis is performed which shows how to translate the
measured temperatures at the bottom of the PC board to the junction of the device. The techniques in this
product note can be extended to other Hittite power amplifiers by using load and temperature data provided
by Hittite Microwave Corporation.
(Endnotes)
1
=TLINE=, transmission line synthesizer software, Eagleware Corporation
2
Arbie, Peter, L.D., “Design of RF and Microwave Amplifiers and Oscillators”, Norwood, MA: Artech
House, 1999
4
=Match=, matching network synthesizer software, Eagleware Corporation
5
Genesys V8, RF and Microwave linear simulation software, Eagleware Corporation
6
15 - 32
=EMPOWER=, electromagnetic simulation software, Eagleware Corporation
For price, delivery, and to place orders, please contact Hittite Microwave Corporation:
12 Elizabeth Drive, Chelmsford, MA 01824 Phone: 978-250-3343 Fax: 978-250-3373
Order Online at www.hittite.com
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