Design of High-Efficiency Interior Permanent
Magnet Synchronous Machine with Stator Flux
Barriers and Single-Layer Concentrated
Windings
Volodymyr Bilyi, Dieter Gerling
Abstract-- This paper presents the design of a highperformance 14-pole, 12-slot interior permanent magnet
synchronous machine with flux barriers inside the stator iron
core and fractional-slot concentrated windings. This work
focuses on electromagnetic, mechanical and thermal aspects of
the machine design. The specific goal of this work was to
design an electrical machine, which will have a small size, high
power-to-weight ratio, high efficiency, especially at part-load
operating points, and high speed capability for traction drive
applications. Therefore the unconventional stator design is
used. The proposed machine design was compared with the
conventional one, concerning the magnetic saturation,
spectrum of the magneto-motive force and produced torque.
To ensure the high speed capability of the motor, the
numerical structural analysis of the rotor is shown.
Temperature behavior inside of the designed motor at steadystate operating point is shown.
Index Terms-- Electromagnetic analysis, finite element
analysis, materials, mechanical analysis, machines, permanent
magnet, thermal analysis, efficiency, electric vehicle,
concentrated winding.
I
I. INTRODUCTION
N the past years, significant improvements in the
production and quality of the electromagnetic materials
were achieved. Modern iron alloys can be used for different
purposes, i.e. to withstand a high mechanical stress, for
example due to the high rotational speed, or such to
minimize the amount of iron loss inside of the core at
certain frequencies. Also, modern rare-earth magnet
materials are becoming better, concerning the maximum
operating temperature and magnetic properties. The
enhancements in the battery production technology have
made them cheaper and increased their lifetime and energy
content. These achievements in the material production and
V. Bilyi is a research assistant at the Universitaet der Bundeswehr
Muenchen, Chair of Electrical Drives and Actuators, Neubiberg 85577,
Germany (e-mail: Volodymyr.Bilyi@unibw.de).
D. Gerling is Full Professor at the Universitaet der Bundeswehr
Muenchen, Chair of Electrical Drives and Actuators, Neubiberg 85577,
Germany (e-mail: Dieter.Gerling@unibw.de)
978-1-4799-7940-0/15/$31.00 ©2015 IEEE
also the recent scientific work on the improvement of
electrical machines have made them great candidates for
the application in electric and hybrid vehicles as a traction
drive.
Every single application has a certain specific set of
specifications. To use the electrical motor as a traction
drive in electric vehicle, it has to fulfill the main
requirements: high overall efficiency, especially in part
load, short active length, high power-to-weight ratio, high
flux-weakening capability on the wide constant-power
speed range, mechanical stability at high speed, low torque
pulsation, low vibration and noise levels, high fault
tolerance and low manufacturing cost.
Permanent magnet synchronous machines (PMSM)
commonly have the highest efficiency and power density
among all electrical machines. Despite the not directly
controllable intensity of the magnetic field due to
permanent magnets, flux-weakening operation is possible.
However, the interior PMSM (IPMSM) have a better fluxweakening capability [1]. Moreover, there is no need for
non-magnetic bandage fixation of the permanent magnets
(PM) due to a high rotational speed [2]. Also, IPMSM tend
to have less amount of magnet loss, due to the deeper
placement of the magnets inside of the rotor core [3].
There are two main winding topologies which are used
for stator excitation. These are the distributed and the
concentrated windings. In this work the concentrated
winding is considered.
The fractional slot concentrated windings (FSCW) are
widely used in different applications in order to increase the
copper fill factor inside of the stator slots and therefore to
minimize copper loss and also to reduce the machine length
in comparison with a distributed winding, due to its shorter
stator end-windings [4]. In addition, modular construction
of the stator core is possible in this case, so the simplified
manufacturing process can be realized [5], which directly
affects the production cost.
In spite of all advantages of the IPMSM with FSCW
there are several disadvantages, such as high number of
space harmonics with high amplitude, which results in
1177
vibration and noise.
To reduce the amplitude or even to completely avoid the
undesirable harmonics in the magneto-motive force (MMF)
spectrum, the novel stator design with flux barriers in stator
core was suggested in [6]. In [7] and [8] authors investigate
PM machines with stator flux barriers and validate the
results on a prototype.
This work focuses on the design of the IPMSM with
stator flux barriers and FSCW for traction drive application
to meet some specific set of requirements. Therefore
electromagnetic, mechanical and thermal analyses are
shown.
II. ELECTROMAGNETIC ANALYSIS
A. Requirements
A set of specifications which is typical for automotive
traction applications is shown in Table I.
TABLE I
SET OF REQIREMENTS
127
550
13000
7000
>95%
137
81
100
110
210
103
130
140
<5%
<135
<185
<190
<25
Fig. 1. Flux density distribution under the rated load conditions in the
conventional machine.
B. Machine Design
In order to get a good performance of the machine with
FSCW, the behavior of the slot and pole numbers have to
be carefully chosen. To simplify the manufacturing process,
a single-layer FSCW is considered. At the beginning of the
design process, a 28-pole, 24-slot topology was
investigated, due to a high fundamental winding factor of
0.966, which directly affects the machine performance.
However, through the relatively small specified stator
diameter this topology was inappropriate, therefore a 14pole, 12-slot IPMSM, with a similar fundamental winding
factor is selected. Every second tooth of the stator is
equipped with a non-overlapping 3-phase concentrated
winding. The number of slots per pole per phase is less than
one. The rotor is equipped with 14 NdFeB permanent
magnets. The following analysis was made for the ambient
temperature of 90°C.
To fulfill all requirements in Table I, the motor with
following data was designed. Steel M235-35A was chosen
as a material of the stator and rotor laminations, where
1
0.9
0.8
Normalized values
Max. phase voltage [VRMS]
Max. phase current [ARMS]
Max. speed [rpm]
Corner speed (rated torque) [rpm]
Efficiency at corner speed
Rated torque at ncorner [Nm]
Torque at nmax [Nm]
Rated power at ncorner [kW]
Power at nmax [kW]
Peak torque at ncorner [Nm]
Peak torque at nmax [Nm]
Peak power at ncorner [kW]
Peak power at nmax [kW]
Torque pulsation
Rotor outer diameter [mm]
Stator outer diameter [mm]
Motor active length [mm]
Total active weight [kg]
single lamination sheet has a thickness of 0.35 mm and the
saturation flux density of the material is 2T. Stacking factor
is set to 0.95. This steel shows relatively good
electromagnetic properties, concerning iron loss and
magnetic saturation, and at the same time very good
mechanical strength. NdFeB was picked as a material of
PMs with coercivity of 957.9 kA/m (at 90°C). Maximum
operating temperature of the magnets is 160°C. The
dimensions of a single magnet segment are 21x5 mm. Airgap thickness between the rotor and the stator is set to 0.9
mm. Stator winding fill factor equals 0.5, which could be
even higher in practice. The number of windings per coil is
set to 4. The active length of the motor is 170 mm. Stator
and rotor outer diameters are 180 and 132 mm,
respectively. Therefore, all geometry requirements (Table I)
were met.
To compare the conventional and proposed motor
designs, flux density distribution and spectrum of the
magneto-motive force are considered. Fig. 1 and 3 and Fig.
2 and 4 show the flux density inside of the machine iron
core at the rated load, and the normalized MMF spectrum
for both designs, respectively.
0.7
0.6
0.5
0.4
0.3
0.2
0.1
0
1
3
5
7
9
11
13
15
17
19
21
23
Harmonic order
Fig. 2. Air-gap magnitude under the rated load conditions in the
conventional machine.
There is a magnetic saturation in case of the first design
(Fig. 1). It is caused by different reasons. One of them is the
presence of harmonics with the lower order than the order
1178
investigated with a 2-D finite elements analysis (FEA). The
results are shown in Fig. 5. The average mechanical torque
production and torque pulsation in the conventional
machine are 176 Nm and 10.5% at the peak load (550A)
and corner speed, respectively. By adding the stator flux
barriers, the value of the average torque is increased by
17% and the pulsation of the torque is only 4.4%. Adding
the semicircular rotor notches has increased the torque by
5.1% and the pulsation of the torque is 2.1% at peak load.
At the rated load of 350A the average torque and the
pulsation of the torque are 137.7 Nm and 4.1%,
respectively.
220
215
210
205
Torque, T [Nm]
of working harmonic in the MMF spectrum and in the airgap (Fig. 2). In the case of 14-pole motor, the 7th harmonic
is the working harmonic. The harmonics of lower order do
not directly contribute to the torque production but have a
negative effect on the motor performance, causing reduced
torque with high pulsations, vibration and noise. In order to
reduce the amplitude of the undesired harmonics, an
alternative stator design is investigated. Fig. 3 shows the 2D model of the proposed machine. As can be seen, every
second tooth is equipped with the stator flux barrier. Nonpermeable, non-conductive material or air can be used to
produce the flux barriers. The rotor of the proposed
machine is also equipped with notches on both sides of
each magnet. This contributes to better flux focusing and to
avoidance of flux short-circuits in the rotor and has a great
effect on the mechanical stress reduction inside of the rotor.
In Fig. 4 the air-gap magnitude of the proposed machine is
shown. Amplitudes of the 1st and 5th harmonics are reduced
in comparison with conventional design by 54% and 32%,
respectively. Flux densities in the stator yoke and in the
rotor core in the proposed design are become lower and
have an optimal value (Fig. 3).
200
no barriers
rotor barriers
stator barriers
stator and rotor barriers
195
190
185
180
175
170
165
0
1
2
3
4
5
6
7
Time, t [ms]
Fig. 5. Effect of the stator flux barriers and rotor notches on the produced
torque.
Fig. 6 shows the n-M curves of the motor under peakload and rated-load conditions. It is seen that the
requirement concerning produced mechanical torque and
mechanical power are met.
220
200
ph
Average torque, Tavg [Nm]
Fig. 3. Flux density distribution under the rated load conditions in the
proposed machine.
1
0.9
Normalized values
0.8
0.7
160
140
120
ph
80
60
40
0.6
20
0.5
0
0
1
2
3
4
5
6
7
8
9
10
11
12
Rotational speed, n [krpm]
0.4
Fig. 6. n-M curves under the rated and the peak load conditions.
0.3
0.2
0.1
0
IRMS = 350A
100
170
160
150
140
130
120
110
100
90
80
70
60
50
40
30
20
10
0
13
Mechanical power, Pm [kW]
IRMS = 550A
180
1
3
5
7
9
11
13
15
17
19
21
23
Harmonic order
Fig. 4. Air-gap magnitude under the rated load conditions in the proposed
machine.
The effect of the stator flux barriers and rotor notches on
the machine performance concerning produced torque was
To see the effectiveness of the machine under different
load conditions, the efficiency map has to be computed. To
be able to compute the efficiency map on the whole
operation range of the designed motor the values of losses
in the machine are needed.
C. Copper Loss Estimation
By the copper loss calculation only DC ohmic loss is
1179
taken into account. The proximity and skin effects are not
considered. The amount of copper losses Pcu can be
expressed as a product of the phase current I to the power
of two and the phase resistance Rph, which depends on the
temperature and on the material properties.
Pcu = 3 ⋅ I 2 ⋅ R ph
(1)
temperature behavior of the motor. This leads to the axial
length of the single magnet segment of 6.8 mm.
Fig. 8 gives the information about the eddy-current loss
prediction inside the PMs under the different load
conditions and with the segmented PMs. There is 244 W
and 400 W eddy-current loss under the rated load (7000
rpm) and the peak load (6000 rpm) conditions, respectively.
The amount of copper loss at the rated and at the peak
load conditions are 1.2 kW and 3 kW, respectively. Endwindings are also taken into account. The calculation is
made for the temperature of 90°C.
1.5
Eddy-current loss in PM, PM [kW]
I
D. Iron Loss Prediction
For the iron loss estimation the numerical 2-D FEA
model is used. The core loss PFe can be represented as a
sum of hysteresis Ph, eddy-current Pc and excess Pe core
loss:
PFe = Ph + Pc + Pe
(2)
PFe = k h fB 2 + k c f 2 B 2 + k e f 1.5 B 1.5
RMS
1.35
= 550A
ph
IRMS = 450A
ph
1.2
IRMS = 350A
ph
1.05
I
RMS
= 250A
ph
0.9
0.75
0.6
0.45
0.3
0.15
0
(3)
0
1
2
3
4
5
Where B is the amplitude of the AC flux component, f is
the frequency, kh is the hysteresis core loss coefficient, kc is
the eddy-current core loss coefficient and ke is the excess
core loss coefficient.
However, the 2-D FEA model does not allow
considering the axial segmentation of the rotor PMs during
the eddy-current loss computation inside the magnets. To
take the segmentation into account and to investigate the
dependency of the eddy-current loss due to the PMs on the
number of magnet segments, for the specific axial machine
length, a 3-D numerical model is made. Fig. 7 shows the
reduction of the eddy-current loss in the magnets with
increased number of magnet segments. This effect is similar
to the effect of the stator and rotor lamination. The
estimated iron loss values are 1.51 kW and 1.59 kW under
the rated load (350 A and 7000 rpm) and peak load (550 A
and 6000 rpm) conditions, respectively.
Eddy-current loss in PM, PM [p.u.]
7
8
9
10
11
12
13
Fig. 8. Eddy-current loss due to the PMs versus rotational speed under the
different load conditions.
E. Air-friction Loss Estimation
The investigation of the mechanical friction loss of the
electrical machine with high rotational speed is usually
needed. This work considers only air-friction loss inside of
the air-gap, due to the rotation of the rotor. Other
mechanical losses are not taken into account. Windage loss
can be derived as a product of the coefficient of the surface
roughness kR (1 for smooth surface), friction coefficient CR,
density of the medium ρ (1.2041 kg/m3 for air at 20°C),
angular velocity of the rotor in the power of three, outer
radius of the rotor Rro in the power of for and axial length
of the rotor Lst [9-11].
PW = k R C R ρπω 3 R ro4 Lst
(4)
where:
1
CR =
0.9
0.8
0.0152
Re δ
(5)
and:
0.7
0.6
Re δ =
0.5
0.4
ρωRro δ
μ
(6)
Where δ is the air-gap thickness and μ is the dynamic
viscosity of the air. The calculated value of the air-friction
loss is 61 W at the rotational speed of 13000 rpm.
0.3
0.2
0.1
0
6
Rotational speed, n [krpm]
0
5
10
15
20
25
30
35
40
Number of segments, NS
Fig. 7. Eddy-current loss vs. the number of axial magnet segments.
For the design of the proposed machine, 25 magnet
segments was chosen in order to reduce the eddy-current
loss inside the PMs, which is very important for
F. Efficiency map computation
After the loss calculation was done, it is possible to
compute the efficiency map of the proposed motor under
different load conditions. Fig. 9 shows the efficiency map
of the designed motor. There is a big area of the highest
1180
efficiency of 97% and high part-loadd efficiency. The
requirements concerning efficiency are met.
concerning main mechanical properties of the rotor
materials, which are used for thhe structural 2-D FEA.
LE II
TABL
PROPERTIES OF RO
OTOR MATERIALS
Material
Steel
M235-35A
Permanent magnets
Fig. 9. Efficiency map of the designeed motor.
G. 3-phase Short-circuit Simulation
Despite having single-layer non--overlapping and
therefore fault-tolerant concentrated windings, it is
important to investigate the machine beehavior underfault
conditions. In this work a symmetrical 3-pphase short-circuit
is simulated. Fig. 10 shows the producedd torque under the
short-circuit conditions versus rotational speed.
s
[p.u.]
Value
7600 kg/m3
185 GPa
0.28
421 - 460 MPa
532 - 580 MPa
7700 kg/m3
140 GPa
0.24
600 MPa
After defining the material properties, the simulation is
performed. Temperature and magnetic
m
effects are not taken
into account. The maximum mechanical
m
stress is achieved
at a rotational speed of 156000 rpm, which is 120% of the
maximum rotational speed of the
t motor. It is seen, than the
value of the stress is 448.1 MPa,
M
which is lower than the
maximum value of the yield strength (460 MPa) for the
chosen material, so there is onnly elastic deformation (Fig.
11). Beside the rotational speedd and the material properties,
following parameters are imporrtant:
•
•
0
•
•
avg
-0.2
Short-circuit torque, T K
Phys. Properties
P
mass dennsity
elastic modulus
m
Poisson‘s ratio
yield streength
tensile sttrength
mass dennsity
elastic modulus
m
Poisson‘s ratio
compressive strength
Outer rotor radius
Size of the iron briddge between the magnets and
the air-gap
Size of the magnetss
Shape of the notchees and their radius
-0.4
-0.6
-0.8
-1
0
1
2
3
4
5
6
7
8
9
10
11
12
13
Rotational speed, n [krpm]
Fig. 10. Produced torque in case of the 3-phaase short-circuit.
The maximum torque of 80 Nm is produced at the
rotational speed of 95 rpm. This value coould be critical for
vehicle applications and should be takenn into account by
adding fault detection systems.
III. MECHANICAL ANALYSIS
Because of the centrifugal forces due to the high
rotational speeds of the rotor and presencee of the permanent
magnets inside of the rotor with a higher mass density than
the mass density of the iron, the mechhanical analysis is
needed, to ensure that the mechanical sttress inside of the
rotor does not exceed the limit. Also, the machine has to be
capable of an overspeed of 120% of
o the maximum
rotational speed [12, 13]. Table II givees an information
Fig. 11. Mechanical stress distributioon over the notch surface, under the
steady-state rotational speed of 15600 rpm.
r
The mechanical stress at 7000 rpm and at 13000 rpm is
90.4 MPa and 311.1 MPa, respectively.
o the effect of the rotor
For better understanding of
notches, the comparison betw
ween two rotor topologies is
made. The results are shown inn the Fig. 12. It can be seen
that the maximum rotational speed of the rotor without
notches, before the plastic defformation, is only 8600 rpm,
which is slightly above the rateed speed of 7000 rpm. That is
why the design without notches is inappropriate in this
1181
case. The proposed design of the rotor makes it possible to
meet the requirements concerning the rotational speed.
200
180
160
550
Temperature, t [°C]
Von Mises stress, σ [MPa]
500
Plastic deformation
450
400
Elastic deformation
350
without notches
300
250
120
100
80
60
40
200
with notches
150
20
100
0
8.6 krpm
50
0
140
0
1
2
3
4
5
6
7
8
9
10 11 12 13 14 15
Yoke
Coil side
End winding
Teeth
Rotor
Fig. 13. Steady-state temperature inside of the motor parts under the rated
load condition.
16
Rotational speed, n [krpm]
Fig. 12. Mechanical stress inside of the rotor versus rotational speed.
V. WEIGHT CALCULATION
IV. THERMAL ANALYSIS
The last but not least step in the design process was the
investigation of the temperature behavior inside the
designed motor.
The temperature prediction is made with a lumped
element model. Lumped element models are widely used
for temperature estimation during the design process. These
models have shown very accurate results and are very timesaving in comparison with FEA methods [14-17].
As a prerequisite for this investigation, the loss
calculation is done as described above. It is assumed that
there are five main loss sources and therefore five heat
sources inside the machine. These are the stator yoke, the
stator teeth, coils, which divided into coil sides and endwinding, rotor with PMs. The parameters of the model are
described as thermal resistor, thermal capacitances, heat
sources and compensation elements to achieve more
accurate results [14]. A conventional cooling jacket is
assumed as heat sink.
Following assumptions are made:
• Internal heat is evenly distributed in the bodies
• Axial heat flow appears only in the shaft
• Ambient temperature is set to 75°C
• Inlet temperature is set to 75°C
Fig. 13 shows the result of the steady-state temperature
estimation for the different regions in the machine under the
rated load conditions. The maximum temperature of 190°C
occurs in the end-winding. The high conductor insulation
class is required. The maximum operating temperature of
the PMs is not exceeded.
The estimation of the temperature ruse in the machine
under the peak load conditions is also done. The simulation
has shown that the designed machine could operate up to 30
seconds under the peak load conditions at 6000 rpm.
The proposed motor makes possible the significant
weight and size reduction in comparison with the
conventional machines due to the stator flux barriers and
short end-windings which make this electrical machine a
great candidate for the application as traction drive.
Table III shows the results of the machine active weight
calculation. The total active machine weight of the motor is
almost 21.1 kg. Concerning the peak power of 157.3 kW
which is produced under peak load conditions at a
rotational speed of 9500 rpm, the peak power-to-weight
ratio is 7.46 kW/kg which is very high. The peak-powerratio at the rated speed is 4.7 kW/kg.
TABLE III
MACHINE WEIGHT
Stator weight (stack of sheets), mS [kg]
Copper weight, mcu [kg]
Rotor weight (stack of sheets), mR [kg]
Magnet weight , mM [kg]
Total weight, mΣ [kg]
Peak power-weight-ratio, [kW/kg]
Rated power-weight-ratio, [kW/kg]
9.86
2.63
6.65
1.92
21.07
7.46
4.7
VI. CONCLUSION
The amount of electric vehicles is becoming higher. In
order to fulfill all requirements to the electric drivetrain,
new technologies have to be investigated and implemented.
This work shows a possible light-weight, high-efficient,
compact and cost-effective solution.
Firstly the comparison between the IPMSM with
conventional and proposed stator was presented. The
saturation effect inside of the stator was discussed. The
proposed design has shown better results concerning core
saturation and produced torque.
Then the loss estimation was made and an efficiency
map was computed. The motor has shown a big area of
highest efficiency and also high part-load efficiency.
1182
The next part of the design process was the structural
analysis of the rotor and the investigation of the influence
of the rotor notches on the produced mechanical stress.
At the end, a lumped element model is made for the
temperature rise estimation inside the designed machine.
And the active weight of the machine is calculated.
The proposed motor has met the entire requirements and
therefore is a great alternative to the conventional motors
which are widely used nowadays.
REFERENCES
[1]
[2]
[3]
[4]
[5]
[6]
[7]
[8]
[9]
[10]
[11]
[12]
[13]
[14]
[15]
[16]
[17]
T. M. Jahns, “Flux-weakening regime operation of an interior
permanent-magnet synchronous motor drive”, IEEE Trans. Ind.
Appl., vol. IA-23, no. 4, July/Aug. 1987.
T. Schneider, T. Koch, A. Binder, “Comparative analysis of limited
field weakening capability of surface mounted permanent magnet
machines”, in Proc. Elec. Power Appl., vol. 151, no. 1, pp. 76-82,
Jan. 2004.
P. B. Reddy, A. M. El-Refaie, K. -K. Huh, J. K. Tangudu, T. M.
Jahns, “Comparison of interior and surface PM machines equipped
with fractional-slot concentrated windings for hybrid traction
applications”, IEEE Trans. Energy Convers., vol. 27, no. 3, pp. 593602, Sep. 2012.
A. M. EL-Refaie, “Fractional-slot concentrated-windings
synchronous permanent magnet machines: opportunities and
challenges”, IEEE Trans. Ind. Electron., vol. 57, no. 1, pp. 107-121,
Jan. 2010.
H. Akita, Y. Nakahara, N. Miyake, T. Oikawa, “New core structure
and manufacturing method for high efficiency of permanent magnet
motors”, Ind. Appl. Conf., vol. 1, pp. 367-372, Oct. 2003.
G. Dajaku, D. Gerling, “Low costs and high-efficiency electric
machines”, EDPC-Konferenz, Nürnberg, 2012.
G. J. Li, Z. Q. Zhu, W. Q. Chu, M. P. Foster and D. A. Stone,
“Influence of flux gaps on electromagnetic performance of novel
modular PM machines”, IEEE Trans. Energy Convers., vol. 29, no.
3, pp. 716-726, Sep. 2014.
G. J. Li, Z. Q. Zhu, M. Foster and D. Stone, “Comparative studies of
modular and unequal tooth PM machines either with or without tooth
tips”, IEEE Trans. Magn., vol. 50, no. 7, July 2014.
A. Borisavljevic “Limits, modeling and design of high-speed
permanent magnet machines”, Ph.D. dissertation, Dept. Elect. Eng,
Math. Comp. Sc., Delft Univ. of Tech., Delft, The Nerherlands,
2013.
F. Magnussen, “On design and analysis of synchronous permanent
magnet machines for field-weakening operation in hybrid electric
vehicles”, Ph.D. dissertation, Dept. Elect. Eng., Royal Inst. of Tech.,
Stockholm, Sweden, 2004.
J. Saari, “Thermal analysis of high-speed induction machines”, Ph.D.
dissertation, Labor. of Electr., Helsinki Univ. of Tech., Helsinki,
Finland, 1998.
A. Binder, T. Schneider, M. Klohr, “Fixation of buried and surfacemounted magnets in high-speed permanent-magnet synchronous
machines”, IEEE Trans. Ind. Appl., vol. 42, no. 4, pp. 1031-1037,
July/August 2006.
E. C. Lovelace, T. M. Jahns, T. A. Keim, J. H. Lang, “Mechanical
design considerations for conventionally laminated, high-speed,
interior PM synchronous machine rotors”, IEEE Trans. Ind. Appl.,
vol. 40, no. 3, pp. 806-812, May/June 2004.
G. Dajaku, “Electromagnetic and thermal modeling of highly utilized
PM machines”, Ph.D. dissertation, Universität der Bundeswehr,
Neubiberg, Deutschland, 2006.
B. Kipp, “Analytische Berechnung thermischer Vorgänge in
permanentmagneterregten
Synchronmaschinen”,
Dissertation,
Universität der Bundeswehr, Hamburg, Deutschland, 2008.
D. Gerling, G. Dajaku, “Thermal calculation of systems with
distributed heat generation”, International Conference for Thermal,
Thermomechanical and Emerging Technology, pp. 645-652, San
Diego (CA), USA, May/June 2006.
G. Dajaku, D. Gerling, “An accurate electromagnetic and thermal
analysis of electric machines for hybrid electric vehicle application”,
The 22nd International Battery, Hybrid and Fuel Cell Electric
Vehicle Symposium & Exposition, pp. 1198-1207, Yokohama, Japan,
Oct. 23-28, 2006.
BIOGRAPHIES
Volodymyr Bilyi; M.Sc. Volodymyr Bilyi was born in 1990. He received
his Master degree in Electrical Engineering from the Otto von Guericke
University Magdeburg, Magdeburg, Germany in 2013. Since 2013 he
works as Research Assistant at the Institute of Electric Drives and
Actuators at the Universitaet der Bundeswehr Muenchen, Neubiberg,
Germany.
Dieter Gerling; Prof. Dr.-Ing. Dieter Gerling was born in 1961 in
Menden/Sauerland, Germany. He received his diploma and Ph.D. degrees
in Electrical Engineering from the Technical University of Aachen,
Aachen, Germany in 1986 and 1992, respectively. From 1986 to 1999 he
was with Philips Research Laboratories, Aachen, Germany as Research
Scientist and later as Senior Scientist. In 1999 he joined Robert Bosch
GmbH, Buehl, Germany, as Director. Since 2001 he has been Full
Professor at the Universitaet der Bundeswehr Muenchen, Neubiberg,
Germany.
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