IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
1595
A Novel Soft-Switching Multiport Bidirectional
DC–DC Converter for Hybrid Energy Storage System
Zhihui Ding, Chen Yang, Zhao Zhang, Cheng Wang, and Shaojun Xie
Abstract—A novel multiport isolated bidirectional dc–dc converter for hybrid battery and supercapacitor applications is presented, which can achieve zero voltage switching for all switches in
the whole load range. The bidirectional power flow between any
two of the ports is free, and the circulating power is low for the
well matching of the transformer voltages of all time regardless of
the voltage variations of the battery and supercapacitor. Moreover,
the current ripples are greatly decreased by interleaved control,
which is good for battery and supercapacitor. The converter topology and the operation principle are introduced. Detailed analysis
on soft-switching of all switches is given. On the basis of theoretical analysis, the principle and method for parameter designing
are provided. A hybrid energy management strategy combining
bus voltage control and energy management of the energy storage
devices is proposed and the control scheme is presented. Moreover, detailed parameter design of a prototype converter is given
for a 380-V dc-bus microgrid lab system. Effectiveness of the control strategy, correctness of the analysis on soft-switching, and the
parameter design methods are verified by the simulation and experimental results.
Index Terms—DC-bus microgrid, hybrid energy management,
hybrid energy storage, multiport bidirectional converter, softswitching.
I. INTRODUCTION
ITH the rapid development of electric vehicles [1], [2],
[9], [14], [15], [18] and distributed renewable energy
power systems [3]–[6], [12], [13], [17], microgrid (ac or dc)
gradually becomes the direction of distributed generation application [19], [20]. As an essential part of the microgrid, an energy
storage system draws wide attention. Battery is the most mature
W
Manuscript received December 3, 2012; revised March 7, 2013; accepted May
9, 2013. Date of current version October 15, 2013. This work was supported
by the National Nature Science Foundation of China under Grant 51077070,
the Science and Technology Support program of Jiangsu Province, China under
Grant BE2010188, Shenzhen Huntkey Chi Yuan Electric Company, Ltd., and
Nanjing Megampere Electrical Technology Company, Ltd. Recommended for
publication by Associate Editor F. Peng.
Z. Ding was with the College of Automation Engineering, Nanjing University of Aeronautics and Astronautics, Nanjing 210016, China. She is now with
Delta Electronics (Shanghai) Company Ltd., Shanghai 201209, China (e-mail:
dingzhihui1988@hotmail.com).
C. Yang, Z. Zhang, and S. Xie are with the College of Automation Engineering, Nanjing University of Aeronautics and Astronautics, Nanjing 210016,
China (e-mail: yangchen_1985@163.com; 275591557@qq.com; eeac@nuaa.
edu.cn).
C. Wang was with the College of Automation Engineering, Nanjing University of Aeronautics and Astronautics, Nanjing 210016, China. He is now
with Texas Instruments Semiconductor Technologies (Shanghai) Company Ltd.,
Shanghai 200003, China (e-mail: wii1225520@gmail.com).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2013.2264596
style of energy storage. It has advantages of high energy density,
long lifespan, low initial cost, and disadvantages of low power
density, slow dynamic response, which makes it unable to satisfy the requirements of sudden change on load or power. Meanwhile, a supercapacitor has short charging/discharging time and
a long cycle life. It is also easy to be well maintained. Therefore,
the hybrid system combined with the battery and the supercapacitor is often used to improve the performance of the energy
storage system. Battery is used for energy storage, while supercapacitor is used for transient energy storage and fast supplement. However, to ensure the hybrid system to be high-efficient,
safe, reliable, and running steady, the design needs be considered
from two aspects: the construction and the energy management
strategy.
For the construction, there are two kinds of hybrid energy
storage structures in common use. One is that the battery connected to the dc bus directly and the supercapacitor linked to
the dc bus through a bidirectional dc–dc converter [1]; the other
is that the battery and supercapacitor both linked to the dc bus
by dc–dc converters [2]. In the former structure, the output
power of the battery is uncontrollable and the voltage of the
battery should be in the same level as the dc bus. The latter
structure controls the input or output power of the storage elements well by a rational control strategy, but the efficiency
and dynamic response are limited. Multiport bidirectional converters for a hybrid energy storage system have been reported
in [3]–[6] with the advantages of low cost, high efficiency, compact structure, and unified power management. Many three-port
bidirectional dc–dc converters have been proposed [7]–[10] and
most of them are based on the dual active bridge (DAB) dc–dc
converters. Suppose a full-bridge structure is called a converter
unit. In [11], three converter units are connected to a threewinding transformer by magnetic coupling, in order to compose
a three-port bidirectional converter. The magnitude and direction of the transferred power can be controlled by the difference
in phase between pulse voltages formed by two converter units
on both sides of the transformer. Besides the voltage-mode converter unit [11]–[17], the current-mode one [15]–[17] can also
be used. In the current-mode structure, the output current on the
bridge side is continuous and its ripples are small, which suits
for battery and supercapacitor applications. In [18], two input
ports are coupled by the dc bus in the three-port bidirectional
converter, and this topology has the advantages of fewer devices
and simpler control strategy. However, exactly voltage matching
with wide input voltages cannot be achieved in these converters, which causes current stress increased and soft-switching
unsatisfied. Although the duty cycle control can be introduced
to extend the zero voltage switching (ZVS) operating range, the
0885-8993 © 2013 IEEE
1596
Fig. 1.
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
Topology of the proposed multiport converter.
voltage matching is still unrealized when two of the three ports
are connected to energy storage elements.
For the energy management strategy, the simple-rule method
[23]–[25], the filter method [26]–[28], the fuzzy-control method
[29], [30], the optimization method [31], and some other methods are commonly used. The simple-rule method is easy to
realize: according to the working mode analysis, a simple control principle is set to realize the control of mode switching.
However, this method is not flexible enough and lacks robustness. In the filter method, the high-frequency pulsation power
is separated by a filter and absorbed by the supercapacitor and
the remaining power is absorbed by the battery. But proper distribution of the power in the long time range cannot be realized.
The control method based on the fuzzy logic can distribute the
power between the battery and supercapacitor in real time. It has
advantages of strong reasoning ability and good robustness, but
high-frequency pulsation current still goes through the battery,
which is bad for the life of battery. The algorithm based on the
optimization method is complex and has mass workload, which
makes it difficult to be realized.
In this paper, a novel multiport phase-shift bidirectional dc–dc
converter is proposed and its operation principle and characteristics are analyzed. To highlight the performance of the converter
and improve its converting efficiency, a proper design method is
proposed in terms of the soft-switching condition and other factors. Meanwhile, the corresponding control strategy is proposed
to realize energy transfer and free mode transition. Detailed
analysis on the control strategy and a corresponding control
scheme are given. Then, the operation principle and hybrid energy storage control strategy of the aforementioned converter
are verified by the experiments on a prototype converter. The
results indicate that ZVS can be achieved for all switches in any
direction of power flow and in the whole load range. It also indicates that the given control strategy and method can cover all
operating conditions of the whole hybrid energy storage system.
II. PRINCIPLE OF THE MULTIPORT PHASE-SHIFT
BIDIRECTIONAL DC–DC CONVERTER
Fig. 1 shows the topology of the proposed converter. The converter units are distributed symmetrically on both sides of the
transformer. The left side (primary side) is defined as the battery
side. On this side, inductors L1 , L2 and switches S1u , S2u , S1d ,
S2d (including body diodes and parasitic capacitors) compose
a two-channel interleaving Buck/Boost construct, in which the
low-voltage port is connected to the battery. UBA is defined as
the battery voltage. The right side (secondary side) is defined as
the dc-bus side. On this side, inductors L3 , L4 and switches S3u ,
S4u , S3d , S4d (including body diodes and parasitic capacitors)
compose a two-channel interleaving Buck/Boost construct, in
which the high-voltage port is connected to the dc bus and the
low-voltage port is connected to the supercapacitor. UDC is the
dc-bus voltage and USC is the supercapacitor voltage. The converter also has an internal high-voltage port on the battery side,
which is connected to a filter capacitor Cc without any power
input or output. Performance of the converter can be improved
by controlling the voltage of this port. So, it is also supposed
that the converter is named as a “3 + 1”−port bidirectional
converter.
Sketch of the drive signals for power switches and the operation waveforms are shown in Fig. 2. The phase-shift angle of
the converter is defined as φ (−π ≤ φ ≤ π). D1 is the duty cycle
of the switches S1u (or S2u ) on the battery side. S1u and S2u
are conducted ON by interleaving the drive signals with 180◦ .
Meanwhile, the drive signals of S1u and S1d are complementary, and so it is with S2u and S2d . D2 denotes the duty cycle of
the switches S3u (or S4u ) on the dc-bus side. S3u and S4u are
also conducted ON by interleaving the drive signals with 180◦ .
Meanwhile, the drive signals of S3u and S3d are complementary,
and so it is with S4u and S4d .
As indicated in Fig. 2(a) and (b), the primary voltage of the
transformer is exactly the voltage across the filter capacitor Cc ,
while the secondary voltage of the transformer is the voltage of
the dc bus. Therefore, no matter how the voltages of battery, supercapacitor, and the dc bus change, the primary and secondary
voltages of the transformer are matching well as long as the
voltage across the capacitor Cc is kept following the voltage of
the dc bus.
The voltage across the capacitor Cc can be controlled by
adjusting the duty cycle D1 of the battery side. The duty cycle
D2 of the dc-bus side depends on the voltage of the dc bus and
supercapacitor. The power flow between the battery side and
the dc-bus side is controlled by the phase-shift angle φ shown
in Fig. 2(c). Therefore, the duty cycle and phase-shift control
are decoupled, which greatly reduces the control complexity of
power flow.
A. Power Flow Calculation
The transferred power changes, while the phase-shift angle
φ and the duty cycle D1 , D2 change. Due to the different duty
cycle regions, analysis of the power flow relations is divided
into 16 kinds of situations shown in Fig. 3. The power flow
is symmetrical about the duty cycle 0.5. Therefore, the power
flow relations of regions “1,” “3,” “13,” and “15” are the same,
those of regions “2,” “4,” “14,” and “16” are the same, those
of regions “6,” “8,” “10,” and “12” are the same, and those of
the rest regions are the same. With the same analysis for regions
“1,” “2,” “9,” and “10,” the power flow relations in the duty cycle
range of [0, 1] can also be deduced.
Take region “1” as an example, with the phase-shift angle φ
changed in the range of [−0.5π, 0.5π], the transferred power
from the battery side to the dc-bus side is analyzed. Voltage and
current waveforms for the transformer of different phase-shift
DING et al.: NOVEL SOFT-SWITCHING MULTIPORT BIDIRECTIONAL DC–DC CONVERTER FOR HYBRID ENERGY STORAGE SYSTEM
1597
Fig. 2. Sketch of the drive signals for power switches and the operation waveforms. (a) Primary side. (b) Secondary side. (c) Both sides and the leakage inductor
current.
Fig. 3.
Symmetrical power distribution in different duty ratio regions.
angles in region “1” are illustrated in Fig. 4. So, the power flow
in region “1” can be derived in (1). If no control is given to the
capacitor Cc and the duty cycle D1 of the battery side is fixed
to 0.5, the power flow is given in the (1) and (2), at the bottom
of this page.
Curves of the transmitted power versus the phase-shift angle
in several special cases are shown in Fig. 5. As an example
PBA
1
PBA
to be analyzed, the battery voltage is considered to be about
50 V. Each curve is under a different situation of the battery
voltage and the duty cycle D2 of the dc-bus side. The solid
lines represent the situation when the primary and secondary
voltages of the transformer match well, while the dashed lines
represent the situation when those do not. When the voltages
of the transformer just match, the curves in two control modes
overlap.
With the same analysis for the other 15 regions, the power
relations in the duty cycle range of [0, 1] are also able to be
deduced. The converter transfers the maximum power with φ =
0.5π, which is shown in Fig. 6. The maximum power reaches
the maximum in the situation of D1 = D2 = 0.5. If the duty
cycle increases or decreases, the maximum power decreases.
According to (1), when the switching frequency is fixed, the
power is related to the leakage inductor besides the duty cycles
and phase-shift angle. Therefore, the leakage inductor should
be designed to ensure the maximum power transferred in the
whole operating range to meet the requirement of the rated
power.
⎧
UC2 c
⎪
⎪
[π 2 − 2π 2 (D1 + D2 − D12 − D22 ) + 2πφ + 2φ2 ],
φ ⊂ [−0.5π, πD1 + πD2 − π]
⎪
⎪
2πωL
⎪
r
⎪
⎪
⎪
⎪
2
⎪
⎪
⎪ UC c [4πD φ + (πD − πD − φ)2 ],
⎪
φ ⊂ [πD1 + πD2 − π, πD1 − πD2 ]
2
2
1
⎪
⎪
2πωLr
⎪
⎪
⎪
⎪
⎨
UC2 c
=
(4πD1 φ),
φ ⊂ [πD1 − πD2 , πD2 − πD1 ]
⎪
2πωLr
⎪
⎪
⎪
⎪
⎪ U2
⎪
⎪
Cc
⎪
[4πD1 φ − (πD1 − πD2 + φ)2 ],
φ ⊂ [πD2 − πD1 , π − πD1 − πD2 ]
⎪
⎪
⎪
2πωL
r
⎪
⎪
⎪
⎪
⎪
⎪ UC2 c
⎪
⎩
[−π 2 + 2π 2 (D1 + D2 − D12 − D22 ) + 2πφ − 2φ2 ], φ ⊂ [π − πD1 − πD2 , 0.5π]
2πωLr
⎧n U U
1 C c DC
⎪
[(πD2 − 0.5π − φ)2 + 2πD2 φ],
φ ⊂ [−0.5π, πD2 − 0.5π]
⎪
⎪
n
πωL
⎪
2
r
⎪
⎪
⎪
⎨n U U
1 C c DC
=
(4πD2 φ),
φ ⊂ [πD2 − 0.5π, 0.5π − πD2 ]
⎪
n2 πωLr
⎪
⎪
⎪
⎪
⎪
⎪
⎩ n1 UC c UDC [−(0.5π − πD + φ)2 + 2πφ − 2πD φ], φ ⊂ [0.5π − πD , 0.5π].
2
2
2
n2 πωLr
(1)
(2)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
Fig. 4. Voltage and current waveforms for the transformer with different phase-shift angles in region “1.” (a) φ ⊂ [−0.5π, πD 1 + πD 2 − π]. (b) φ ⊂
[πD 1 + πD 2 − π, πD 1 − πD 2 ]. (c) φ ⊂ [πD 1 − πD 2 , πD 2 − πD 1 ]. (d) φ ⊂ [πD 2 − πD 1 , π − πD 1 − πD 2 ]. (e) φ ⊂ [π − πD 1 − πD 2 , 0.5π].
Fig. 7.
RMS value of L r in two control modes.
B. RMS Current Through Leakage Inductor Lr
Fig. 5. Curves of the transmitted power versus the phase-shift angle in some
special cases (pu = (n 1 U D C /n 2 )2 /(2πωL r ), (n 1 /n 2 )U D C = 100 V, and
U B A = 40 V∼50 V).
Fig. 6.
Maximum power transferred from the battery side.
When the rated power is transferred from the battery side, the
RMS current through leakage inductor Lr is
2π
1
IL r rm s =
iL r (θ)2 dθ.
(3)
2π 0
The RMS current through Lr in two control modes in the
situation of (n1 /n2 )UDC = 100 V is shown in Fig. 7. Under no
voltage matching control, the duty cycle of D1 is fixed to 0.5.
In most regions of the operating range (regions “1” and “2”),
the RMS current through the leakage inductor Lr is greatly decreased after the capacitor Cc port is under the voltage matching
control. The leakage inductor current under the voltage matching control is a little greater than that under no control in regions
“3” and “4.” It is because when the capacitor Cc port is under
no control, the capacitor Cc voltage UC c = 2UBA , which influences the magnitude of the transferred power and the leakage
inductor current.
DING et al.: NOVEL SOFT-SWITCHING MULTIPORT BIDIRECTIONAL DC–DC CONVERTER FOR HYBRID ENERGY STORAGE SYSTEM
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TABLE I
TRANSFORMER CURRENT FOR THE BATTERY SIDE BEFORE POWER SWITCHES
ON IN REGION “1”
Fig. 8. Curves of K Q P versus the phase-shift angle ((n 1 /n 2 )U D C =
100 V, U B A = 40 V/45/50 V).
C. Active and Reactive Power Calculations
The leakage inductor of the transformer exchanges reactive
power as well as active power.
In a switching period, the RMS voltage across the leakage
inductor Lr is
2π
1
(uab (θ) − ucd (θ))2 dθ.
(4)
UL r rm s =
2π 0
The reactive power of the leakage inductor is
QL r = UL r
rm s IL r rm s .
(5)
The ratio of the reactive and active powers is
KQP =
QL r
.
PBA
(6)
Curves of KQP versus the phase-shift angle φ with
(n1 /n2 )UDC = 100 V in the situation of two control modes
are shown in Fig. 8. They are drawn under the condition of
(n1 /n2 )UDC = 100 V. When the voltages match, the battery
voltage UBA = 50 V. When the voltage of the capacitor Cc
port is not controlled, as the battery voltage deviates from 50 V,
the voltage mismatching is increasingly serious and the ratio of
the reactive power and the output power is increasingly greater.
By contrast, the reactive power decreases apparently after the
capacitor Cc port is under the voltage matching control. Meanwhile, in order to reduce the reactive component and improve
the efficiency, the phase-shift angle should be decreased as much
as possible. Therefore, a leakage inductor is designed small to
satisfy the fixed demand on transferred power.
III. ANALYSIS ON THE SOFT-SWITCHING CONDITION AND
DESIGN OF THE CONVERTER
A. Analysis on ZVS Operating Ranges on the Battery Side
When the switch is OFF, sudden changes on the switch voltage cannot be made due to the parallel junction capacitor or
parallel capacitor, so ZVS can be seen achieved. If the antipar-
Fig. 9.
Curves of iL r 1 versus φ in region “1.”
allel diode is turned ON before the switch, this also makes it
possible to achieve ZVS [8], [21].
The inductor current iL 1 , iL 2 on the battery side, the inductor
current iL 3 , iL 4 on the dc-bus side, and the transformer current
iL r 1 , iL r 2 are shown in Fig. 1. The positive reference direction
is defined as shown in Fig. 1.
For the battery side, the soft-switching conditions are
⎧
iL r 1 ≤ iL 1 ,
S1u
⎪
⎪
⎪
⎨
iL r 1 ≥ iL 1 ,
S1d
(7)
⎪
−iL r 1 ≤ iL 2 , S2u
⎪
⎪
⎩
−iL r 1 ≥ iL 2 , S2d
where iL 1 = iL 2 = 2DP1BUAC c , and the current ripple is ignored.
The leakage inductor current iL r 1 at the turn-on time of each
switch also changes, as the phase-shift angle φ and the duty
cycles D1 , D2 change. Take region “1” ((D2 < D1 < 0.5) ∩
(D2 + D1 > 0.5)) in Fig. 3 as an example; the soft-switching
range of each switch on the battery side is analyzed when the
phase-shift angle varies in the range of [−0.5π, 0.5π]. Table I
shows the expressions of the transformer current for the battery
side before power switches ON in region “1.”
Curves of iL r 1 versus φ in [−0.5π, 0.5π] in the situation of
D1 = 0.4 and D2 = 0.45, shown in Fig. 9, are drawn according
to Table I. As indicated in Fig. 9, the switches on the battery
side can satisfy (7). Substituting the expression in Table I into
(7), ZVS conditions of S1u in region “1” are, as shown (8), at
the bottom of the next page.
All the five equations are always tenable in each phase-shift
angle regions, so S1u can achieve ZVS in region “1.” The transformer current when S1d and S1u are ON is centered on the
origin, so ZVS of S1d can also be achieved in region “1.” The
ZVS operating range of S2u is the same as that of S1u , and so
as the ranges of S2d and S1d .
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
TABLE II
TRANSFORMER CURRENT FOR THE DC-BUS SIDE BEFORE POWER SWITCHES
ON IN REGION “9”
Fig. 10. Soft-switching operating range for the battery side. (a) Soft-switching
range for S1 u , S2 u . (b) Soft-switching range for S1 d , S2 d .
With the same analysis for the other 15 regions, the softswitching operating range for the battery side can be obtained,
which is shown in Fig. 10. When φ < 0 and D2 approaches
0.5, the ZVS operating range of S1u and S2u becomes narrow.
Meanwhile, when φ > 0 and D2 approaches 0.5, the ZVS range
of S1d and S2d becomes narrow. It can be seen that all switches
of the battery side are able to achieve ZVS at any output power
when D1 ≤ 0.5 which can be satisfied by the right transformer
turns ratio.
B. Analysis on the ZVS Ranges for the DC-Bus Side
The dc bus exchanges energy with the battery side as well as
the supercapacitor.
1) DC Bus and the Supercapacitor: The dc bus and the supercapacitor are connected together by a two-channel interleaving Buck/Boost converter. The upper and down switches in each
channel are conducted ON by complementary drive signals. The
inductors are designed properly to keep the inductor current
working with positive and negative states during a switching
period so that ZVS can be achieved regardless of the direction
of power flow [22].
The expressions of one-channel inductor current and its fluctuation are as follows.
The average inductor current is
IL = ±
PSC
.
nUSC
(9)
The inductor current fluctuation is
ΔI =
(UDC − USC )USC
.
LUDC f
(10)
The maximum inductor current is
PSC
1
(UDC − USC )USC
im ax = IL + ΔI = ±
.
+
2
nUSC
2LUDC f
(11)
The minimum inductor current is
PSC
1
(UDC − USC )USC
−
im in = IL − ΔI = ±
2
nUSC
2LUDC f
(12)
where IL = IL 3 = IL 4 , L = L3 = L4 , and n equals the number
of interleaving phases. If the converter works in the Buck mode,
IL is positive; and if in the Boost mode, IL is negative.
In the Buck mode, im in < 0 must be ensured to keep the
upper switches S3u and S4u achieving ZVS. So, as indicated in
(12), as long as im in < 0, when the output power PSC reaches
its maximum and the dc-bus voltage UDC reaches its minimum,
im in < 0 in the whole load range can be ensured.
In the Boost mode, im ax > 0 must be ensured to keep the
down switches S3d and S4d achieving ZVS. So, as indicated in
(11), as long as im ax > 0, when the output power PSC reaches
its maximum and the dc-bus voltage UDC reaches its minimum,
im ax > 0 in the whole load range can be ensured.
2) DC-Bus Side and the Battery Side: When the dc-bus side
is transmitting energy with the battery side, analysis on ZVS
operating ranges for the dc-bus side is similar to those for the
battery side.
ZVS conditions for the switches S3u , S3d , S4u , and S4d on
the dc-bus side are
⎧
iL r 2 ≥ im in ,
S3u
⎪
⎪
⎪
⎨ iL r 2 ≤ im ax ,
S3d
(13)
⎪
−i
≥
i
,
S
⎪
L
r
2
m
in
4u
⎪
⎩
−iL r 2 ≤ im ax , S4d
where the expressions of im ax and im in are in (11) and (12). As
indicated in the analysis above, im in < 0, im ax > 0.
Take region “9”(region (D2 < D1 < 0.5) ∩ (D2 + D1 >
0.5)) in Fig. 3 as an example, the soft-switching operating range
for the dc-bus side when φ varies in [−0.5π, 0.5π] is analyzed.
Table II shows the transformer current for the dc-bus side before
power switches ON in region “9,” where N represents n1 /n2 .
Fig. 11 shows the curves of iL r 2 versus φ when D1 = 0.45 and
D2 = 0.4, which indicates that the switches on the dc-bus side
cannot satisfy (13) in the whole power range. For S3u and S4u ,
⎧ 2
2φ + (2π − 4πD1 )φ + π 2 − 2π 2 (D1 + D2 − D12 − D22 ) ≥ 0,
⎪
⎪
⎪
⎪
2
⎪
⎪
⎨ (φ + πD2 − πD1 ) ≥ 0,
0 ≥ 0,
⎪
⎪
⎪
⎪
φ2 − (2πD1 + 2πD2 )φ − 3π 2 D12 + π 2 D22 + 2π 2 D1 D2 ≤ 0,
⎪
⎪
⎩ 2
2φ − (2π + 4πD1 )φ + π 2 + 2π 2 (D1 − D2 + D22 ) − 6π 2 D12 ≤ 0,
φ ⊂ [−0.5π, πD1 + πD2 − π]
φ ⊂ [πD1 + πD2 − π, πD1 − πD2 ]
φ ⊂ [πD1 − πD2 , πD2 − πD1 ]
φ ⊂ [πD2 − πD1 , π − πD1 − πD2 ]
φ ⊂ [π − πD1 − πD2 , 0.5π].
(8)
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be achieved for all switches on the dc-bus side in the whole load
range.
C. Design Procedures of the Converter
Fig. 11.
Curves of iL r 2 versus φ in region “9.”
Fig. 12. Soft-switching operating range for the dc-bus side (a) Soft-switching
range for S3 u , S4 u . (b) Soft-switching range for S3 d , S4 d .
ZVS can be achieved completely when φ > 0; but when φ < 0,
iL r 2 has the minimum of (n1 /n2 )UC c (πD2 − πD1 )/(ωLr ), so
ZVS can be achieved as long as im in is less than this value. For
S3d and S4d , ZVS can be achieved completely when φ < 0; but
when φ > 0, ZVS can be achieved only when im ax > (n1 /n2 )
UC c (πD1 − πD2 )/(ωLr ) is ensured.
ZVS operating ranges for other regions on the dc-bus side
can be analyzed similarly, and the results are also similar.
Fig. 12 shows the soft-switching operating range for dc-bus
side. As D1 approaches 0.5, ZVS operating range becomes
narrow. According to (13) and Fig. 12, if the inductors L3 and
L4 are properly chosen to make im in less than the minimum
transformer current iL r 2(m in) before S3u and S4u are conducted
ON and im ax greater than the maximum transformer current
iL r 2(m ax) before S3d and S4d are conducted ON, then ZVS can
im in
im ax
1) Design Principles for the Parameters of the Power Circuit:
a) Turns ratio of the transformer: ZVS can be achieved for
all switches on the battery side in the whole load range when
D1 <= 0.5. Therefore, turns ratio should be chosen under the
condition that D1 <= 0.5 should be ensured when the battery
voltage is maximum and the dc-bus voltage is minimum.
b) Leakage inductance of the transformer: When φ = 0.5π,
the power transmitted between the battery side and the dc-bus
side reaches the maximum. When the leakage inductance is
chosen, it should be ensured that the maximum power transmitted at every operating point in the operating range meets the
requirement of the rated power.
c) Inductors on the dc-bus side: To ensure ZVS can be
achieved for all switches when three ports work together, im in
needs to satisfy (14), at the bottom of this page, for the upper
switches S3u and S4u and im ax needs to satisfy (15), at the
bottom of this page, for the down switches S3d and S4d .
d) Inductors on the battery side: When designing the inductor
on the battery side, high charging/discharging current ripple
should be prevented so as not to reduce the battery’s lifespan.
So the fluctuation of the inductor current is usually set to be
10–20% of the maximum inductor current.
2) Designing Example: Take a hybrid energy storage prototype applied to a 380-V dc microgrid experiment system for
example, and it is designed in the foregoing method according to the design requirements. Design specifications and key
parameters of the converter are as follows.
DC-bus voltage UDC = 350–420 V (380–420 V when charging, 350–380 V when discharging); battery voltage UBA = 40–
56.4 V, rated battery power PBA = 1 kW; supercapacitor voltage
USC = 150–300 V; rated supercapacitor power PSC = 10 kW;
and switching frequency f = 20 kHz.
a) Turns ratio of the transformer and the leakage inductance:
Select float charge voltage of the battery, 56.4 V, as the voltage
point when D1 = 0.5. To ensure that D1 ≤ 0.5 is satisfied when
the dc bus reaches its minimum voltage, the turns ratio is chosen
as n1 :n2 = 1:3.1 so that the operating ranges for D1 is 0.3 ≤
D1 ≤ 0.5.
⎧
(n1 /n2 )UC c (πD2 − πD1 )/(ωLr ),
⎪
⎪
⎪
⎨ (n /n )U (π(1 − D ) − πD )/(ωL ),
1
2
Cc
2
1
r
< min
⎪
(n1 /n2 )UC c (πD2 − π(1 − D1 ))/(ωLr ),
⎪
⎪
⎩
(n1 /n2 )UC c (πD1 − πD2 )/(ωLr ),
⎧
(n /n )U (πD1 − πD2 )/(ωLr ),
⎪
⎪ 1 2 Cc
⎪
⎨ (n1 /n2 )UC c (πD1 − π(1 − D2 ))/(ωLr ),
> max
⎪
(n1 /n2 )UC c (π(1 − D1 ) − πD2 )/(ωLr ),
⎪
⎪
⎩
(n1 /n2 )UC c (πD2 − πD1 )/(ωLr ),
D1 ≤ 0.5, D2 ≤ 0.5
D1 ≤ 0.5, D2 > 0.5
D1 > 0.5, D2 ≤ 0.5
(14)
D1 > 0.5, D2 > 0.5
D1 ≤ 0.5, D2 ≤ 0.5
D1 ≤ 0.5, D2 > 0.5
D1 > 0.5, D2 ≤ 0.5
D1 > 0.5, D2 > 0.5.
(15)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
Fig. 13. Curves of im a x and im in versus U S C . (a) Curves of im in versus
U S C in the Buck mode (U D C = 380 V, P S C = 10 kW, L = 40–60 μH).
(b) Curves of im a x versus U S C in the Boost mode (U D C = 350 V, P S C =
10 kW, and L = 40–60 μH).
The duty cycle D2 of the dc-bus side is determined by the
voltage ranges of the dc bus and the supercapacitor to work in
the range of 0.35 ≤ D2 ≤ 0.86.
According to Section II, the maximum power transferred by
the converter is
Pm ax =
u2C c
· 4π 2 D1 (1 − D2 )
2πωLr
(16)
where UC c = (n1 /n2 )UDC .
To ensure that the maximum power transmitted can meet the
requirement of the rated power when the duty cycles D1 and
D2 change in their own ranges, D1 is set to 0.3 and D2 is set to
0.86. According to (16) and the requirement of the rated power,
the leakage inductance Lr is designed to be 25 μH.
b) Inductors on the dc-bus side: When UDC = 350 V, USC =
300 V, and UBA = 56.4 V, (14) reaches its minimum of −13 A,
and (15) reaches its maximum of 13 A, which means when the
inductors on the dc-bus side are being designed, im in <−13 A,
im ax >13 A should be satisfied.
Fig. 13(a) shows the curves of im in versus USC in the Buck
mode, while Fig. 13(b) shows curves of im ax versus USC in
the Boost mode. With the consideration of the dead time, the
absorption capacitor of the switch, loss of the real circuit, and
Fig. 13, L3 , L4 are designed to be L3 = L4 = 45 μH.
c) Inductors on the battery side: The battery side is a twochannel interleaving Buck/Boost converter, and the total output
current ripple of two channels is shown as follows:
⎧
UC c − 2UBA UBA
⎪
Δio =
·
,
⎪
⎪
f ·L
UC c
⎨
⎪
⎪
⎪
⎩ Δio = 2UC c − 2UBA · UBA − 1 ,
f ·L
UC c
2
0 < D1 <
1
2
1
< D1 < 1.
2
(17)
When UC c = 135.5 V and UBA = 40 V, the output current
ripple reaches its maximum. Fluctuation in the inductor current
is set to be 20% of the maximum inductor current, and the
maximum battery current is set to be IBA(m ax) = 25 A, so
ΔIm ax = 5 A and L1 = L2 = 164 μH.
Fig. 14.
Control logic diagram of the converter in energy storage state.
IV. CONTROL STRATEGY OF THE PROPOSED HYBRID
ENERGY STORAGE SYSTEM
Battery has high energy density and is usually used as the
main energy storage element in the whole microgrid system
to reduce the steady fluctuation in the microgrid. Supercapacitor has the advantages of short charging/discharging time, long
circle life, and easy maintenance, and is used to absorb and
provide pulsation power. Assume that the battery has a large
capacity and the status of fully charged and discharged is not
taken into consideration; only the current status of the dc bus
and supercapacitor should be considered. In terms of the capacity of the supercapacitor, the voltage should be maintained
middle USC m id under normal operating conditions, in order
to respond to the next fluctuation of the dc-bus power in time.
Assume the maximum supercapacitor voltage is USC m ax , and
the minimum is USC m in . The supercapacitor voltage should
not be greater than the maximum or less than the minimum in
the operation. Suppose PDC represents the dc-bus power, and
PBA m ax represents the maximum battery charging/discharging
power.
According to the operating status of the battery, the whole
control strategy is divided into two parts: energy storage and
energy release.
A. States of Energy Storage
When the dc-bus power PDC > 0, which means the distributed
generation power is greater than that the load consumed, the redundant energy needs to be stored in the energy storage system;
the converter now works in the energy storage state. According to the current status of the supercapacitor, the control logic
diagram of the converter is shown in Fig. 14.
Energy storage state 1 USC < USC m id : The supercapacitor
voltage is less than the middle voltage. No matter how much
the dc-bus power is, the supercapacitor is charged first and the
battery is not charged.
Energy storage state 2 USC = USC m id : If the dc-bus power
PDC is less than the maximum battery charging power PBA m ax ,
then the battery is charged; the charging current of the battery is
determined by the bus energy; the supercapacitor is controlled by
limiting its voltage to keep the middle voltage USC m id constant.
DING et al.: NOVEL SOFT-SWITCHING MULTIPORT BIDIRECTIONAL DC–DC CONVERTER FOR HYBRID ENERGY STORAGE SYSTEM
Fig. 15.
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Control logic diagram of the converter in energy release state.
If PDC > PBA m ax , the system is turned to energy storage state
3.1.
Energy storage state 3.1 USC m id < USC < USC m ax : If
PDC > PBA m ax , the battery is charged by the constant maximum current and the redundant power is absorbed by the supercapacitor until the supercapacitor is fully charged to its highest
voltage and the state is turned to energy storage state 4. If
PDC < PBA m ax , the system is turned to energy storage state
3.2.
Energy storage state 3.2 USC m id < USC < USC m ax : If
PDC > PBA m ax , the system is turned to energy storage state
3.1. If PDC < PBA m ax , the battery is still charged by the constant maximum current; the supercapacitor is discharged to
maintain the steady of the dc bus from the voltage drop; at the
same time, the charge state of the supercapacitor is adjusted until
the supercapacitor voltage reaches the middle voltage USC m id ,
and the state is turned to energy storage state 2.
Energy storage state 4 USC = USC m ax : the supercapacitor
reaches its maximum voltage. If PDC > PBA m ax , the supercapacitor voltage must be restricted and it is not charged, then
the dc-bus voltage rises and triggers the overvoltage warning. If
PDC < PBA m ax , the system is turned to energy storage state
3.2.
Under normal energy storage conditions, the whole energy
storage port unit works in energy storage state 2, the supercapacitor maintaining the middle voltage, and responding to the
dc-bus fluctuation in time.
B. Energy Release States
When the dc-bus power PDC < 0, which means the distributed
generation power is less than the load consumed, the energy
storage system must provide energy to maintain the steady of
the dc-bus voltage; the converter now works in the energy release
state. According to the current status of the supercapacitor, the
control logic diagram of the converter is shown in Fig. 15.
Energy release state 1 USC > USC m id : The supercapacitor
voltage is greater than the middle voltage. No matter how much
the dc-bus power is, the supercapacitor is discharged first and
the battery does not act until the supercapacitor USC = USC m id ,
and the system is turned to energy release state 2.
Energy release state 2 USC = USC m id : If the power of the dc
bus PDC is less than the maximum battery discharging power
Fig. 16.
Control block diagram of the hybrid energy storage system.
PBA m ax , the battery is discharged as power needed and the
supercapacitor is controlled to keep the middle voltage USC m id
constant. If PDC > PBA m ax , the system is turned to energy
release state 1.
Energy release state 3.1 USC m in < USC < USC m id : If
PDC > PBA m ax , the battery is discharged by its constant maximum current and the supercapacitor is discharged to provide
other power until the supercapacitor reaches its allowed lowest
voltage USC m in , and then the system is turned to energy release state 4. If PDC < PBA m ax , the system is turned to energy
release state 3.2.
Energy release state 3.2 USC m in < USC < USC m id : If
PDC > PBA m ax , the system is turned to energy release state
3.1. If PDC < PBA m ax , the battery is still charged by the constant maximum current and the redundant power charges the
supercapacitor until it reaches the middle voltage USC m id and
then the system is turned to energy release state 2.
Energy release state 4 USC = USC m in : the supercapacitor voltage is its allowed lowest operation voltage. If PDC >
PBA m ax , the supercapacitor needs to be limited to the low
voltage and is not discharged, so the dc-bus voltage drops and
triggers the low-voltage warning. If PDC < PBA m ax , the system is turned to energy release state 3.2.
Under normal energy release conditions, the whole energy
storage unit works in energy release state 2, the supercapacitor
maintaining the middle voltage, and responding to the dc-bus
fluctuation in time.
C. Design of the Control Scheme
Fig. 16 shows the control block diagram of the hybrid energy storage system based on the analysis on the hybrid energy
storage control strategy. It mainly contains the dc-bus voltage
loop, the supercapacitor voltage loop, the supercapacitor current
loop, the battery current loop, the voltage matching loop, the
logic synthesis block, and battery SOC estimate block, where
∗
∗
∗
, IBA
, and ISC
are the references of the dc-bus voltage,
UDC
the battery, and the supercapacitor charge or discharge currents
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
Fig. 17. Simulation waveforms with voltage matching control. (a) U B A = 40 V, U S C = 180 V (charging simultaneously). (b) U B A = 56 V, U S C = 300 V
(charging simultaneously). (c) U B A = 50 V, U S C = 240 V (discharging simultaneously).
∗
∗
∗
respectively, USC
m ax , USC m id , and USC m in are the references
of the highest, middle, and lowest supercapacitor voltage, re∗
∗
∗
spectively, and ISC
m ax , ISC m id , and ISC m in are the outputs
of the supercapacitor voltage loop, respectively. The controller
needs to sample six variables of the dc-bus voltage uDC , the
supercapacitor voltage uSC , the battery voltage uBA , the filter
capacitor voltage uC c , the supercapacitor current iSC , and the
battery current iBA . cSOC is a logical control signal determined
by the battery SOC estimate block. The algorithm of the battery
SOC estimate block is not taken into consideration in this paper.
The output value of the PI compensator in the dc-bus voltage loop after amplitude limitation is defined as I ∗ , which
∗
∗
∗
with ISC
m ax , ISC m id , ISC m in , and cSOC together turns into
∗
and the supercapacitor chargthe battery current index IBA
∗
after logic synthesis block.
ing/discharging current index ISC
Detailed logic synthesis is according to the energy management
strategy in Sections IV-A and IV-B. When the supercapacitor voltage approaches its highest, middle, or lowest voltage,
the logic synthesis block can automatically adjust the super∗
, slowly decreasing its charging/
capacitor current index ISC
discharging current to limit the change of the supercapacitor
voltage. The control of the battery current iBA is realized by
adjusting the phase-shift angle. The output of the PI compensator goes through a phase-shifted controller and an oscillator
and gets a triangular carrier with a certain phase-shift angle. The
greater the power of the battery side, the bigger the phase-shift
angle. The output of the supercapacitor current loop intersects
with the triangular carrier and gets the drive signal on the dc-bus
side. Meanwhile, the voltage matching loop uses a PI compensator to control the filter capacitor voltage uC c to follow the
dc-bus voltage (n1 /n2 )uDC . When (n1 /n2 )uDC > uC c , the ontime of S1d and S2d under closed-loop control increases and
uC c rises so that real-time dynamic voltage matching can be
realized. When (n1 /n2 )uDC < uC c , it is in a similar way.
V. SIMULATION AND EXPERIMENTAL RESEARCH
A. Simulation Research
1) Comparing the Working With Voltage Matching Control
or Not: The simulation parameters are as follows: dc-bus volt-
age UDC = 400 V; Battery voltage UBA = 40–56.4 V, rated
power PBA = 1 kW; supercapacitor voltage USC = 150–300 V,
pulsation power PSC = 10 kW; turns ratio of the transformer
n1 :n2 = 1:3.1, the leakage inductance Lr = 25 μH; the inductors L1 = L2 = 164 μH, L3 = L4 = 45 μH; and switching
frequency f = 20 kHz.
Fig. 17 shows simulation waveforms with voltage matching
control. Fig. 17(a) and (b) shows simulation waveforms for
the dc bus charging the battery and the supercapacitor. Due to
charging the battery, uab lags behind (n1 /n2 )ucd with a certain
phase-shift angle. The battery voltage in Fig. 17(a) UBA = 40 V,
while it changes to 56 V and the duty cycle D1 changes correspondingly to ensure voltage matching of all time. At the same
time, the supercapacitor voltage changes from 180 to 300 V and
the duty cycle on the dc-bus side D2 adjusts itself automatically without affecting the voltage matching control. Fig. 17(c)
shows simulation waveforms for the battery and the supercapacitor discharging simultaneously. The battery is discharged and
transfers energy to the dc-bus side, so uab leads (n1 /n2 )ucd with
a certain phase-shift angle. As indicated previously, no matter
how voltages of the battery and the supercapacitor change and
whether they are charging or discharging, as long as the duty
cycle on the battery side D1 is controlled to keep the capacitor
Cc voltage following (n1 /n2 )UDC of all time, voltage matching
can always be maintained.
Fig. 18 shows simulation waveforms without voltage matching control while circuit parameters are the same.
Comparing Fig. 17 with Fig. 18, it can be found that the
voltage matching can effectively reduce the winding current
through the transformer (i.e., the power switch current) and
decrease the cycling reactive power, which is good for improving
the converter’s efficiency.
2) ZVS Working: The design example in Section III-C is
simulated to verify the correctness of the analysis on ZVS operating ranges and effectiveness of the method for designing
parameters.
Figs. 19 and 20 show simulation waveforms of soft-switching
operation for the battery side under different power conditions.
In Fig. 19, UDC = 380 V, USC = 190 V, UBA = 56.4 V, so D2 =
0.5 and now is the hardest condition for the upper switches
DING et al.: NOVEL SOFT-SWITCHING MULTIPORT BIDIRECTIONAL DC–DC CONVERTER FOR HYBRID ENERGY STORAGE SYSTEM
1605
Fig. 18. Simulation waveforms without voltage matching control. (a) U B A = 40 V, U S C = 180 V (Charging simultaneously). (b) U B A = 56 V, U S C = 300 V
(charging simultaneously). (c) U B A = 50 V, U S C = 240 V (discharging simultaneously).
Fig. 19.
Simulation waveforms of the soft-switching operation for the battery side (φ < 0). (a) Full load. (b) Half-load. (c) 20% load.
Fig. 20.
Simulation waveforms of the soft-switching operation for the battery side (φ > 0). (a) Full load. (b) Half-load. (c) 20% load.
on the battery side to achieve ZVS. As the figure indicates,
even at light load, the switches on the battery side still satisfy
(13). Therefore, ZVS for all switches on the battery side can be
achieved in the whole power range and directions.
Fig. 21 shows the simulation waveforms of soft-switching
operation for the dc-bus side. Fig. 21(a) shows the waveforms
for the battery discharging and the supercapacitor charging, with
USC = 300 V, UBA = 56.4 V, when it is the hardest condition for
the upper switches on the bus side to achieve ZVS. Fig. 21(b)
shows the waveforms for the battery and the supercapacitor
discharging, USC = 300 V, UBA = 56.4 V, when it is the hardest
condition for the down switches on the bus side to achieve ZVS.
As indicated in the figure, the filter inductor on the bus side
L3 = L4 = 45 μH can ensure ZVS for all switches when three
ports work together. Theoretically, it is the hardest moment for
ZVS to be achieved when UDC = 350 V, but in practical, the
bus voltage should be maintained steady at 380 V.
B. Experimental Research
The parameters of the experimental prototype are same as
that of the simulation converter.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
Fig. 23. Experimental waveforms of power flow from the dc bus to the battery
and supercapacitor. (Ch1: u G E (S 1 u ) , 20 V/div; Ch2: u c d , 400 V/div; Ch3: u a b ,
200 V/div; and Ch4: iL r , 10 A/div). (a) (U D C = 380 V, U S C = 200 V, and
U B A = 50 V). (b) (U D C = 380 V, U S C = 237 V, and U B A = 42 V).
Fig. 21. Simulation waveforms of the soft-switching operation for the dc-bus
side. (a) Battery discharging, supercapacitor charging. (b) Battery charging,
supercapacitor discharging.
Fig. 22. Experimental waveforms of power flow between the dc-bus side
and the battery side (Ch1: u c d , 400 V/div; Ch2: u a b , 200 V/div; Ch3: iB A ,
6.67 A/div; and Ch4: iL r , 10 A/div) (a) From the dc-bus side to the battery side
(U D C = 380 V, U B A = 50.3 V, and IB A = −20.3 A). (b) From the battery
side to the dc-bus side (U D C = 354 V, U B A = 50.6 V, and IB A = 20.6 A).
The control system of the converter is based on
TMS320F28335. The foregoing energy management strategy
is realized by programming.
1) Steady Operation: When the system operates in steady
state, energy can be transferred between the three ports of the
multiport bidirectional converter and the accessional fourth port
is used for voltage matching control so that the voltages of the
transformer still match well even if the voltages of the battery
and the supercapacitor fluctuate or change.
Energy can be transferred between the dc-bus side and the battery side through the phase-shift transformer. When the voltage
of the converter unit on the dc-bus side leads that on the battery
side, the battery is charged. When the voltage of the converter
unit on the dc-bus side lags that on the battery side, the battery
is discharged. The resistor is used to simulate the supercapacitor, and the duty cycle on the dc-bus side is fixed to D2 =
0.5. Fig. 22 shows the experimental waveforms of power flow
between the dc-bus side and the battery side. Fig. 22(a) shows
the experimental waveforms for the dc-bus side while charging
the battery, ucd leading uab ; Fig. 22(b) shows the experimental
waveforms for the dc-bus side while discharging the battery,
uab leading ucd . As indicated in the waveforms for the uab and
iL r , the voltage matching loop works and voltage matching is
realized perfectly by adjusting the duty cycle D1 of the battery
side.
Energy is transferred from the dc bus to the supercapacitor through the Buck/Boost converter. The supercapacitor is
Fig. 24. Experimental platform of the multiport bidirectional dc–dc converter.
(a) Experimental frame. (b) Experimental setup.
charged and the converter works in the Buck mode, meanwhile
the dc bus charges the battery through the phase-shift transformer. Fig. 23 shows the experimental waveforms of power
flow from the dc bus to the battery and supercapacitor and when
the voltages of the battery and the supercapacitor both change. In
Fig. 23(a), USC = 200 V and UBA = 50 V, while in Fig. 23(b),
USC = 237 V and UBA = 42 V. The voltages of the battery
and the supercapacitor both change. The waveforms for uab
and ucd are observed. The voltage matching loop adjusts D1
so that the filter capacitor voltage follows the dc-bus voltage to
ensure voltages always match; the dc-bus voltage outer loop and
supercapacitor current inner loop adjusts D2 correspondingly.
Besides, the battery current loop adjusts the phase-shift angle
φ to control the battery to be charged with constant 20 A. In
this way, the phase-shift angle φ, the duty cycles D1 and D2 are
decoupled.
2) Experiment Research on the Control Scheme: In this section’s experiments, to simulate the operation condition of the
dc bus and the energy storage elements and to take photos of
the dynamic experimental waveforms for the converter between
different modes, a dc source in series with a resistor and in parallel with another resistor is used to simulate the dc bus, and a
dc source in parallel to a resistor or a resistor only is used to
simulate the discharging or charging state of the supercapacitor
and battery, respectively, as shown in Fig. 24.
When the dc-bus voltage is greater than 380 V (uin > 380 V),
the energy storage port unit works in energy storage state, with
battery charged. The energy storage consists of five states: state
DING et al.: NOVEL SOFT-SWITCHING MULTIPORT BIDIRECTIONAL DC–DC CONVERTER FOR HYBRID ENERGY STORAGE SYSTEM
Fig. 25. Experimental waveforms of modes transition from energy storage
state 1 to 2.
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Fig. 27. Experimental waveforms of modes transition from energy storage
state 3.1 to 3.2. (a) From energy storage state 3.1 to 3.2. (b) From energy
storage state 3.2 to 3.1.
Fig. 26. Experimental waveforms of modes transition from energy storage
state 2 to 3.1.
1, 2, 3.1, 3.2 and 4. If the supercapacitor voltage and the dc-bus
voltage change, the converter can be switched to the corresponding operating state.
Fig. 25 shows the experimental waveforms of modes transition from energy storage state 1 to 2. Before t1 , the system works
in energy storage state 1, uSC < 225 V, and the dc-bus energy
is absorbed by the supercapacitor. At t1 , uin rises to simulate
dc-bus power increasing, and then the supercapacitor charging
current increases correspondingly, absorbing the bus energy to
keep the dc-bus voltage steady, while the supercapacitor voltage rises (the supercapacitor is simulated by a resistor). At t2 ,
the supercapacitor voltage rises to 225 V and the system turns
to energy storage state 2, the supercapacitor voltage steady at
the middle voltage, and the dc-bus energy absorbed by the battery, causing the battery charging current to increase. At t3 , the
dc-bus power is steady and the system works steady in energy
storage state 2.
Fig. 26 shows the experimental waveforms of modes transition from energy storage state 2 to 3.1. Before t1 , the system
works in energy storage state 2, uSC = 225 V. At t1 , the dc-bus
power rises. The battery absorbs the bus energy causing the battery voltage to rise (the battery is simulated by a resistor) and
the supercapacitor voltage is maintained at the middle voltage.
Then, the bus power continues to increase. At t2 , the battery
reaches the maximum charging power. At the same time, the
supercapacitor continues to absorb additional energy of the bus,
causing the supercapacitor voltage to rise (the supercapacitor is
simulated by a resistor, too), and the battery charges with the
maximum power. At t3 , the dc-bus power becomes constant and
the system works steadily in energy storage state 3.1.
Fig. 27 shows the experimental waveforms of modes transition between energy storage state 3.1 and 3.2.
Fig. 27(a) shows the experimental waveforms of modes transition from energy storage state 3.1 to 3.2. Before t1 , the dc-bus
energy charges the battery with its maximum power and addi-
Fig. 28. Experimental waveforms of modes transition from energy storage
state 3.1 to 4.
tional power is absorbed by the supercapacitor, through which
the current is positive. From t1 , the dc-bus power gradually decreases and the supercapacitor charging current decreases. At t2 ,
the supercapacitor current crosses zero and go negative. Then,
the supercapacitor turns to be discharged to maintain the steady
of the dc-bus voltage to prevent it from dropping and the system
enters energy storage 3.2. At t3 , the dc-bus power is steady and
the system works steady in energy storage state 3.2.
Fig. 27(b) shows the experimental waveforms of modes transition from energy storage state 3.2 to 3.1. Before t1 , the system
works in energy storage state 3.2 where the dc bus and the
supercapacitor together provide energy to charge the battery
with the maximum power. From t1 , the dc-bus power gradually
increases and the supercapacitor discharging current gradually
decreases. At t2 , the supercapacitor current goes across zero and
turns to charging from discharging. The increased bus energy
is absorbed by supercapacitor to keep the dc-bus voltage steady
and the system enters energy storage state 3.1. At t3 , the dc-bus
power stops increasing and the system works in energy storage
state 3.1.
Fig. 28 shows the experimental waveforms of modes transition from energy storage state 3.1 to 4. Before t1 , the system
works in energy storage state 3.1, where the battery is charged
with the maximum power and the additional power of the bus
is absorbed by the supercapacitor. From t1 , the dc-bus power
increases and the supercapacitor charging current increases correspondingly. At t2 , the supercapacitor voltage reaches the highest, 300 V. The system controls the supercapacitor by limiting
voltage and the supercapacitor stops being charged. With the
dc-bus power increasing further, the voltage rises further and
the bus voltage is out of control and will trigger the overvoltage
warning.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 4, APRIL 2014
The experiment process and working conditions of the energy
release state are similar.
VI. CONCLUSION
A novel multiport bidirectional dc–dc converter, which can
be named as a “3 + 1”−port bidirectional converter, is proposed. In it, one port is connected to the dc bus, two ports are
connected to battery and supercapacitor, respectively, and the
accessional fourth port is used for voltage matching control, by
which voltage matching can be kept perfectly even when two energy storage elements are connected simultaneously. Compared
to the topology without the fourth port or without controlling
the fourth port, the reactive power and circulating current are
greatly reduced in this topology. The converter units on both
sides of the transformer use interleaving control so that the output current ripple is small. In terms of the port extension, this
novel converter can be extended into a serial “n+(n − 2)” multiport bidirectional dc–dc converter by magnetic coupling. It suits
for hybrid applications of more energy storage elements to use
internal ports of the converter for voltage matching control.
The ZVS conditions and ranges for all switches of the converter are analyzed and some reference principles are proposed
for designing the parameters of the power circuit.
1) When the duty cycle of the battery side D1 ≤ 0.5, no
matter whether the battery is charging or discharging, no
matter how much the transmitted power is, ZVS for all
switches on the battery side can be achieved. Therefore,
turns ratio of the transformer should be properly selected
to make the duty cycle of the battery side satisfy D1 ≤
0.5.
2) The upper and down switches of each channel in the twochannel interleaving Buck/Boost converter on the dc-bus
side are conducted on complementarily. The inductors are
designed properly to keep the inductor current working
in a positive and negative state during a switching period,
which can decrease the inductance, improve the dynamic
response, and provide the ZVS conditions for switches.
3) When the battery and the supercapacitor work together, the
ZVS range on the dc bus is affected by the supercapacitor
charging/discharging current. ZVS for all switches can be
achieved by designing the filter inductor.
Based on the 380-V dc-bus microgrid experimental system,
the battery and supercapacitor hybrid energy storage control
strategy is researched. The novel multiport bidirectional converter and the hybrid energy storage control strategy can realize
transmission and proper distribution of the energy. It can always
keep the dc-bus voltage steady according to the current status
of the dc bus and the energy storage elements, which is verified
effectively by the experimental results.
The experiment mainly includes principle validation without
the operating modes of the fully charged or fully discharged
state of the battery. A complete test for the converter combining
practical application is not run yet. SOC of the battery will be
taken into consideration and a complete design and test for the
converter combining the 380-V dc microgrid application will
also be done in the future.
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Zhihui Ding was born in Jiangsu, China, in 1987. She
received the B.S. and M.S. degrees in electrical engineering from the Nanjing University of Aeronautics
and Astronautics, Nanjing, China, in 2009 and 2012,
respectively.
She is currently at Delta Electronics (Shanghai)
Company, Ltd., Shanghai, China.
Chen Yang was born in Shaanxi, China, in 1985. He
received the B.S. and M.S. degrees in electrical engineering from the Nanjing University of Aeronautics
and Astronautics, Nanjing, China, in 2008 and 2011,
respectively, where he is currently working toward
the Ph.D. degree in electrical engineering.
His research is mainly focused on high-voltage
dc–dc converters for photovoltaic applications and
dc microgrid.
1609
Zhao Zhang was born in Jiangsu, China, in 1993. He
is currently working toward the B.S. degree in electrical engineering at the Nanjing University of Aeronautics and Astronautics (NUAA), Nanjing, China.
He is currently with the Graduate School of Electrical Engineering, NUAA.
Cheng Wang was born in Jiangsu, China, in 1988. He
received the B.S. and M.S. degrees in electrical engineering from the Nanjing University of Aeronautics
and Astronautics, Nanjing, China, in 2010 and 2013,
respectively.
He is currently at Texas Instruments Semiconductor Technologies (Shanghai) Company, Ltd., Shanghai, China.
Shaojun Xie (M’05) was born in Hubei, China, in
1968. He received the B.S., M.S., and Ph.D. degrees
in electrical engineering from the Nanjing University
of Aeronautics and Astronautics (NUAA), Nanjing,
China, in 1989, 1992, and 1995, respectively.
In 1992, he joined the Faculty of Electrical Engineering, Teaching and Research Division, NUAA,
where he is currently a Professor with the College
of Automation Engineering. He has authored more
than 150 technical papers in journals and conference
proceedings. His main research interests include aviation electrical power supply systems and power electronic conversion.