An Interleaved Buck-Boost-Converter Combined
with a Supercapacitor-Storage for the Stabilization
of Automotive Power Nets
Johannes Kloetzl1), Dieter Gerling2)
FEAAM GmbH / 2)Universitaet der Bundeswehr Muenchen
Werner-Heisenberg-Weg 39
Neubiberg, Germany
Tel.: +49 (0) 89 - 6004-4726
Fax: +49 (0) 89 - 6004-3718
E-Mail: johannes.kloetzl@unibw.de
URL: http://www.feaam.de
Abstract-The quality of the supply voltage in automotive power
nets is deteriorating because of the electrification of auxiliary
components with high and dynamic power demands. This
electrification, on the one hand, serves the reduction of CO2emissions and, on the other hand, it is required in vehicles with
an electric drive train. For the stabilization of the supply voltage
active units can be used. In this paper the construction and first
simulation results of an interleaved 2-phase buck-boost
converter in combination with a supercapacitor-storage for this
purpose are described. Furthermore insight into the control and
the operational strategy is delivered.
I.
INTRODUCTION
The voltage of low voltage (LV) automotive power nets
(PN), normally rated at about 14 V, is getting more and more
unstable [1]. One main reason for this is the increasing level
of electrification. In vehicles with a combustion engine a lot
of auxiliary components, like power steering, fuel pump or
air-conditioning compressor, are electrified, because it is
inefficient to operate them in a fixed ratio with the engine
speed. With electrified components the power consumption
gets correlated with the actual power demand. This is an easy
way to reduce fuel consumption and thus CO2-emissions [2].
In vehicles with electric or hybrid drive train the
electrification of these and other components is an absolute
requirement, because their function is also needed when the
combustion engine, as far as existing, is stopped.
The power demand of these and other components, like driver
assistance systems, can be very dynamic and in superposition
very high. In vehicles with combustion engines the generator
is not able to follow this demand, because its dynamic is
limited through the current construction type. So it is buffered
by the battery, combined with voltage deviations depending
on the battery’s size and condition. In vehicles with electric
drive train the LV-power net is normally energized by the
high voltage (HV) power net via a DC/DC-converter. These
converters are much more dynamic in their power conversion
ability so that the battery size can be reduced. But they are
not designed to cover the maximum power demand of the LV
power net because of cost, space and efficiency reasons. As a
consequence critical voltage deviations can also occur here.
978-1-61284-246-9/11/$26.00 ©2011 IEEE
Too high voltage deviations can cause malfunctions or a
breakdown of the ECUs (electronic control units), so they
have to be limited.
Out of the definition for automotive power net stability given
in [3] a great number of options for stabilization, passive as
well as active, can be obtained. One of these is that every
component has to be able to cover its power demand itself. So
a stability level of 100 % could be reached, independent of
the vehicle configuration. A first step in this direction would
be stabilization units assigned to single components. The
exact design can differ from low power support (the unit can
just limit the voltage deviations) to full power support (the
unit can cover the full power demand).
The device presented in this paper will be used to close or to
reduce the gap between power demand and power generation
and therefore reduce voltage deviations to an uncritical level.
The energy needed to fill this gap will be stored in an
electrochemical double layer capacitor (EDLC) stack which
is charged/discharged by a DC/DC-converter. The device is
assigned to one single component (a driver assistance
system).
II.
CONCEPT
Devices like the one mentioned above and shown in Fig. 1
already exist in principle in different variations as presented
e.g. in [4]. They differ, e.g., in the DC/DC-topology or in the
used storage voltages and capacities. The main differences to
the concept presented here are the choice of the capacity and
the designated application. It will be assigned to a special
component and not to the whole power net or a bigger part of
it. And it will be able to supply the component in stand-alone
in case of a breakdown of the power net, e.g. in case of a
DC
LV PN
DC
Fig. 1. Principle concept of the device
EDLC
1)
failure of the DC/DC-converter between HV and LV power
net in full electric vehicles. So a controlled and save
shutdown of the vehicle will be possible if multiple devices
like this are used for every critical component of the vehicle.
Consequently, the size of the LV-PN battery can be reduced
and chosen independently from the electrical configuration of
the vehicle, because the battery size can mainly be chosen
according to the power demand for starting the engine.
Furthermore it offers additional possibilities like the
reduction of the wiring harness, because ideally it just has to
deal with the average current of the component.
The EDLC-stack voltage is chosen lower than the rated
PN-voltage. This enables to use a simple DC/DC topology
and makes the control strategy very easy: Step up operation
always leads to a power net stabilization and step down
always charges the stack.
III.
CONSTRUCTION
A. EDLC-Stack
The EDLC-stack’s capacity is designed to store enough
charge for a stand-alone supply of a component, in this case a
driver assistance system, in an acceptable voltage range. For
this, the worst case current demand of this component was
measured and integrated over time. The needed energy at a
reduced voltage level of 12 V amounts to about 600 Ws. With
a supposed average efficiency of the DC/DC-Converter of
about 70 %, 870 Ws are needed to be stored in an acceptable
voltage range.
For this a capacity of 37.5 F was chosen and is realized
with 12 capacitors of 50 F each, connected four times in
series and three times in parallel. This needs a little bit more
space than a 4 times serial connection of 150 F capacitors but
the internal resistance is reduced by about 30%. A low
internal resistance is strictly required to be able to transfer the
needed power out of the stack into the PN. To reduce the risk
of overvoltage at single capacitors and for a longer lifetime of
the stack, a charge balancing unit is indispensable [5]. For
this purpose, a circuit similar to [6] is realized. In the chosen
design it balances charge differences in about 50 s. But it
causes a high self-discharge of the stack in continuous
operation so that it has to be switched off after balancing. The
maximum voltage of the stack is chosen to 10.5 V and it is
not discharged below 8 V in normal operation.
B. DC/DC-Converter
Because of the chosen maximum voltage, the voltages of
the stack and the power net overlap completely and so the
stack can be easily charged with a normal buck converter.
The discharge of the stack and thus the stabilization of the
power net voltage can be done with a simple boost converter.
A galvanic separation of the stack and the PN is not useful,
because an automatic power transfer to the PN would not be
possible in case of a voltage drop in the PN below the actual
stack voltage. Because of these reasons, the low number of
active parts and the capability for bidirectional energy
transfer a buck-boost converter is chosen as DC/DC converter
for the device. Furthermore a high efficiency could be
assumed for this converter type.
Because of the high space requirements for vehicle usage,
the size of the converter has to be reduced as far as it is costeffective. So a high switching frequency of about 130 kHz
was chosen to reduce the size of the converters inductor. A
high switching frequency is also needed to enable a
sufficiently fast compensation of voltage deviations and to
reduce the size of the filtering elements. A further reduction
of these elements is reached through a splitting of the
converter into two phases which are controlled with a phaseshift of half a period; generally known as interleaving [7]. So
a massive reduction of the EMI-filters is possible, if the phase
shift is strictly adhered to [8].
Because all the internal resistances of the EDLC-stack, the
inductor and the switches limit the maximum output power in
boost operation and consequently the stabilization capability
of the module, every part is chosen with a focus on low
internal resistances. Therefore the switches are oversized for
about three times concerning the ampacity. This also leads to
reduced conduction losses, but of course at higher costs. To
further decrease conduction losses the switches of every half
bridge are driven by complementary gate signals. This
synchronous rectification unloads the freewheeling (body)
diodes, which normally have a much higher RDSon than the
switches, almost for the whole time. The freewheeling diodes
are only used in a short time in which both switches of the
half-bridge are not driven to avoid shoot-through effects. In
addition, the converter always operates in continuous
conduction mode which simplifies the control of the
converter, because the nonlinear control areas shrink [9].
Because of all the reasons presented above similar
topologies are often chosen for automotive applications [8]. A
simple schematic of the converter’s topology is shown in
Fig. 2.
C. Measurements
For the control of the converter the measurement of both
voltages, VPN and VEDLC, is needed. Furthermore a current
measurement is needed for a faster control. In this case one
current sensor per phase is used, because during operation the
currents in the different phases can differ massively [10],
Fig. 2. Schematic of a two phase interleaved buck-boost converter
even when they are absolute symmetric. This leads on the one
hand to an unbalanced loading of the inductors and the
switches, which has bad influences on durability and
efficiency. On the other hand it causes EMI problems because
the filters are designed in assumption of balanced currents
and consequently a perfect ripple cancelation. For
guaranteeing a good ripple cancelation here IL1 and IL2 are
measured.
IV.
CONTROL AND OPERATIONAL STRATEGY
A. Boost Control
For the design of the control the average model of the
converter can be used. The design of the control loop will be
described here exemplarily for the boost control, equivalent
to a power transfer from the EDLC-stack to the PN. The
average model of the almost ideal converter can be described
in the frequency domain for a one phase converter in the
following way [11]:
VL= VEDLC – VTr = VEDLC – VBN (1–D)
(1)
ID = IL – ITr = IL (1-D)
(2)
ID – IPN = IC
(3)
UPN = IC / (CPN s)
(4)
IL = (1/RL) / (1+L/RL s) UL.
(5)
D = ton / (ton + toff).
(6)
Tr indicates here the switching transistor and D the
freewheeling transistor. D itself is the duty cycle and RL the
internal resistance of the inductor. The signs of the currents
are here the inverted to Fig. 2. The resulting plant structure is
shown in Fig. 3.
The nonlinearity at the input of the plant [12] and the
following disturbance caused by the output voltage can be
compensated easily, because VPN as well as VELDC are
measured. In contrast, the multiplication of IL with (1-D)
cannot be compensated because it is located in the inner part
of the path. For the design of the controller it is assumed to be
constant here. The resulting inaccuracy is acceptable, because
(1-D) varies in normal operation in a range of 0.2 to 0.6. The
chosen constant value is arranged inside this range.
For a fast control of the power net voltage a cascade
control is chosen here. The inner part controls the inductors
Fig. 3. Plant structure for a single phase boost-converter
current IL, the outer part the converters output voltage. For the
current control loop a dead time for the switching delay and a
PT1-filter of the current measurement have to be considered.
This filtering is needed to eliminate the switching frequency
and its harmonics in the current, because the control is based
on the average model. Both is considered as one combined
time constant. So the plant for the current controller can be
modeled as PT2-type. For this case a simple PI-controller is
chosen and designed by optimum magnitude. The suboptimal
disturbance response of the resulting control loop is not
relevant here, because the single disturbance (VELDC) should
be almost compensated.
In the chosen converter topology there are two parallel
phases which correspond to two parallel current plants. Here
two independent current controllers can be chosen, because
two current sensors were implemented. A functional
demonstration of this concept will be presented in the next
chapter. This concept requires a drive which is able to vary
the duty cycle of the two phases independently.
For the higher-level voltage controller this concept has no
relevant consequences, because the currents ID1 and ID2
superimpose to IC. Furthermore both current controllers get
the identical desired value IL*. So it is just a question of an
adjustment of the voltage controller’s gain. The resulting
plant of for the voltage controller is of IT1-type so again a PIcontroller is chosen to guarantee stability. The complete
resulting control loop with the described compensations is
shown in Fig. 4.
A. Buck Control
The plant for the buck control is very similar. In case of the
current plant it is identical except that the manipulated
variable is D instead of (1-D). It is also controlled via a PIcontroller. In the voltage plant the multiplication with (1-D)
does not exist, so after the same compensations as shown for
the boost control the plant is totally linear. Furthermore for
the output capacitor, which is the EDLC-stack in this case,
the internal resistance is considered. Consequently the
capacitors transfer function is of PI-type instead of just I-type.
For the voltage controller a P-type controller is chosen,
because of the high capacity of the output capacitor. The
control should be comparatively slow to limit the charge
current and to not load the PN too intensively. This would be
absolutely counterproductive. The gain of the P-Controller
can be easily used for this purpose, because the controllers
Fig. 4. Control loop structure for a two phase boost-converter
output is the desired inductor current which corresponds to
the capacitors charge current in buck mode.
B. Operational Strategy
The operational strategy defines the global behavior of the
complete unit. It is based on state machine with 4 states at
present. There is one state for buck and boost operation each,
an idle or standby state and a fail state. In normal operation
the idle state is the central state which is used at every change
between buck and boost mode. One exception is the transition
to the fail mode which is possible from every state, e.g.
because of overcurrent. Another exception is presented later.
The principle structure is shown in Fig. 5.
There is no off-state implemented at present, because the
unit cannot really be switched off. In idle state, when all
switches are open, nevertheless a current can flow out of the
unit into the power net, if the voltage of the power net is
lower than the voltage of the storage capacitors. This function
is desired, as mentioned above, but it can lead to problems in
maintenance. So such a function is in consideration at the
moment. Mainly it is just about adding one more switch,
either between the PN and the unit or between the DC/DCconverter and the storage.
In principal synchronous rectification would enable the
converter to change directly between buck and boost mode
and the other way round, because the actual operation mode
is just a function of the duty cycle. This ability is not used
here because of three reasons. First it is assumed that two
independent controllers are better to optimize for their
particular use than one single controller. Second it is more
efficient, because it is avoided that the converter reaches an
operating point in which only low power or, in worst case, no
power is transferred but switching and low conduction losses
are produced. Third and mainly overshooting e.g. in the case
of a load dump is reduced, because not the controller has to
compensate the load dump. It is just done, because the unit is
set into idle state and no more output current is generated as
long as the voltage stays higher than the activation level.
In the buck mode the buck control mentioned above is
active. The mode gets activated if the SOC of the stack is
under a specified limit and the terminal voltage is higher than
the present desired value of the power net voltage minus a
constant value. It gets deactivated if the terminal voltage is
lower than the present desired value of the power net voltage
minus another greater constant value or the SOC reaches an
upper limit. Because the present buck control is optimized for
charging the stack and it is consequently slow the buck mode
cannot be used to effectively reduce overvoltage. This
function is not the primary focus of the unit so that this
function will be added in a further step with a second buck
controller.
In the boost mode the boost control mentioned above is
active. The boost mode gets activated when the SOC is higher
than a specified limit and the terminal voltage is lower than
the deactivation voltage of the buck mode. It gets deactivated
if the terminal voltage reaches a voltage higher than the
activation voltage but lower than the buck activation voltage,
if there is a current flow into the stack detected or if the SOC
gets too low for a sufficient functionality. So a permanent
change between buck and boost mode is avoided. Because of
the same reason there is a minimum time defined when the
boost mode has to be active. This function would be also
useful in buck mode but it is not implemented because it
could lead to a delay in the voltage stabilization when a
voltage drop occurs.
One exception for the transition between states was
mentioned before. There exists one unidirectional transition
between buck and boost mode. This transition gets active
when during the buck mode a high negative edge in the
terminal voltage is detected. In this case no time is wasted by
changing first to idle and then to boost mode.
Another characteristic of the operational strategy is that the
states of the PI controllers in boost mode are not initialized as
zero and reset to this initialization values between two
activations. This leads to a faster reaction in the boost mode
and it is avoided that the converter passes through a range of
(1-D) where the output power is very high. A possible
occurring overshoot of the output voltage is accepted here. In
buck mode the procedure is similar with a similar the
background. Here a possible pass through of a short time
boost operation should be avoided or damped. This behavior
is just normal for converters with synchronous rectification,
because the operation mode is a function of D and if the states
of the controllers are zero at startup the duty cycle can pass
through values that cause the wrong operation. In buck mode
the controller states are not reset between two activations in
contrast to boost mode.
At present the operational strategy has no information
about the activation of the component it is assigned to. This
option will be added later, because it can help to improve the
system response. If the component informs the unit about its
activation a kind of feedforward control becomes possible.
V.
Fig. 5. Principle structure of the operational strategy
SIMULATION AND RESULTS
In this chapter some examples for the functionality and the
current capability will be given. All simulations were done
with Matlab Simulink. The electrical part is modeled with
SimPowerSystems Toolbox. The parasitic values of all parts
were considered as far as available. The controllers were
Fig. 6. Simulation results of output power test
implemented as continuous, only for the operational strategy
a time discrete implementation was done.
The first simulation was done to check the output power of
the unit. For this it was modeled in standalone operation, so
without a coupling to a power net, just coupled to a switched
resistive load of 0.2 Ω. The maximum output current the unit
is designed for is 60 A at 12 V. The initial voltage of the
stack was 9.6 V which corresponds with a SOC of 84%. Just
a rudimental controller design was used here, so no statement
about dynamic can be given out of this simulation. The
results for the output voltage and the output current are shown
in Fig. 6. As it can be seen the required 60 A are reached
without problems at the rated voltage. The overshooting after
the load is switched off is very high, because the full output
current charges into the output capacitor. In coupling with a
power net, the overshooting would be lower but still
problematic. In full operation it is reduced by the operational
strategy as presented before, because the boost controller is
switched off at the beginning of the overshoot.
The inductor currents for a very similar simulation without
a balancing control are shown in the upper part of Fig. 7. In
this simulation only some small changes in SOC, etc. were
done to show a worse case. Both inductors have absolute
identical parameters. In the lower part the results with the
current balancing control described above are presented. As it
can be seen the current balancing works properly. Almost no
Fig. 7. Comparison of inductor currents without (upper part) and with (lower
part) balancing control
difference between IL1 and IL2 can be seen anymore.
Furthermore can be seen, that in case of no load currents the
converter is active and there are inductor currents with a
mean value of zero, which are only causing needless losses.
This is also avoided by the operating strategy.
Another example shows what effect the presetting of the
controller states has. For this the simulation results of a
similar simulation are shown. The difference is that in this
case the described operational strategy is partly implemented,
so that the controllers are only active, if the corresponding
state is too. The controller parameters are still the rudimental
ones. The unit now is coupled to a simple power net model
consisting of simple battery model with 13 V no-load voltage
and some line resistances. Again a load of 0.2 Ω is switched
on. The controller states are preset to zero at the beginning.
After switching off the states are not reset and so the control
behavior is much better at the second switch on, as it can be
seen in Fig. 8.
It is also observable, that in the boost mode the output
voltage of the unit overshoots over the desired value of 12 V
during the first switching. This is caused by the controller
which first has to pass through a range of (1-D) with higher
output power than needed. The same reason would cause a
short time of boost operation in buck-mode. This cannot be
seen here, because the states of the buck-controller were
perfectly preset in this example, so that it starts right in buck
operation at 0 s. For sure the controller preset will not fit in
every case as perfectly as in the chosen example, but it can
definitely help to improve the system response especially for
high output currents. The best results can be achieved with
switched loads of almost constant amplitude. This matches
with the characteristic of the assigned driver assistance
system.
This characteristic can be seen in Fig. 9. In this example a
Fig. 8. Comparison of the system response depending on the controller state
preset
they are too short for e.g. causing a reset of the ECU.
Furthermore there is seen potential to reduce these deviations
in the further design process.
VI.
CONCLUSION AND OUTLOOK
The concept and the construction of a unit for the
stabilization of the supply voltage of one assigned component
in automotive power nets were presented in this paper.
Furthermore the control and the operational strategy were
elucidated in detail. The functionality of the unit was
validated successfully with several simulation results.
At the moment a prototype is built almost completely and
tested in parts. The implementation of the control on a DSP
and following system tests are planned. In the end the unit
will be tested on the test bench shown in [13]. If these tests
are successfully the concept will be transferred to other
components.
Fig. 9. Comparison of the main power net sizes with and without the use of
the described unit
full simulation is shown. The complete operational
characteristic and faster controller parameters are
implemented. The load is implemented as a controlled current
source which is driven by a measured current profile. In
Fig. 9 the main sizes of the PN are presented in comparison
with and without the active unit during the startup procedure
of the driver assistance system. As it can be seen, the voltage
deviation is reduced significantly. The voltage is only for
short times a little bit lower than the rated 12 V for boost
operation. The stability reserve as it is defined in [3] is
doubled from 32 % to 65 % for a rated PN voltage of 13 V.
In standalone operation of the unit, e.g. in case of a failure
in the wiring harness, the functionality of the system can be
guaranteed too, as it can be seen in Fig. 10. The dynamic
behavior is not as satisfying as it is in interaction with an
active power net, but still acceptable. There occurs
overshooting if there is a high current gradient, but its
amplitude is uncritical for automotive components. The short
low voltage peaks also should cause no problems, because
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Fig. 10. Terminal sizes of the unit in standalone operation
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