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IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 1, JANUARY 2006
Development of Multiband Phase Shifters
in 180-nm RF CMOS Technology
With Active Loss Compensation
Chao Lu, Student Member, IEEE, Anh-Vu H. Pham, Senior Member, IEEE, and Darrell Livezey
Abstract—We present the design and development of a novel integrated multiband phase shifter that has an embedded distributed
amplifier for loss compensation in 0.18- m RF CMOS technology.
The phase shifter achieves a measured 180 phase tuning range in
a 2.4-GHz band and a measured 360 phase tuning range in both
3.5- and 5.8-GHz bands. The gain in the 2.4-GHz band varies from
0.14 to 6.6 dB during phase tuning. The insertion loss varies from
3.7 dB to 5.4-dB gain and 4.5 dB to 2.1-dB gain in the 3.5- and
5.8-GHz bands, respectively. The gain variation can be calibrated
by adaptively tuning the bias condition of the embedded amplifier
to yield a flat gain during phase tuning. The return loss is less than
10 dB at all conditions. The chip size is 1200 m 2300 m including pads.
Index Terms—CMOS analog integrated circuits
distributed amplifiers, phase shifters, phased arrays.
(ICs),
I. INTRODUCTION
M
ULTIPLE input and multiple output (MIMO) transceivers have recently gained attention for the development of broad-band wireless applications. As a special case of
MIMO, adaptive phased-array antennas can effectively combat
co-channel interferences and deal with multipath fading [1]. By
controlling the time delay and gain of the signal in each antenna
independently, phased-array antennas can form beams and
nulls in desired directions. This kind of beamforming improves
antenna gain to yield higher signal-to-noise ratio and provides
spatial diversity for higher data rate transmission.
RF phase shifters are key elements of analog phased-array
antennas and have been mostly implemented in GaAs integrated circuits (ICs). Recently, Si-based RF phase shifters have
emerged as a new platform for wireless integrated transceivers
[2]–[5]. Both digital and analog phase shifters have been
demonstrated in RF silicon ICs. A 6-bit digital phase shifter
was reported in SiGe technology over a 7–11-GHz frequency
band [2]. A dual-band 5.2/2.4-GHz 4-bit phase shifter designed
by combining two single-band phase shifters was reported in
[3]. Low- factor passive devices and a small tuning ratio of
varactor capacitance (typically 2 4) in CMOS represent challenges in the implementation of multiband continuous phase
Manuscript received April 22, 2005; revised August 30, 2005. This work was
supported in part by UC MICRO, by Tahoe RF Semiconductor Inc., and by the
National Science Foundation under Contract ECS-0401375.
C. Lu and A.-V. H. Pham are with the Department of Electrical and Computer
Engineering, University of California at Davis, Davis, CA 95616 USA (e-mail:
cflu@ucdavis.edu; pham@ece.ucdavis.edu).
D. Livezey is with Tahoe RF Semiconductors Inc., Auburn, CA 95602 USA
(e-mail: dlivezey@tahoeRFsemiconductor.com).
Digital Object Identifier 10.1109/TMTT.2005.860892
Fig. 1. Circuit topology of a multiband phase shifter with broad-band active
loss compensation.
shifters with low loss and large phase tuning ranges. A continuous phase shifter with a 105 phase tuning range at 2.4 GHz
has recently been reported in 0.18- m CMOS technology [4].
A 360 phase shifter at 8 GHz consuming 170 mW of power
was proposed in [5] by embedding a varactor-tuned LC ladder
network between driving amplifiers.
In this paper, we present the development of a multiband
RF CMOS phase shifter that employs a distributed amplifier
for loss compensation since the conventional loss compensation
methods, such as negative resistance [4], become ineffective for
broad-band and multiband applications. In this phase shifter,
two varactor-tuned LC networks have been designed to provide
both phase shifting and broad-band impedance matching. The
impedance-matching networks allow the distributed amplifier to
have high input and output impedance for the reduction of power
consumption. The distributed amplifier compensates loss from
2.4 to 6 GHz and provides the calibration of gain variation. The
phase shifter achieves a measured continuous 180 and 360
phase tuning at 2.4- and 3.5/5.8-GHz bands, respectively. The
gain is as high as 6.6 dB without calibration and the return loss is
less than 10 dB from 2.4 to 6 GHz. The phase shifter requires
only one phase control bias and, therefore, one digital-to-analog
converter (DAC). To the best of our knowledge, this is the first
multiband continuous phase shifter reported in RF CMOS.
Section II describes the architecture of the proposed embedded amplifier topology. The detailed analysis of distributed
amplifiers and phase transmission functions are presented.
Section III describes the prototype implementation and measurements of the multiband phase shifter.
II. PHASE-SHIFTER CIRCUIT TOPOLOGY
Fig. 1 illustrates the multiband phase shifter that has a distributed amplifier for broad-band loss compensation. The proposed topology in this design yields significant loss compensation with less power consumption by increasing the characteristic impedance of gate and drain lines of the distributed amplifier to be above 50 . Varactor-tuned LC networks are employed
0018-9480/$20.00 © 2006 IEEE
LU et al.: DEVELOPMENT OF MULTIBAND PHASE SHIFTERS IN 180-nm RF CMOS TECHNOLOGY
41
Fig. 2. Cascaded single-stage distributed amplifier.
to achieve a compact chip size and to provide broad-band 50matching and phase tuning. Though the internal characteristic
, the input
impedance of the distributed amplifier is
and output impedances of the whole phase-shifter circuit are
maintained to be around 50 during phase tuning.
A. Distributed Amplifiers
In distributed amplifiers, gain stages are connected so that
output currents are combined coherently while their capacitances are synthesized in parallel to form artificial transmission
lines, namely, gate and drain lines. The artificial transmission-line topology gives distributed amplifiers a wide bandwidth. Due to its gain advantage over conventional distributed
amplifiers, a cascaded single-stage distributed amplifier [6]–[8]
is employed in this design. When the operation frequency is far
below the cutoff frequency, the gain of a distributed amplifier
with cascaded single stages is simplified as [6]
Fig. 3. Schematic diagram of varactor-tuned LC network.
B. Varactor-Tuned LC Network
Fig. 3(a) shows a conventional varactor-loaded transmissionline phase shifter, where several lumped-element -sections are
cascaded to implement an equivalent transmission line for the
compact chip size. Similar to that in [9], the transmission phase
of a single -section can be derived as
(2)
When
tions is
, the transmission phase of
-sec-
(1)
(3)
is the transconductance of each stage, and
and
where
are the characteristic impedances of gate and drain artificial
transmission lines, respectively. To enhance gain, the number of
of each stage can be
stages ( ) or the transconductance
increased with a penalty of more power consumption. Furthermore, the gain can also be increased by designing high interstage impedances, but significantly sacrificing bandwidth [8].
Our strategy is to increase the characteristic impedances ( )
of gate and drain lines at the input and output of the amplifier to
achieve high gain, while the inter-stage impedances are maintained at the intermediate level, i.e., 100 . If we define the
, the power consumpimpedance transformation ratio
tion of distributed amplifiers can be theoretically reduced by
times for the same gain compared with
for
design. We have chosen
for both
and
to balance power consumption and bandwidth. Fig. 2
shows a three-stage cascaded single-stage distributed amplifier
where the gate/drain characteristic impedances are designed to
and the gate/drain parasitic
be equally 100
capacitances of transistors are adopted into the artificial transmission lines.
The simulated results show that the distributed amplifier can
yield 14–15-dB gain up to 8 GHz. The distributed amplifier
draws maximum 25-mA current from 1.8-V power supply and,
therefore, the maximum power consumption is 45 mW.
where
is the operation frequency. The group delay is then
(4)
This group delay is independent on frequency and is referred as
a true time delay, and this delay can be adjusted by changing inductance or capacitance or both. Therefore, varactor-load transmission-line phase shifters are also suitable for broad-band applications [10].
For a given capacitance tuning ratio , which is defined to be
, the relative phase shifting is controlled by varying
the capacitance . The relative transmission phase tuning range
is
(5)
We can see that the phase tuning range increases with the section
number . For a typical silicon-based varactor, the capacitance
tuning ratio is typically 2 4. To reach a symmetrical variation,
we specify [11]
(6)
42
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 1, JANUARY 2006
Fig. 4. Peak Q frequency of spiral inductors and cutoff frequency of
-sections in 180-nm RF CMOS technology.
Assuming
, we obtain
(7)
parameter variations resulting from device mismatching. Each
varactor-tuned LC network in Fig. 1 is composed of the circuit
shown in Fig. 3(b). The input varactor-tuned LC network
, and the output one transforms
transforms 50 to
to 50 . These two varactor-tuned LC networks
employ an identical architecture and can share one control
voltage because of the symmetry.
C. Hyperabrupt (HA) Varactor
At these worst conditions, the return loss is calculated as
dB
Fig. 5. Tuning capacitance of MOS varactors and HA varactors at 3 GHz.
(8)
This indicates that the phase shifter can achieve a good
impedance matching through the full phase tuning range.
From (5), we can also see that the phase tuning range will
increase as the inductance value increases in each section.
However, a larger nominal capacitance value of varactors is
required to maintain the characteristic impedance around
and, therefore, the cutoff frequency of the -sections is degraded, as shown in Fig. 4. This figure also indicates the peak
quality factor ( ) frequency of spiral inductors drops as the
inductance increases. To alleviate the effect of parasitic capacitance related with spiral inductors and satisfy the condition of
, we chose the inductance
nH and,
therefore,
pF. At these conditions,
pF
pF. The desired phase tuning range can be
and
achieved by appropriately designing section number . The
section number should be at least 20 to yield 360 phase
tuning range at 3.5 GHz.
One contribution of this study is to design a varactor-tuned
LC network that provides both phase shifting and broad-band
matching. To provide broad-band impedance matching,
three-section Chebyshev impedance transformers [12] [see
Fig. 3(b)] have been designed, and each section is composed
of an equal “length” lumped varactor-loaded transmission
line [see Fig. 3(a)]. Each section has a different characteristic
impedance, i.e., 57.4, 70.7, and 87.1 , which are determined by
Chebyshev polynomials. Since the frequency band of interest
GHz in the design of multiis 2–6 GHz, we chose
section Chebyshev transformers. The three-section Chebyshev
transformer has 12 lumped -sections and, therefore, 12 inductors. All inductors are kept uniformly to be 1.0 nH, and
all capacitors are comprised of the same unit varactor with
different multipliers. Therefore, the circuit is insensitive to
RF CMOS technology typically offers two options for varactor implementation: MOS accumulation varactors and HA
junction varactors. The HA varactor utilizes a retrograde implant to modify a standard pn junction that results in a nonlinear
n-type dopant across the depletion region. As the reverse bias is
increased, the nonlinear doping profile causes a greater capacitance change in the HA junction diode and greatly enhances the
tunability and linearity [13]. HA varactors have a larger tuning
ratio and better capacitance–voltage (CV) linearity compared
with MOS varactors, as shown in Fig. 5. The linear capacitance
control property relaxes the resolution requirement on DACs.
The tuning ratio of HA varactors is as high as 5 with the control voltage changing from 0 to 7 V. However, as the control
voltage approaches at a high value, the linearity of capacitance
versus control voltage is degraded. In this design, the control
voltage from 0 to 4.5 V is chosen to result in a tuning ratio of 4.
D. Gain Variation Calibration
to yield difAs varactors are tuned and controlled by
ferent phases, the quality factor of the varactors and the cutoff
frequency of varactor-tuned LC networks will vary and cause
changes in the insertion loss of the phase shifter. Generally,
this kind of gain shift, as well as that resulting from temperature drifting needs to be calibrated in most applications [14],
and the calibration is conventionally performed through a variable attenuator. Our proposed architecture provides an option
for gain calibration by adaptively adjusting the bias condition
of the embedded amplifier, as indicated by (1). For the operation frequencies far less than the cutoff frequency of the artificial gate and drain transmission lines, the transmission phase of
the distributed amplifier is independent on the bias condition,
as long as all transistors are maintained in the saturation region.
Therefore, the gain (loss) calibration will not impact the transmission phase of the phase shifter. The calibration can be conducted to yield a flat gain by using a simple lookup table [3].
LU et al.: DEVELOPMENT OF MULTIBAND PHASE SHIFTERS IN 180-nm RF CMOS TECHNOLOGY
Fig. 6.
Die photograph of the multiband phase shifter in 0.18-m RF CMOS.
Fig. 8.
43
Measured transmission phase at 2.4, 3.5, and 5.8 GHz.
TABLE I
COMPARISON OF PHASE TUNING RANGE WITH V
Fig. 7.
FROM 0 TO 4.5 V
Measured: (a) input return loss and (b) output return loss.
III. MEASUREMENT RESULTS OF PHASE SHIFTERS
The novel phase shifter was implemented in IBM 180-nm RF
CMOS. The varactor-tuned LC networks are on the left- and
right-hand sides, while the embedded distributed amplifier is
in the middle of the chip. The chip size is 2.3 mm 1.2 mm
including pads. As shown in Fig. 6, the grounding path and all
biases are connected onto an evaluation board using bond wires.
A.
-Parameter Measurements
-parameter measurements were conducted at room temperature using a Cascade Microtech probe station and an Agilent
E8364B Performance Network Analyzer.
Fig. 7 shows that the measured return loss is less than 10 dB
from 2.4 to 6 GHz with control voltage
tuning from 0 to
measurements are focused on three frequency
4.5 V. The
bands, i.e., 2.4–2.48, 3.4–3.5, and 5.73–5.83 GHz, and the meafrom 0 to 4.5 V
sured relative phase tuning ranges with
are 220 , 360 , and 660 , respectively (Fig. 8). The measured
Fig. 9.
Measured S
in 2.4-GHz frequency band.
phase tuning ranges are consistent with those in the simulation,
as compared in Table I. Fig. 8 indicates a nearly linear relation, as predicted
ship between the relative phase shifting and
in Section II.
For narrow-band applications, the phase variation at a given
control voltage versus operation frequency is an important
specification for continuous phase shifters. The phase variation
versus frequency determines the achievable resolution and,
therefore, the performance of phased array antennas. The
-parameter measurements show that the phase variation for
20-MHz bandwidth is 3 , and the gain ripple within the same
bandwidth is less than 0.1 dB, as shown in Figs. 9–11. The
results shown below are based on the measurements with the
180 and 360 continuous phase tuning ranges in 2.4- and
3.5/5.8-GHz bands, respectively.
44
Fig. 10.
Fig. 11.
IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 54, NO. 1, JANUARY 2006
Measured S
Measured S
Fig. 12.
Calibration results at 2.4 GHz.
Fig. 13.
Calibration results at 3.5 GHz.
Fig. 14.
Calibration results at 5.8 GHz.
in 3.5-GHz frequency band.
in 5.8-GHz frequency band.
Fig. 9 shows that the relative phase tuning range is 180 in
the 2.4-GHz band as
changes from 0.1 to 3.0 V, and the
gain varies from 0.1 to 6.6 dB. This result indicates that the
insertions loss caused by lossy CMOS passive devices has been
significantly compensated.
The phase tuning range in the 3.5-GHz band is 360 as
changes from 0 to 4.0 V, and the insertion loss is reduced from
3.7- to 5.4-dB gain, as shown in Fig. 10. The phase tuning
changes from 1.0
range in the 5.8-GHz band is 360 as
to 4.5 V. The insertion loss is also compensated to vary from
4.5 dB to 2.1-dB gain during the phase tuning, as illustrated
by Fig. 11.
The distributed amplifier provides a constant gain of 14 dB up
to 8 GHz. However, the loss of the phase shifter and matching
networks increases significantly at 5.8 GHz and up to 18 dB.
Hence, the compensated gain at 5.8 GHz is less than that at 2.4
and 3.5 GHz.
B. Gain Variation Calibration
As we can see from the -parameter measurement results, the
changes for difgain (loss) of the phase shifter varies as
ferent phase conditions. The gain variation of the phase shifter
can be calibrated by adaptively adjusting the gate bias of the
distributed amplifier through a lookup table. The calibration results at 2.4 GHz is shown in Fig. 12. We can see that a flat gain
around 0 dB is achieved compared with the gain shifting from
0 to 6.6 dB before the calibration. On the other hand, the phase
tuning versus control voltage curve with the calibration is almost identical as that without the calibration through the whole
180 tuning range.
LU et al.: DEVELOPMENT OF MULTIBAND PHASE SHIFTERS IN 180-nm RF CMOS TECHNOLOGY
The gain variation calibration has been also conducted at 3.5
and 5.8 GHz, as shown in Figs. 13 and 14, respectively. A flat
insertion loss is achieved at 3.7 and 4.5 dB, respectively,
across the 360 phase tuning range. Similar to that at 2.4 GHz,
the transmission phase is almost unaffected through the calibration process. The independence of phase tuning on gain calibration simplifies the design of control (calibration) algorithms.
IV. CONCLUSION
This paper has presented the development of a multiband
phase shifter in 180-nm CMOS technology. The phase shifter
requires only one control voltage for phase tuning. The proposed novel topology yields significant loss compensation with
moderate power consumption [5], [15], [16]. The measurement
results have shown that the phase shifter can achieve more than
180 tuning range over 2.4- to 6-GHz bands. The phase variation
within 20-MHz bandwidth is 3 , and the time-delay nature
also makes this phase shifter suitable for wide-band applications. The gain can be as high as 6.6 dB without calibration. The
gain variation during phase tuning can be calibrated by adaptively adjusting the bias condition of the embedded amplifier.
The calibration through a lookup table has been conducted to
yield a flat gain through the whole phase tuning range.
[11] F. Ellinger, H. Jackel, and W. Bachtold, “Varactor-loaded transmissionline phase shifter at C -band using lumped elements,” IEEE Trans. Microw. Theory Tech., vol. 51, no. 4, pp. 1135–1140, Apr. 2003.
[12] D. M. Pozar, “Impedance matching and tuning,” in Microwave Engineering, 3rd ed. Hoboken, NJ: Wiley, 2005, ch. 5, pp. 250–255.
[13] J. S. Dunn, D. C. Ahlgren, D. D. Coolbaugh, N. B. Feilchenfeld, G.
Freeman, D. R. Greenberg, R. A. Groves, F. J. Guarin, Y. Hammad, A.
J. Joseph, L. D. Lanzerotti, S. A. St. Onge, B. A. Orner, J.-S. Rieh, K.
J. Stein, S. H. Voldman, P.-C. Wang, M. J. Zierak, S. Subbanna, D. L.
Harame, D. A. Herman, Jr., and B. S. Meyerson, “Foundation of RF
CMOS and SiGe BiCMOS technologies,” IBM J. Res. Dev., vol. 47, no.
2/3, pp. 101–138, Mar./May 2003.
[14] G. Tsoulos and M. Beach, “Calibration and linearity issues for an
adaptive antenna system,” in Proc. Vehicular Technology Conf., vol. 3,
Phoenix, AZ, May 1997, pp. 1597–1600.
[15] H. Hayashi and M. Mauraguchi, “An MMIC active phase shifter using a
variable resonant circuit (and MESFETs),” IEEE Trans. Microw. Theory
Tech., vol. 47, no. 10, pp. 2021–2026, Oct. 1999.
[16] D. Viveiros, Jr., D. Consonni, and A. K. Jastrzebski, “A tunable all-pass
MMIC active phase shifter,” IEEE Trans. Microw. Theory Tech., vol. 50,
no. 8, pp. 1885–1889, Aug. 2002.
Chao Lu (S’05) received the B.E. and M.S. degrees
in electronic engineering from Tsinghua University,
Beijing, China, in 1999 and 2002, respectively, and
is currently working toward the Ph.D. degree at the
University of California at Davis.
From 2002 to 2003, he was also a Design Engineer with Intel Technology Ltd., Shanghai, China,
where he developed mixed-signal ICs. His current
research interests include advanced communication
architectures, wide-band/multiband RF integrated
circuits (RFICs), and mixed-signal ICs.
ACKNOWLEDGMENT
The authors wish to acknowledge the IBM Corporation,
Essex Junction, VT, for the chip fabrication.
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45
Anh-Vu H. Pham (SM’03) received the B.E.E.
(with highest honors), M.S., and Ph.D. degrees from
the Georgia Institute of Technology, Atlanta, in
1995, 1997, and 1999, respectively.
In 1997, he co-founded RF Solutions, LLC, an
RFIC company that was acquired by Anadigics in
2003. He has held faculty positions with Clemson
University and the University of California at Davis,
where he is currently an Associate Professor. He is
also active as a consultant to the industry. He has
authored or coauthored over 50 technical journal and
conference papers. His research interests are in the area of RF and high-speed
packaging and signal integrity, RFIC design, and wireless sensors.
Dr. Pham serves as a member of the IEEE Microwave Theory and Techniques
Society (IEEE MTT-S) International Microwave Symposium (IMS) Technical
Program Committee (TPC) on Power Amplifiers and Integrated Circuits. He
has been the chair of the IEEE MTT-12 Microwave and Millimeter Wave Packaging and Manufacturing Technical Committee of the IEEE MTT-S. He was
the recipient of the 2001 National Science Foundation CAREER Award on millimeter-wave organic packaging.
Darrell Livezey received the B.S.E.E./C.S. degree
from the University of Colorado at Boulder, in 1986.
In 1986, he joined the Boeing Company, where
he designed electrooptic systems for aircraft. In
1996, he joined CommQuest (an IBM company)
and developed several RF and mixed-signal ICs
for telecommunication applications. Since 2003, he
has been a Senior Engineer with Tahoe RF Semiconductor Inc., Auburn, CA, where he currently
develops RF and mixed-signal ICs for telecommunication, satellite, and automatic test equipment (ATE)
applications.