16th International Power Electronics and Motion Control Conference and Exposition
Antalya, Turkey 21-24 Sept 2014
A galvanic isolated evaluation of a NTC
thermistor using a logarithmic amplifier
Yevgen Polonskiy, Denis Surmann, Christian Laudensack, Dieter Gerling
Φ
Abstract -- This paper presents a possibility to evaluate a
NTC (negative temperature coefficient) thermistor, which
exhibits an exponential behavior, using a logarithmic circuit.
The presented method allows a quicker and precise processing
compared to look-up tables or linearizing circuits. The
resistance value is converted into a galvanic isolated voltage
signal. Furthermore the presented design is optimized for a
low current consumption. Both features allow a direct
application on the power stage side with a supply given by
IGBT driver auxiliary supply.
Index Terms—Isolated circuit; logarithmic amplifier; NTC
thermistor; operational amplifier; temperature monitoring.
I.
NOMENCLATURE
B: thermistor temperature constant [°K]
k: Boltzmann constant [§1.38Â10-23 J/K]
RT: resistance of the thermistor [Ÿ]
RT,0: reference resistance value of the thermistor [Ÿ]
T: temperature of the thermistor [°K or °C]
T0: reference temperature of the thermistor [°K]
TA: (ambient) temperature of the electronics [°K or °C]
UT: temperature voltage (= kÂTA/q) [V]
q: electron charge [§1.602Â10-19 C]
II.
given in this paper describes a circuit requiring less parts
and calibration effort. The improvements are largely thanks
to the modern electronics, which provide much more
freedom and better accuracy than in the past. The presented
solution is intended to provide an easy and continuous data
processing, low current consumption and galvanic isolation
for the control part.
III. BEHAVIOR OF A NTC THERMISTOR
The thermistor is a semiconductor compound, which
resistance depends on temperature in a nonlinear way. A
simple form to describe its resistance is given by
B 1/ T −1/ T
0
(1)
RT = R0 e
To evaluate the temperature (1) is reordered
T = 1 / 1 / T + ln R / R / B
(2)
0
T
0
For the chosen IGBT module [11] the resistance behavior is
shown in Fig. 1 below.
(
(
)
(
) )
INTRODUCTION
T
he current application is a 3 phase power electronics
based on the topology [1], which feeds a switched
reluctance drive [2]. The power stage is built up of 9
IGBT modules. Each of them is equipped with a NTC
thermistor. The motivation to use a logarithmic circuit arose
from a wish to implement an easy processing of temperature
data from a multiple number of IGBT modules. An
additional isolation via an optocoupler protects the control
part in case of a blow-up in a power module [3].
The nonlinear behavior of a NTC thermistor has enforced
different approaches to linearize it. The methods vary from
passive resistor bridge circuits [4]-[5], advanced active
circuits [6]-[9] to mathematic curve fitting functions [10].
Each way has its own benefits and shortcomings, so the user
should meet the decision based on the application needs.
From our point of view the idea to use a logarithmic circuit
to overcome the thermistor’s nonlinearity was proposed in
[6] at first and shows a proven implementation. The design
Φ
The authors gratefully acknowledge financial support from Federal
Ministry for Economic Affairs and Energy (BMWi).
Y. Polonskiy is with the Department of Electrical Drives and
Actuators, Universitaet der Bundeswehr, Muenchen, Germany (e-mail:
yevgen.polonskiy@unibw.de).
D. Surmann is with the Department of Aerospace, Universitaet der
Bundeswehr, Muenchen, Germany (e-mail: denis.surmann@unibw.de).
C. Laudensack is with the Department of Electrical Drives and
Actuators, Universität der Bundeswehr, Muenchen, Germany (e-mail:
christian.laudensack@unibw.de).
D. Gerling is Full Professor and Head of the Department of Electrical
Drives and Actuators, Universitaet der Bundeswehr, Muenchen, Germany
(e-mail: dieter.gerling@unibw.de).
PEMC 2014
Fig. 1. Behavior of the thermistor resistance, values from [11]
IV. LOGARITHMIC CIRCUIT
A.
Basic theory
To process (2) easily there should be a possibility to
evaluate ln(RT) directly. This task is accomplished by a
logarithmic amplifier, which basic circuit is shown in Fig. 2.
The functional details can be recalled from [12].
Fig. 2. Basic logarithmic circuit proposed in [12]
This topology is lightly adopted to fit the application as
shown in Fig. 3. The thermistor is supplied by the amplifier
IC1B, configured as a unit buffer. The gates IC1A and IC1D
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16th International Power Electronics and Motion Control Conference and Exposition
Antalya, Turkey 21-24 Sept 2014
Fig. 3. Logarithmic amplifier circuit for the evaluation of the NTC thermistor
work identically to the logarithmic amplifier from Fig. 2
while IC1A converts the thermistor resistance value into a
voltage signal ULOG, the IC1D helps to get rid of the typical
transistor transfer characteristics as current gain ȕ and
reverse saturation current IB0 in the final equation (9). The
mesh equation for the voltage over resistor R6 is
(3)
+U
−U
=0
R6
BE,1
BE,2
For the base emitter voltages the usual equation for the
linear transistor operation is used (4)
U
§ I1 ·
§ I2 ·
¸ ;U
¸ (4)
= U ln¨
= U ln¨
¸ BE,2
BE,1
T ¨β ⋅I
T ¨β ⋅I ¸
©
B0 ¹
©
B0 ¹
Under assumption, that the voltage divider R5, R6 is almost
unloaded because the resistance R6 is much smaller than the
base-emitter resistance, the relation for voltage signal ULOG
is valid:
R +R
5
6 =K U
U
≅U
(5)
LOG
R6
N R6
R
6
with KN = (R5+R6)/R6. Combining (3) and (4) results in
U
§I ·
≅ U ln¨ 2 ¸
(6)
R6
T ¨I ¸
© 1¹
The amplifiers IC1A and IC1D force the collector currents
I1 and I2 to be as following
U
⋅K
R
1
S
2
I =U
⋅
= 15V int
(7)
1
15V int R + R + R R
R
1
2
4
T
T
U
U
I = 15V int = 15V int
(8)
2 R +R
R
8
16
R
with the factor KS = R2/(R1+R2+R4) and the resistor sum
RR = R8+R16. Equations (6), (7) and (8) can be inserted
into (5) yielding in the final solution
U
§ R ·
(9)
≅ U K ln¨ T ¸
LOG
T N ¨K R ¸
© S R¹
From (9) one recognizes the benefit of a logarithmic circuit,
which avoids any steady state dependency on the supply
voltage (here U15Vint). So there is no need for a precise
current or voltage reference. Still there is the temperature
dependent voltage UT, which can be compensated by
choosing the resistor R6 with a temperature coefficient of
+3500ppm/K [12].
U
PEMC 2014
B.
Circuit components
As already mentioned the purpose of the current circuit is
to simplify the temperature monitoring of IGBT modules
under laboratory conditions. Hence the component values in
Fig. 3 are chosen regarding a realistic temperature range
between 20°C and 130°C. In order to simplify the transfer
of ULOG via an analog optocoupler the voltage signal should
be greater or equal to 0V. This requirement constrains the
ratio of RT/(KSRR) for the maximum temperature of 130°C
to
R (@ 130°C )
T
≥1 Ÿ K R ≤ 243Ω
(10)
S R
K R
S R
To simplify the following calculations the lowest
presumable resistance value of RTmin is set to 250Ÿ.
The factor KS influences the current I1, which is supplied to
the thermistor. I1 should be kept low to meet the
requirements on the low consumption, self-heating of the
thermistor and the logarithmic operation range of the
transistor T1. For the minimum resistance value from (10)
the maximum allowable current is set to 0.5mA. With the
nominal supply voltage U15Vint=15V the factor KS results in
0.5mA ⋅ R
T min = 1
(11)
120
S
U
15V int
To keep the steady state current consumption low the
branch R1, R2, R4 is set to 120kȍ common resistance.
From (10) and (11) the resistance RR is determined in (12),
yielding in a sufficient high resistance value.
250Ω
R =
= 30kΩ
(12)
R K
S
Considering (9) and temperatures between 20°C and 130°C
ULOG ranges from 0V to ca. 0.92V.
The capacitors C9 and C10 add a low pass filter for the
supply voltage U15Vint with a cut-off frequency of about
212Hz. The capacitors C3, C4 stabilize the feedback path of
the amplifier gates IC1A and IC1D. The value of 100nF is
chosen intuitively; the current application does not require
any high frequency bandwidth.
The transistor pair T1 should be a matched one with high
current gain ȕ. Unfortunately, there is only a small amount
of suitable parts, which are available from common
distributors. In case costs are not a key factor SSM2212
[13] is a good match. For the current application PMP4201
[14] is chosen, offering a compromise between costs and
precision.
As already stated the resistor R6 should compensate the
K
=
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16th International Power Electronics and Motion Control Conference and Exposition
temperature dependency of UT. Here it is also difficult to
find suitable parts, which are available from common
distributors. Applicable parts can be sourced from
Panasonic or KOA. For the current application Panasonic’s
ERAS33 type [15] is chosen. It possesses a temperature
coefficient of +3300ppm/K. It should be noticed from (9),
that effective temperature compensation requires a high
ratio of R5/R6.
Obviously one of the important parts is the operational
amplifier IC1. During the selection a low current
consumption, offset voltage and offset bias currents are
considered. For the final design ADA4096-4 [16] is opted.
A good alternative is MAX44245 [17].
Other circuit components fulfill the supplementary
functions. The diode D1 protects T1 from reversed baseemitter voltage; resistors R17, R18 compensate bias
currents. The resistor R7 carries the sum of I1+I2 and is
usually set to a value, which allows the amplifier to stay
within the supply voltage range.
V.
ISOLATED SIGNAL TRANSFER
A.
Basic theory
The voltage signal ULOG is transferred to the control part
via the analog optocoupler IC2 as shown on the top in Fig.
4. Thereafter, the signal is amplified by IC3A to make better
use from the A/D-converter input range, see bottom side
Fig. 4.
Fig. 4. Isolated signal transfer
The depicted circuit is a rather simple “photovoltaic”
operation mode [18], which is used for a precise signal
transfer. A positive voltage ULOG forces the amplifier’s
IC1C output to become negative. This leads to a current
flow through the transmitter (LED) diode of IC2 (pins 1,2).
Accordingly the “servo”-photodiode (pins 3,4) becomes
active and drives the current IS. The overall process is
balanced in case IC1C provides the virtual ground for the
voltage ULOG. On the secondary side the receiver diode of
IC2 (pins 5,6) becomes active too and forces a current flow
IR. The current in the “servo”- and the receiver photodiode
are matched by a common named transfer gain K3, which is
given in the optocoupler datasheet.
I
K = R
(13)
3 I
S
The current IR passes the T-network of the amplifier IC3A
giving the output voltage UO. The overall transfer ratio of
the voltage ULOG to UO is
PEMC 2014
U
O
U
=K
Antalya, Turkey 21-24 Sept 2014
R11 ⋅ R15
R3
=K K
3 A
R9
R11 + R15 +
3
(14)
LOG
with the gain KA = (R11+R15+R11ÂR15/R3)/R9. Combining
(9) and (14) results in
§ R ·
(15)
≅ U K K K ln¨ T ¸
O
T 3 A N ¨K R ¸
© S R¹
Finally (15) has to be inserted into (2). First two
“correction” factors C0 and C1 are introduced. Assume that
U
R ·
1 §¨
(16)
C
ln C T ¸ = U
1B ¨ 0 R ¸
O
©
0¹
By a simple comparison of coefficients with (15) it is
obvious that C1 = BUTK3KAKN and C0 = R0/(KSRR). Now
logarithmic rules can be used and (2) is expressed as
C
1
1
(17)
T=
=
R
C
C
R
C
§
·
§
·
1
1 ¨ T¸
1 + 1 ln¨ C T ¸ − 1 ln C
+ ln
0
T
B ¨R ¸ T
B ¨ 0 R ¸ B
© 0¹
©
0
0
0¹
It can be easily recognized that the second denominator
term in (17) equals (16). The final expression for the
temperature is given by (18)
C
1
(18)
T=
C
C
1 − 1 ln C + U
0
O
T
B
0
Due to the logarithmic circuit the temperature calculation in
Kelvin is simplified to one sum and one division operation.
For a result in Celsius an additional subtraction is needed.
The form (18) can be processed more quickly, than e.g.
look-up tables or the approximated by Taylor series
logarithm.
The dual diode D3 and the resistor R24 have an optional
functionality to provide a hardware temperature threshold
monitoring. If ULOG reaches a level ” 0V the output of IC1C
swings to the positive supply rail. The right diode D3
protects the transmitter diode of IC2 from a reverse voltage.
The left diode D3 together with R24 and R20 forms a
voltage divider and forces the positive input of IC1C to
some positive voltage level. Now IC1C “locks” itself and
works as a comparator until ULOG tops the voltage at the
positive input. By connecting the output of IC1C with e.g.
pre-biased open-collector transistor a trigger for the error
input of the IGBT driver [19] can be realized.
Regarding temperature compensation of UT it should be
noted from (15), that a better possibility is to use R9 with
the suitable temperature coefficient rather than R6.
Unfortunately, there is no part available to fulfill this.
B. Circuit components
The resistor R9 is set to 150kŸ to insure a low current
consumption. The transfer gain KA is set to 10 to resolve the
low voltage signal of ULOG properly. The simplest way to
amplify ULOG would be to put a feedback resistor over
IC3A. Thus the required value should be 1.5MŸ to achieve
the desired gain. Such a high value could impose problems
with leakage currents. Therefore the decision to use a low
impedance T-network was made. On the other side the
voltage offset and noise are amplified with the gain of
1+R15/R3, while a simple feedback resistor does not
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16th International Power Electronics and Motion Control Conference and Exposition
contribute such drawbacks; the reader should consider [20]
for a discussion of pros and contras of the different
feedback circuits. To achieve KA = 10 using practical
resistance values R11 and R15 are set to 56kŸ while R3
measures 2.26kŸ.
The resistor divider R24, R20 determines the threshold
where IC1C recovers to the normal operation. During usual
operation R20 compensates the input bias current and its
value is set to 100kŸ, being closely to R9. Choosing R24 =
2Mȍ results in a recovery threshold of approx. 0.72V, or
expressed in temperature the recovery happens when the
thermistor cools down below 43°C, Fig. 13. The capacitors
C5, C8 stabilize the feedback path of the amplifiers and are
set intuitively to 100nF. R10 decouples the output of IC1C
from the low dynamic resistance of the transmitter diode.
R12 is used for the compensation of the bias currents. The
requirements for the op-amp IC3A are moderate compared
with IC1, here TL062AC [21] as a general purpose
amplifier is chosen.
VI. MEASUREMENTS
The circuit discussed above is built up with the possibility
to test two optocouplers – LOC117 [22] and HCNR200
[23], Fig. 5. The optocouplers can be connected alternately
via simple switches. Additionally, the circuit is equipped
with a sub-D connector to route important test points to the
acquisition board NI-6221 [24]. The electronics is tested at
different ambient temperatures TA to verify its functionality.
The circuit is supplied with a bipolar voltage of +15V/-8V.
The thermistor resistance is emulated by a potentiometer.
Antalya, Turkey 21-24 Sept 2014
Fig. 6. Output voltage ULOG of the logarithmic amplifier
ULOG shows a negligible dependency on the ambient
temperature. The relation between the ideal and measured
curves is showed exemplary on the profile at TA = 50°C.
Beside the high deviation nearby 0V (due to offset errors)
the ratio stays almost constant at a mean value of 1.09. This
facilitates a calibration capability as discussed in section
VII. The next Fig. 7 and 8 show curves of the gain KA at
different ambient temperatures. With lower resistance
values the voltage ULOG tends to 0V and offset errors distort
the gain factor.
Fig. 7. Amplifier gain KA with HCNR200
Fig. 5. PCB with the implemented circuit
The reason to implement two optocouplers is to test the
transfer behavior at low voltage levels of ULOG. For example
[23] shows, that with lower current IS the required LED
current increases rapidly. This effect could lead to an
unwanted higher current consumption. In contrast [22]
states, that so called servo gain increases at low LED
currents. This could counteract a higher current
consumption.
First the behavior of the voltage ULOG is tested at different
ambient temperatures TA and compared with the ideal curve
calculated from (9), Fig. 6.
PEMC 2014
Fig. 8. Amplifier gain KA with LOC117
The transfer gain of LOC117 tends to be slightly lower than
of HCNR200. Together with a too high voltage ULOG, Fig.
6, the overall error with LOC117 appears lower than with
HCNR200, referred to Fig. 9. The thermistor (real)
temperature is calculated from (2), while the evaluated
temperature comes from (18).
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16th International Power Electronics and Motion Control Conference and Exposition
Antalya, Turkey 21-24 Sept 2014
Fig. 12. Current consumption with LOC117
Fig. 9. Error between the thermistor (real) and evaluated temperature
The maximum error is about 11°C, which is measured with
HCNR200 at 85°C ambient temperature. During the usual
operation under laboratory conditions the temperature
inside the housing of the power electronics could reach
about 50°C to 70°C. For this condition the maximum error
with HCNR200 is 4°C to 8°C accordingly. For LOC117 the
maximum error reaches 2°C to 5°C. This error is acceptable
for a usual monitoring of power modules. The next Fig. 10
gives an example for the temperatures evaluated from both
ULOG (9) and UO (18).
Finally, the function of the threshold triggering as discussed
in section V.A is depicted in the Fig. 13.
Fig. 13. Threshold triggering
VII.
CALIBRATION POSSIBILITY
In case the application requires a precise temperature
monitoring the transfer function can be calibrated by
operating the circuit at the expected ambient temperature
and adjusting the coefficients to meet the real transfer
characteristics. The equation (18) can be inversed giving a
simple linear function in form f(x) = a + bx
Fig. 10. Calculated temperatures from ULOG and UO, HCNR200, TA=85°C
The overall current consumption of the circuit (thermistor
side) is depicted below, Fig. 11 and 12. The initial concerns
about a possible excessive current consumption at low
voltage levels ULOG (thus at high temperatures) proved to be
unfounded. The use of LOC117 exhibits a lower current
consumption due to a higher servo gain compared with
HCNR200.
1 §¨ 1 ln C0 ·¸ 1
(19)
=
−
+
U
T ¨T
B ¸ C1 O
© 0
¹
Regarding the chosen component values the coefficients of
the function (19) can be easily calculated. The straight line
given by (19) is an ideal transfer function, which is naturally
distorted, Fig. 14. The user emulates known resistance
values and measures the output voltage UO. Thus, the real
curve and its linearized function can be obtained.
Fig. 14. Ideal and measured temperature curves
Fig. 11. Current consumption with HCNR200
The coefficients gained from the measurement in Fig. 14 are
inserted into (19) and the temperature is calculated with the
PEMC 2014
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16th International Power Electronics and Motion Control Conference and Exposition
corrected factors. The measurement accuracy can be
improved gratefully as shown in Fig. 15.
[9]
[10]
[11]
[12]
[13]
[14]
[15]
[16]
[17]
Fig. 15. Influence of the calibration on the measurement error
Keep in mind that the calibration routine is valid only for a
specific ambient temperature. It is possible to implement an
additional ambient temperature sensor; therefore the control
part can adjust the coefficients according to the stored data.
VIII.
CONCLUSION
The paper presented a possibility to obtain the temperature
of a NTC thermistor as an analog voltage signal and transfer
it via an optocoupler to insure galvanic isolation. The
presented method uses a logarithmic amplifier circuit,
which converts the logarithmic temperature dependency in a
linear one. The theoretical background and design are
verified by measurements. The transfer ratio of the circuit
can be calibrated to improve the accuracy at a given
ambient temperature.
IX. REFERENCES
[1]
[2]
[3]
[4]
[5]
[6]
[7]
[8]
M. Hiller and R. Marquardt, "A new converter concept for switched
reluctance drives with multiple energy sources," IEEE Power
Electronics and Drive Systems PEDS 2003, vol. 2, pp. 1235-1240,
Nov. 2003.
C. Laudensack, Q. Yu and D. Gerling, "Dynamic design tool for
canned switched reluctance machines," International Aegean
Conference on Electric Machines and Power Electronics &
Electromotion Joint Conference (ACEMP), Istanbul, Turkey, pp.
775-780, Sept. 2011.
Eupec, "Application Note: Using integrated NTC with reliable
isolation", 2001.
Available: www.igbt.cn/UserFiles/Support_IGBT/file_071.pdf
H. J. Hoge, "Comparison of circuits for linearizing the temperature
indications of thermistors," Review of Scientific Instruments, vol.
50, issue 3, pp. 316-320, May 1979.
Microchip, "Application Note AN685: Thermistors in single supply
temperature sensing circuits", 1999.
Available:
http://ww1.microchip.com/downloads/en/AppNotes/00685b.pdf
R. K. Chakravarty, K. Slater and C. W. Fischer, "Linearization of
thermistor resistance-temperature characteristics using active
circuitry," Review of Scientific Instruments, vol. 48, issue 12, pp.
1645-1649, Dec. 1977.
A. A. Khan and R. Sengupta, "A linear temperature/voltage
converter using thermistor in logarithmic network," IEEE
Transactions on Instrumentation and Measurement, vol. 33, issue 1,
pp. 2-4, March 1984.
R. N. Sengupta, "A widely linear temperature to frequency converter
using a thermistor in a pulse generator," IEEE Transactions on
Instrumentation and Measurement, vol. 37, issue 1, pp. 62-65,
March 1988.
PEMC 2014
[18]
[19]
[20]
[21]
[22]
[23]
[24]
Antalya, Turkey 21-24 Sept 2014
A. R. Sarkar, D. Dey and S. Munshi, "Linearization of NTC
thermistor characteristics using op-amp based inverting amplifier,"
IEEE Sensors Journal, pp. 4621-4626, Dec. 2013.
Y. Cong, Z. Wang-chao, S. Bin and Z. Hang-xia, "Study on NTC
thermistor characteristics curve fitting methods," International
Conference on Computer Science and Network Technology
(ICCSNT), Harbin, China, vol. 4, pp. 2209-2213, Dec. 2011.
SEMiX553GAR128Ds: IGBT module, datasheet, Semikron.
Texas Instruments, "Application Note AN-311: Theory and
Applications of Logarithmic Amplifiers", revised 2013.
Available: www.ti.com/lit/pdf/snoa575
SSM2212: dual NPN matched transistor pair, datasheet, Analog
Devices.
PMP4201: dual NPN matched transistor pair, datasheet, NXP
Semiconductors.
ERAxxx: metal film thermosensitive chip resistors, datasheet,
Panasonic.
ADA4096: micropower, rail-to-rail input/output amplifier, datasheet,
Analog Devices.
MAX44245: precision, low power amplifier, datasheet, Maxim
Integrated.
IXYS Integrated Circuits Division, "Application Note AN-107: LOC
Series Linear Optocouplers", 2013.
Available:
http://www.ixysic.com/home/pdfs.nsf/www/AN107.pdf/$file/AN-107.pdf
Skyper 32Pro R: IGBT driver core, datasheet, Semikron.
Burr Brown (currently Texas Instruments), "Application Note
SBOA061: Designing photodiode amplifier circuits with OPA128",
1994.
Available: www.ti.com/lit/an/sboa061/sboa061.pdf
TL062: low power, JFET-input operational amplifier, datasheet,
Texas Instruments.
LOC117: linear optocoupler, datasheet, IXYS Integrated Circuits
Division.
HCNR200: linear optocoupler, datasheet, AVAGO Technologies.
NI USB-6221: M series multifunction DAQ for USB, datasheet,
National Instruments.
X.
BIOGRAPHIES
Yevgen Polonskiy was born in Dnepropetrowsk, Ukraine on March 24,
1982. He graduated from the technical high school in Konstanz in 2002
and started studying Electrical Engineering at the University of Applied
Sciences, Konstanz. He completed with M.Eng. in 2008 and worked from
2009 to 2011 as development engineer at Maccon GmbH, Munich. Since
August 2011 he is with Institute of Electrical Drives and Actuators at the
Universitaet der Bundeswehr Muenchen, Neubiberg. His research interests
are control of switched reluctance drives and power electronics.
Denis Surmann was born in Hoexter on August 15, 1989. He graduated
from high school in Bad Driburg in 2009 and was awarded by the German
Physics Society. Thereafter, he joined the German Air Force and began his
officer’s career. Surmann is currently a graduate student in the aerospace
program at the Universitaet der Bundeswehr Muenchen, Neubiberg and
received his B.Sc. in Aerospace Engineering in 2013. His bachelor thesis
focused on a switched reluctance drive control.
Christian Laudensack was born in Germany, 1981. He graduated in
2007 in University of Federal Defense, majoring in aerospace engineering.
His employment experience included the German Federal Armed Force,
the IAB GmbH, Ottobrunn, the Systemzentrum für Luftfahrzeugtechnik,
Erding, and the Institute of Electrical Drives and Actuators, Universitaet
der Bundeswehr Muenchen, Neubiberg. His special fields of interest
include the design analyses of switched reluctance drives.
Dieter Gerling, born in 1961, got his diploma and Ph.D. degrees in
Electrical Engineering from the Technical University of Aachen, Germany
in 1986 and 1992, respectively. From 1986 to 1999, he was with Philips
Research Laboratories in Aachen, Germany as Research Scientist and later
as Senior Scientist. In 1999, Dr. Gerling joined Robert Bosch GmbH in
Bühl, Germany as Director. Since 2001, he is Full Professor and Head of
the Institute of Electrical Drives and Actuators at the Universitaet der
Bundeswehr Muenchen, Neubiberg, Germany.
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