Utilizing 650 V MOSFETs
W
hi
Version 1, Mai 2012
With fast Body Diodes for Motor Control
te
pa
p
Prof. Dr.-Ing. Jens Onno Krah (FH Köln), Andreas Rath (FH Köln), Markus Höltgen (FH Köln), Rolf Richter (EBV)
er
Efficient energy conversion is more and more important. Latest power MOSFET technology with fast body diodes offers new opportunities for motor control. These MOSFETs combine a high blocking voltage of 650V with low Rdson with improved body diode
ruggedness during reverse recovery in hard switching applications. Using moderate switching frequencies, excellent efficiencies
in the power range between 200 W and 2 kW are possible. FPGA based control allows to achieve high bandwidth control even at
these moderate switching frequencies.
1. Introduction
Power MOSFETs are well known in low voltage (< 200 V) motor control applications. Key reason is the low RDSon of the channel
which also reduces the conduction losses of the freewheeling current of the body diode if the MOSFET is switched on. This active
bypassing of the freewheeling diode is also used in efficient buck converters and synchronous rectification.
2. Estimation of the losses
The following calculations are based on a hard switched single phase inverter leg
driving inductive load, fig. 1. The output voltage depends on the switching frequency combined with the duty cycle. The discussed device is an ST FDmesh™ V
power MOSFET with 43 mΩ RDSon [1].
Conducting losses
The conducting losses are at low current mainly i² * RDSon. The duty cycle determines in which transistor (high side or low side) the losses will appear.
Pcond = i² * RDSon
(1)
Fig.1: Inverter leg (half bridge) using 650 V
MOSFETs with fast body diodes
As long as the current is lower than approximately 10 A the low voltage drop (< 0.7 V) ensures that the current is flowing through
the MOS channel and that the body diode will take overthe current only during the dead time, which is significantly less than
0.5 μs. At higher current values the conducting losses are shared between MOS channel and body diode.
Switching losses
The switching losses depend on the dc-link voltage vDC, the phase current iPh, the rising and falling times
tr + tf = tsw and the switching frequency fs.
Psw ≈ fs * tsw * vDC * iPh
(2)
PSW ≈ 5 kHz * 50 ns * 300 V * 1 A = 75 mW
Switching losses are insignificant at moderate switching frequencies.
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All statements are without any engagement. Subject to modifications and amendments.
Utilizing a power MOSFET with an RDSon of 43 mΩ (25 °C), a 1 A current will result in 43 mW conducting losses. At operating
temperature (100 °C) the conducting losses will be approximately 70 mW due to the positive temperature coefficient of RDSon (»70 mΩ).
Utilizing 650 V MOSFETs
With fast Body Diodes for Motor Control
W
hi
Version 1, Mai 2012
te
pa
p
Prof. Dr.-Ing. Jens Onno Krah (FH Köln), Andreas Rath (FH Köln), Markus Höltgen (FH Köln), Rolf Richter (EBV)
er
Reverse Recovery Losses
The dominating losses at low current are the reverse recovery losses Prr of the intrinsic body diode. At rated current these losses are
proportional to the switching frequency, the dc-link voltage and the reverse recovery charge Qrr of the device:
Prr = fs * vDC * Qrr (Qrr is specified in the data sheet at IPh = 65 A)
(3)
Prr = 5 kHz * 300 V * 2.1 µC = 3.15 W
Datasheet values are measured with a defined di/dt in A/μs, the diode is exposed to the rated drain current and the conduction
time before turn off is in the range of up to several hundred μs and more. This leads to a maximum possible Qrr. Furthermore
the measurement method of the JEDEC standard includes besides the Qrr also some part of the output charge of the MOSFET,
which results in a very large value not representing reality. The main influence on Qrr comes from the magnitude of the current and the flooding time of the diode. The longer the time that the body diode conducts current before turn off, the higher
the reverse recovery charge will be. Therefore the Qrr values relevant for the application are much lower, due to currents that
do not exceed half the maximum drain current and a diode flooding time of only 50 ns to 150 ns. Minimizing the dead time is
consequently an important topic [2].
The function Qrr(i) is not linear with respect to the current iPh and there is no well-known model to estimate this relation. According
to laboratory measurements the following calculations are based on a simplified model to estimate the relation between Qrr and
iPh at 100 °C.
iPh
(4)
Q (i )≈Q Data sheet * iData sheet rr
Ph
rr
It is self-evident that power MOSFETs with no fast body diode cannot be used in such an application. It is also to be noted that 5
kHz is not a really highswitching frequency. This lower switching frequency is acceptable in case of medium dynamic motor control
applications where the inductance to smooth the current is part of the motor and not increasing cost.
Gate Driving Losses
At moderate switching frequencies latest power MOSFETs
do not need a lot of gate driving power.
(5)
Pg = 185 nC * 5 kHz * 15 V = 13.875 mW
This is not much compared with the standard gate driver quiescent supply power of approximately 15 V * 5 mA = 75 mW.
Using the described equations the losses are mainly a function
of the current. At low current, reverse recovery losses Prr are
dominating. At higher currents (> 5 A) the conducting losses
Pcond. become a more important issue, fig.2. As a reference the
losses of a comparable IGBT are shown. There is no shielded
motor cable considered. The significantly higher conducting
losses of the IGBTs are there reason that the MOSFET power
stage produces in average only 30% of the losses.
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Fig. 2: Losses of a MOSFET inverter leg (single phase) is mainly a function of the
phase current iPh. Switching frequency fs is set to 5 kHz. As a reference the losses of
a comparable IGBT are shown. Gate driver, control and cable related losses are not
considered.
All statements are without any engagement. Subject to modifications and amendments.
Pg = Qg * fs * vgate
Utilizing 650 V MOSFETs
With fast Body Diodes for Motor Control
W
hi
Version 1, Mai 2012
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pa
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Prof. Dr.-Ing. Jens Onno Krah (FH Köln), Andreas Rath (FH Köln), Markus Höltgen (FH Köln), Rolf Richter (EBV)
er
Current Ripple
In applications with considerable low current periods a significant current ripple can help to reduce the reverse recovery losses
even more. Under these circumstances the current never floods the body diode. The current ripple is inversely proportional to
the switching frequency fs. By decreasing the switching frequency the range of current values without reverse recovery losses
can be expanded.
Motor Cable
Especially in the power range up to 2 kW switching losses increase significantly if the motor is connected via a shielded cable
that is several meters long. Using a 10 m cable with 50 pF/m the cable capacity results in Ccable = 500 pF.
Psw cable = u² * Ccable * fs (6)
Psw cable = (300 V)² * 500 pF * 5 kHz = 225 mW
Using 16 kHz switching frequency, 20 m cable and 600 Vdc results in reasonable switching losses in the power electronics due to
the cable:
Psw cable = (600 V)² * 1000 pF * 16 kHz = 5,76 W
This single phase example illustrates that long cables in combination with high switching frequencies and high voltages are
preventing desired efficiencies.
3. Inverter Efficiency
Evaluating the losses in relation to the output power for an
inverter efficiency calculation leads to very interesting results.
Especially at low currents the efficiency of the power MOSFET
inverter leg compared with an IGBT half bridge is significantly better, fig 3. There is again no shielded motor cable
considered.
Using 300 V dc-link voltage a 3 A current equals up to 300
W single phase power or roughly 1 kW in a 3-phase system.
Below 200 W a 48 V dc-link design can help to reduce size and
cost due to less isolation requirements. Above 2 kW IGBT
solutions using 600 V dc-link voltage should be considered.
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Fig. 3: Efficiencies of a MOSFET and an IGBT inverter leg (single phase) are a function of the phase current iPh. Switching frequency fs is set to 5 kHz. Output voltage is
110 Vac. Gate driver, control and cable related losses are not considered.
All statements are without any engagement. Subject to modifications and amendments.
η = Pout / (Ptot. + Pout)
Utilizing 650 V MOSFETs
With fast Body Diodes for Motor Control
W
hi
Version 1, Mai 2012
te
pa
p
Prof. Dr.-Ing. Jens Onno Krah (FH Köln), Andreas Rath (FH Köln), Markus Höltgen (FH Köln), Rolf Richter (EBV)
er
4. Current Loop Bandwidth
Due to parallel processing inside a Field Programmable Gate Array (FPGA), the control algorithm computing time can be significantly less than 1 μs. Together with advanced control technologies in combination with a current observer the bandwidth of fast
switching power stages is not limited by the delay time of high precision (EMI filtered) current measurement any longer. Using
that technology, high control bandwidth in conjunction with high precision current control is now possible at no trade off. The
control strategy relies on a simplified machine model without incurring performance degradations. At 5 kHz switching frequency
more than 2 kHz current loop bandwidth is achievable. This is twice of today’s industrial standard [3].
5. Conclusion
Utilizing latest 650 V power MOSFET technology to build hard switching inverters in the power range from 200 W to 2 kW results
in very high efficiencies at moderate switching frequencies. MOSFET power stages are producing in average only 30% of the
losses of an equivalent IGBT design.
[1] Preliminary data sheet: FDmesh V Power MOSFET, STW77N65DM5, 43 mΩ device, ST, 2012.
[2] “
Improving Efficiency of Synchronous Rectification by Analysis of the MOSFET Power Loss Mechanism”, Infineon Application
Note, Rev. 2.0, June 2009.
[3] C
h. Klarenbach, J.O. Krah: “Fast and High Precision Motor Control for High Performance Servo Drives”, PCIM Power Conversion Intelligent Motion, Nürnberg, May 2010, pp. 326-333.
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All statements are without any engagement. Subject to modifications and amendments.
The efficiency will be reduced by using long motor cables due to the cable capacity. To achieve servo like current loop bandwidth
an FPGA with its rapid parallel processing can be used. Drawback of MOSFETs is their poor overload capability compared with
the common IGBT power stages. An up to 10 μs short circuits withstand capability is not specified, but the MOSFET intrinsic current limiting will allow a short circuit protection together with fast FPGA control systems.