IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 17, NO. 1, JANUARY 2002
101
Design of a Novel ZVT Soft-Switching Chopper
Huijie Yu, Member, IEEE, Byeong-Mun Song, Member, IEEE, and Jih-Sheng (Jason) Lai, Senior Member, IEEE
Abstract—This paper presents a simple soft-switching chopper
scheme with fixed timing control. A near-zero-voltage transition
(near-ZVT) switching condition is realized by adding one auxiliary resonant snubber branch to a full-bridge two-quadrant
chopper. The turn-off loss is reduced by lossless snubber capacitors in parallel with the main switches. The proposed design
approach is to realize true ZVT at the rated load and near-ZVT
under all other load current conditions. Computer simulation
and hardware experiments have been implemented to verify
the proposed concept, and the resulting voltage and current
waveforms are shown from 30% to 150% load conditions in this
paper. Under near-ZVT switching, the switching loss and
of the power device can be significantly reduced, and the reverse
recovery problem of main switches can be avoided as compared
to the hard-switching case. The design criteria for the resonant
components are described with a practical example that has been
used in a commercial magnetic levitation system. Under different
load current conditions, the proposed design methodology has
been fully verified with simulation and experimental results.
Index Terms—Chopper, resonant, soft-switching, snubber, zero
voltage transition.
I. INTRODUCTION
S
OFT switching techniques, especially the zero-voltage
transition (ZVT) have become more and more popular
in the power electronics industry. The goal of soft switching
is to achieve less switching losses and noises than that of the
traditional hard switching. There are numerous examples of
soft-switching techniques [1]–[4]. Some are quite promising
in some particular applications, but may not be suitable for
inverter applications. One major issue is to adapt to load current
change and to achieve ZVT conditions at all load conditions.
Variable timing control needs to be used to make soft switching
more efficient than hard switching, especially for insulated-gate
bipolar transistor (IGBT) based inverter applications. However,
the control of variable timing is very complicated, and the
implementation requires numerous iterations and tunings.
Fixed timing is preferred to variable timing for control simplification, but how to achieve ZVT with fixed timing under
all load condition is quite challenging. The objective of this
paper is to present the design criteria of a novel fixed-timing
soft-switching chopper using the auxiliary resonant snubber
to replace a conventional RCD-snubber based hard-switching
Manuscript received July 20, 1999; revised January 15, 2000. Recommended
by Associate Editor J. H. R. Enslin.
H. Yu and J.-S. Lai are with the Center for Power Electronics Systems,
The Bradley Department of Electrical and Computer Engineering, Virginia
Polytechnic Institute and State University, Blacksburg, VA 24061-0111 USA
(e-mail: huyu@vt.edu).
B.-M. Song is with the Center for Power Electronics Systems, The Bradley
Department of Electrical and Computer Engineering, Virginia Polytechnic Institute and State University, Blacksburg, VA 24061-0111 USA and also with the
General Atomics, San Diego, CA 92186-9784 USA (e-mail: songb@gat.com).
Publisher Item Identifier S 0885-8993(02)02165-8.
chopper. The tested chopper is being used in a magnetic driver
for a commercial magnetic levitation (MAGLEV) system.
and loss reduction, turn-on
The new design achieves
current spike and noise reduction, and finally the improvement
of efficiency and associated heat sink size reduction. Because
the load current of the chopper is only in one direction, only
one auxiliary branch is needed and the control is relatively
simple. The basic cell topology is similar to that of a family of
auxiliary resonant snubber presented in [5]. Main features of
the proposed circuit are
1) main switch turns on at near zero-voltage condition;
2) auxiliary switch turns on and off under zero-current condition;
3) main switch turn-off loss is reduced by lossless snubber
capacitors;
4) near zero-voltage turn-on condition is created by a current injection from the auxiliary branch that contains a
resonant inductor, an auxiliary switch, and snubber capacitors.
These features are similar to those of the resonant snubber inverters described in [5], except that the main switch operation is
near ZVT in the proposed circuit because its initial inductor current is not controllable, and the capacitor voltage at the end of
resonance cannot reach true zero due to the effect of the dissipative components in the resonant branch. The loss due to
near-ZVT is related to the design methodology.
This paper emphasizes the design methodology that allows
a two-quadrant chopper to have an efficient near-ZVT operation for a wide-range load conditions. Using the proposed design criteria, a 10-kW hard-switched commercial chopper that
was used in a magnetic levitation (MAGLEV) system was retrofitted with the proposed resonant circuit and fully executed
with conceptual design, computer simulation, and hardware implementation for near-ZVT operation. Simulation and experimental results prove that the proposed design methodology is
effective, and the proposed soft-switching circuit is well suited
for two-quadrant chopper applications.
II. OPERATION PRINCIPLE
Fig. 1 shows the proposed soft-switching chopper circuit
with auxiliary resonant snubber for a two-quadrant chopper.
The chopper bridge consists of two synchronously switching
- . The diodes provide a
pairs, switches - and diodes
freewheeling current path and a reverse voltage across the load
for two-quadrant operation. The lossless snubber capacitors are
added across main devices, and the auxiliary branch is added
in between two phase-legs. The auxiliary branch consists of
one auxiliary switch, one fast reverse recovery diode, and one
resonant inductor. Since the load current flows in uni-direction,
only one auxiliary branch is needed to achieve soft switching.
0885–8993/02$17.00 © 2002 IEEE
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 17, NO. 1, JANUARY 2002
Fig. 1. Proposed soft-switching chopper circuit.
Fig. 2 illustrates the operation modes for the proposed softswitching scheme. The basic control is to turn on the auxiliary
, before turning on the main switch, and . The
switch,
auxiliary branch takes over the current from the freewheeling
diode and resonates with capacitors in parallel with the main
switch. The main switch is turned on while the voltage across
the main switch drops nearly to zero after resonance. Although
only near-zero-voltage switching can be achieved with the proposed fixed timing control, the diode reverse recovery and
problems of the chopper circuit are effectively solved. The circuit operation modes are described in Fig. 3(a)–(f).
Initially at time , all switches are off, and the load current is
and
as shown in Fig. 3(a). Operation
freewheeling through
modes for a complete cycle are described in detail as follows.
Fig. 2.
Key waveforms of the proposed scheme.
itor voltage drops to zero without proper sensing,
the main switch can thus be turned on at a nearzero-voltage condition. This near zero-voltage is
created by the auxiliary resonant circuit for a
short period, which can be considered as “near
zero-voltage transition” or near-ZVT. After the
main switches turn on, the inductor current decreases linearly due to reverse voltage polarity.
Mode e
Mode a
:
Assume that load current is positive when
and
are conducting the load current, and the
and
are off.
main switches
:
Following the pulse-width-modulation (PWM)
turns on at
command, the auxiliary switch
, the current in
increases linearly and the
and
decreases linearly.
current in diodes
The auxiliary branch diverts the current from the
freewheeling diode gradually.
:
After the resonant current decreases to zero at
, the auxiliary switch gate signal can be turned
off at . The main switches then conduct the
load current, and the auxiliary switch is turned
off under zero-current condition.
:
Main switches turn off with lossless snubber
and
are
capacitors. Once the capacitors
, and
and
are discharged
charged to
to 0, the load current is transferred to diodes
and
, and the circuit operation returns to
Mode a.
Mode b
Mode c
:
After the auxiliary branch current is larger than
and
turn off
the load current at , diodes
naturally. Then all four snubber capacitors resonate with the auxiliary inductor. The capacitor
across the switch discharges with a finite rate to
allow the switch voltage drop to zero.
Mode d
:
At the end of the resonant stage, the snubber capacitors are discharged to zero voltage at . At
this moment, the main switch can be turned on at
zero-voltage condition. In reality, the dissipative
components in the resonant branch may prevent
the voltage from swinging down to true zero but
close enough. However, even if the voltage can
swing to true zero, it is difficult to turn on the
main switch at the exact moment that the capac-
Mode f
III. DESIGN CRITERIA
A. Design Analysis
, the voltage across
At the end of the resonant stage
the main switch should be fully discharged so that the main
switches can be turned on under zero voltage. The key design
point is how to catch the zero-voltage instant and turn on the
main switches exactly at or as close as possible to . The following design analysis will focus on this particular resonant
stage to ensure a proper resonant operation. Once the resonant
stage is well designed, the component value and control timing
can be determined. As long as the resonant inductor current
begins to resonate with the careaches the load current at
pacitors. The equivalent circuit during the resonant period can
be shown in Fig. 4.
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YU et al.: DESIGN OF A NOVEL ZVT SOFT-SWITCHING CHOPPER
103
(a)
(b)
(c)
(d)
(e)
(f)
Fig. 3. Operation stages of ZVT chopper.
2) The duration of the resonant stage is fixed at half of the
natural resonant cycle of resonant tank .
The resonant capacitor voltage and inductor current can be expressed as
(2)
(3)
where
Fig. 4.
Equivalent circuit of resonant stage.
(4)
To simplify the circuit,
is flipped down, and
is flipped
up. The initial condition (IC) of the resonant tank is given in
Fig. 5. Finally, a very simple circuit can be drawn as shown in
and
are the equivalent resonant
Fig. 5(c). In this figure,
capacitor and inductor during the resonant stage, i.e.,
(1)
, we have
.
In the case of
The final equivalent circuit is a very simple LC resonant tank
with zero initial condition. Here we notice two important points.
1) The resonant stage is independent of load current condition.
The current stress on the auxiliary branch can be obtained as
(5)
, is the interval
The auxiliary switch pre-turn-on time,
to , which is the sum of inductor charging time
from
and the resonant stage duration, . A quality factor
is
to , as shown in
defined here as the ratio of
(6)
(7)
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 17, NO. 1, JANUARY 2002
(a)
(b)
(c)
Fig. 5. Simplification of resonant stage circuit.
Note that if
and
can be chosen such that
is sufficiently large with a fixed pre-turn-on time,
, where
is the charging
time under normal load condition, the near-ZVT can then be
is much larger than , even if the main
obtained. Since
switch is turned on a little earlier or later due to the load current
variation, the voltage will only swing back to a finite amplitude,
but close enough to zero-voltage condition.
To reduce the peak resonant current so as to reduce the cirand small .
culating energy, it is desirable to have large
However, for a wide range of near-ZVT operation, it is desirand a small
so that
conable to have a large
dition is satisfied. Since a typical MOS gated device can withstand a high peak over-current in a short period, a larger
and a smaller
may cause a high peak current but not cause a
problem of finding an economical device to handle it. In other
words, a small tank impedance is desirable in the most cases,
and thus the tank impedance becomes an important design
factor. The capacitor value can be selected based on the
requirement and turn-off loss test. The resonant inductor value
can be calculated with the predetermined , and the pre-turn-on
time of the auxiliary switch is optimized at the rated load conequals
under the rated load
dition. That is to let
condition. As a result, the worst case happens under no-load and
heavily overload conditions.
or
.
Fig. 6. Ratio of T to T with respect to normalized impedance Z
B. Design Procedure Example
Fig. 7.
Normalized resonant branch peak current I
.
as a function of Z
If the main switch is turned on precisely with
delay
after the auxiliary switch, and the circuit components are lossless, the exact ZVT condition can be achieved. It should be noted
is load current dependent, it is necesthat according to (6),
sary to adjust the pre-turn-on time of the auxiliary switch to meet
different load current condition if an exact ZVT is desired. To
implement this it is necessary to use variable timing control to
according to the load current condition. However,
change
such a variable timing control requires current sensing and additional complicated control circuitry. It is desirable to look for
a simple solution with fixed-timing control but not losing ZVT.
The proposed approach is described as follows.
In a commercial MAGLEV chopper application, the nominal
is 25 A, and the dc bus voltage
is 300 V.
load current
The design procedure can be described as follows.
Step 1) Decide resonant tank impedance so that the quality
is large enough to satisfy near-ZVT
factor
condition with fixed timing control. However, the
must be limited to avoid
peak resonant current
excessive loss in the auxiliary branch. To facilitate
the comprehensive of the design under different condition, the tank impedance and resonant peak current
is normalized as follows:
(8)
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YU et al.: DESIGN OF A NOVEL ZVT SOFT-SWITCHING CHOPPER
Fig. 8.
105
Turn-off energy as a function of C under different load conditions.
(0.25, 0.4, 0.542, 0.8).
Fig. 9. Variation of T as a function of C and Z
Thus (5) and (7) can be rewritten as
(9)
(10)
As shown in Fig. 6, is chosen as 0.542 which
value of 6.0. In this case,
corresponds to a a
the estimated normalized peak resonant current is
2.845 as indicated in Fig. 7. The selection process
can also start with limiting the peak current first, and
to allow a wide range near-ZVT
check with
condition.
and
so the
requirement can be
Step 2) Select
is proper for acsatisfied, and the resonant cycle
tual implementation. Since the main switch turn-off
loss can only be reduced by snubber capacitors, it
is necessary to perform device test to determine a
proper value for . Fig. 8 shows the test results of
turn-off energy under different snubber capacitance
value so
and load current conditions. Select a
will not significantly
that further increments of
further reduce turn-off loss. In the meantime, it is
necessary to let resonant cycle, , be a reasonable
value so that it is not too small for practical implementation and not too large to avoid loss of duty
cycles. Fig. 9 shows the changes of resonant cycle,
, under different
and values. Based on the
above criteria, a value of 0.1 F was chosen for .
The resonant inductor value is then calculated by
H, and the tank resonant cycle
is around 4 s.
Step 3) Determine the pre-turn-on time of the auxiliary
, and the turn-on duration of the auxilswitch,
.
is the sum of the pre-charging
iary switch,
and resonant period .
is load current
time
dependent and can be chosen under static load
is much larger than
current condition. Since
, the variation of
will not affect much of the
near zero-voltage condition. In this example
Fig. 10.
Simulated key waveforms of near-ZVT chopper scheme.
is chosen as 2.3 s.
is the turn-on duration of
is not critical because
the auxiliary switch.
the auxiliary switch can be turned off after the
should be larger
current reduces to zero. So
, and the selection in this case is
than
s.
Step 4) Summarize the design parameters and select proper
auxiliary switch and passive components. Up to this
point, the major design has been completed. The remaining jobs such as switch selection and magnetic
design can be left to practicing engineers. The complete design summary is listed as follows:
V,
A,
F,
H,
s,
s,
s,
A,
.
IV. SIMULATION RESULTS
The effectiveness of the proposed control scheme can be
simulation with the above design paramproved by
eters. Actual commercial IGBT SPICE models are used, but
parasitic parameters and dissipative components like capacitor
ESR and ESL are not included in the simulation.
Fig. 10 shows the key waveforms of the chopper operation.
, and current,
,
Fig. 10(a) shows the main switch voltage,
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 17, NO. 1, JANUARY 2002
Fig. 11. Resonant current I
under incorrect timing.
(A) and switch voltage V
(a)
(c)
Fig. 12. Waveforms of resonant current I
under different load current conditions.
(V) waveforms
Fig. 13.
Experimental waveforms of the ZVT chopper scheme.
Fig. 14.
Switch voltage waveform under incorrect timing.
(b)
(d)
(A) and switch voltage V
(V)
waveforms. It can be seen that the main switches operate well
is
in near zero-voltage turn-on condition. The turn-on
and the turn-off
controlled by the resonant time constant,
is proportional to the load current but is limited by the
added snubber capacitors. The main switch diode reverse recovery problem is eliminated, and thus there is no current spike
during main switch turn-on. Fig. 10(b) shows that the auxiliary
, is around 70 A, as expected in the
branch peak current,
previous design section. Fig. 10(c) shows the main switch gate
, and auxiliary switch gate signal,
.
drive signal,
It is possible that near zero-voltage turn-on condition may be
lost if the timing is not controlled properly. Fig. 11 indicates that
the main switch is turned on while the switch voltage swings up
to a certain value with the situation that the pre-turn-on time
is longer than the designed value.
Fig. 12(a)–(b) show the simulated waveforms of the voltage
across the main switch and the resonant current during turn-on
process with designed control timing under different load currents: (a) 7.5 A, (b) 17 A, (c) 28 A, and (d) 37 A. It can be seen
that near-ZVT turn-on of the main switch is satisfied for all load
current conditions.
In this simulation, the load current is considered as a constant
current source during the switching period with a value corresponding to the actual experimental load current as described in
the next section for the comparison purpose.
V. EXPERIMENTAL RESULTS
The above-designed soft-switching chopper has been fully
tested with the same parameters that were used in the simulation. Fig. 13 shows experimental key waveforms of load cur, resonant current,
, switch voltage,
, and dc
rent,
input current, . The measurement of dc input current is for the
purpose of loss calculation. To verify loss of ZVT with inappropriate timing, an experiment was conducted under the condition
addressed in Fig. 11. Fig. 14 shows the corresponding test results of losing ZVT condition. As can be seen from Fig. 14 that
an oscillation occurs when the switch turns on after the voltage
has been swung back to a relatively high voltage level.
Fig. 15(a)–(d) shows experimental waveforms of the load cur, the resonant current,
and the voltage across the
rent,
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YU et al.: DESIGN OF A NOVEL ZVT SOFT-SWITCHING CHOPPER
107
TABLE I
COMPARISON OF CALCULATED, SIMULATED, AND EXPERIMENTAL RESULTS
(a)
(b)
(c)
(d)
Fig. 15. Waveforms of resonant current and switch voltage under different
load current condition.
experimental results match well with the designed value in all
different load conditions.
VI. CONCLUSION
Fig. 16.
Loss comparison between hard- and soft-switching choppers.
switch,
, under different load current conditions that are corresponding to Fig. 12(a)–(d) conditions. The timing design is to
ensure that the main switch turns on at near zero voltage under
the nominal operation condition (25 A for the example chopper
case). Waveforms indicate that even at extreme conditions such
as 30% (lightly loaded) in Fig. 15(a) and 150% (overloaded) in
Fig. 15(d), the switching waveform is clean, and the near-ZVT
condition is well satisfied. Loss evaluation results indicated that
the total loss reduction was 31% at the nominal load condition,
as indicated in Fig. 16.
Table I compares the calculated, simulated and experimental
, and its conducresults for resonant branch peak current,
, under different load current,
, condition time,
tions. Although there are some minor differences due to negligence of parasitics and dissipative components, simulation and
In this paper, the design criteria of a novel near-ZVT softswitching chopper are presented with verification of both simulation and experimental results. The resonant tank impedance
was found to be the most critical parameter for ZVT design and
should be selected properly. A step-by-step design procedure
was described with a practical example. The proposed simple
fixed-timing control scheme is proven to be effective to achieve
near-ZVT for a wide range of load conditions. The example
soft-switching chopper also performs significantly better than
its hard-switching counterpart [6] in switching loss and
reduction.
REFERENCES
[1] B. Serge et al., “Chopper and PWM inverter using GTO’S in dual
thyristor operation,” in Proc. EPE Conf., 1987, pp. 383–389.
[2] R. W. DeDoncker and J. P. Lyons, “The auxiliary resonant commutated pole converter,” in Proc. IEEE IAS Ann. Meeting, 1990, pp.
1228–1235.
[3] W. McMurray, “Resonant snubbers with auxiliary switches,” IEEE
Trans. Ind. Applicat., vol. 29, pp. 355–362, Mar./Apr. 1993.
[4] J. S. Lai et al., “A delta configured auxiliary resonant snubber inverter,”
IEEE Trans. Ind. Applicat., vol. 32, pp. 518–523, May/June 1996.
[5] J. S. Lai, “Fundamentals of a new family of auxiliary resonant snubber
inverter,” in Proc. IEEE IECON Conf., Nov. 1997, pp. 645–650.
[6] B. M. Song, J. S. Lai, D. Qu, H. Yu, and H. K. Sung, “A novel softswitching chopper using auxiliary resonant snubbers for a MAGLEV
system,” in Proc. 16th VPEC Sem., Blacksburg, VA, Sept. 1998, pp.
279–284.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 17, NO. 1, JANUARY 2002
Huijie Yu (M’99) was born in Hunan, China in
1972. He received the B.S. and M.S. degrees in
electrical engineering from Tsinghua University,
Beijing, China, in 1994 and 1997, respectively, and
is currently pursuing the Ph.D. degree at the Virginia
Polytechnic Institute and State University (Virginia
Tech), Blacksburg.
In 1997, he joined the Center for Power Electronics
System, Virginia Tech. His research interests include
soft switching power converters, modeling and control of inverter motor drive systems, and high temperature power converters.
Byeong-Mun Song (M’90) received the B.S. and
M.S. degrees in electrical engineering from the
Chungnam National University, Taejon, Korea,
in 1986 and 1988, respectively, and is currently
pursuing the Ph.D. degree at the Virginia Polytechnic
Institute and State University, Blacksburg.
From 1988 to 1994, he was a Senior Research Engineer of the Power Electronics Division, Korea Electrotechnology Research Institute (KERI), Changwon,
Korea. In 1991, he was a Visiting Research Engineer
at the Technical University of Braunschweig, Germany, where he worked for developing the magnetic levitation (Maglev) system.
Since August 2000, he has been with General Atomics, San Diego, CA, as a
Staff Engineer. He is working on advanced Maglev projects and leads for power
electronics systems. His research interests are soft-switching inverters, motor
drives, multilevel converters, and active power filters.
Mr. Song received three awards including an IPEC-2000 Paper Award in
2000, one award from the Korea Ministry of Science and Technology in 1993,
and one award from KERI in 1992.
Jih-Sheng (Jason) Lai (SM’90) received the M.S.
and Ph.D. degrees in electrical engineering from
the University of Tennessee, Knoxville, in 1985 and
1989, respectively.
From 1980 to 1983, he was the Head of the Electrical Engineering Department, Ming-Chi Institute of
Technology, Taipei, Taiwan, R.O.C., where he initiated a power electronics program and received a grant
from his college and a fellowship from the National
Science Council to study abroad. In 1986, he became
a Staff Member at the University of Tennessee, where
he taught control systems and energy conversion courses. In 1989, he joined
the Electric Power Research Institute (EPRI), Power Electronics Applications
Center (PEAC), where he managed EPRI-sponsored power electronics research
projects. In 1993, he worked with the Oak Ridge National Laboratory as the
Power Electronics Lead Scientist, where he initiated a high power electronics
program and developed several novel high power converters including multilevel converters and auxiliary resonant snubber based soft-switching inverters.
Since August 1996, he has been with the Virginia Polytechnic Institute and State
University as an Associate Professor. His main research areas are in high power
electronics converter topologies, motor drives, and utility power electronics interface and application issues. He has published more than 105 technical papers
and two books. He received eight U.S. patents in the area of high power electronics and their applications.
Dr. Lai received several distinctive awards including a Technical Achievement Award in Lockheed Martin Award Night, two IEEE IAS Conference Paper
Awards from Industrial Power Converter Committee, one IEEE IECON Best
Paper Award, and an Advanced Technology Award from Inventors Clubs of
America. He is a member of Phi Kappa Phi and Eta Kappa Nu. He is the Chair
of the IEEE Power Electronics Society Standards Committee. He chaired a
Technical Committee for the 2001 DOE Future Energy Challenge and IEEE
COMPEL-2000 Workshop.
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