CNY70
Vishay Semiconductors
Reflective Optical Sensor with Transistor Output
Marking area
FEATURES
• Package type: leaded
• Detector type: phototransistor
• Dimensions (L x W x H in mm): 7 x 7 x 6
E
21835
D
Top view
19158_1
DESCRIPTION
The CNY70 is a reflective sensor that includes an infrared
emitter and phototransistor in a leaded package which
blocks visible light.
• Peak operating distance: < 0.5 mm
• Operating range within > 20 % relative collector
current: 0 mm to 5 mm
• Typical output current under test: IC = 1 mA
• Emitter wavelength: 950 nm
• Daylight blocking filter
• Lead (Pb)-free soldering released
• Compliant to RoHS directive 2002/95/EC
accordance to WEEE 2002/96/EC
and
in
APPLICATIONS
• Optoelectronic scanning and switching devices i.e., index
sensing, coded disk scanning etc. (optoelectronic encoder
assemblies).
PRODUCT SUMMARY
PART NUMBER
DISTANCE FOR
MAXIMUM CTRrel (1)
(mm)
DISTANCE RANGE FOR
RELATIVE Iout > 20 %
(mm)
TYPICAL OUTPUT
CURRENT UNDER TEST (2)
(mA)
DAYLIGHT
BLOCKING FILTER
INTEGRATED
0
0 to 5
1
Yes
CNY70
Notes
CTR: current transfere ratio, Iout/Iin
(2) Conditions like in table basic charactristics/sensors
(1)
ORDERING INFORMATION
ORDERING CODE
CNY70
PACKAGING
VOLUME (1)
REMARKS
Tube
MOQ: 4000 pcs, 80 pcs/tube
-
Note
MOQ: minimum order quantity
(1)
ABSOLUTE MAXIMUM RATINGS
PARAMETER
(1)
TEST CONDITION
SYMBOL
VALUE
UNIT
mW
COUPLER
Tamb ≤ 25 °C
Ptot
200
Ambient temperature range
Tamb
- 40 to + 85
°C
Storage temperature range
Tstg
- 40 to + 100
°C
Tsd
260
°C
Total power dissipation
Soldering temperature
Distance to case 2 mm, t ≤ 5 s
INPUT (EMITTER)
Reverse voltage
VR
5
V
Forward current
IF
50
mA
Forward surge current
Power dissipation
tp ≤ 10 µs
IFSM
3
A
Tamb ≤ 25 °C
PV
100
mW
Tj
100
°C
Junction temperature
Document Number: 83751
Rev. 1.7, 17-Aug-09
For technical questions, contact:
www.vishay.com
1
CNY70
Reflective Optical Sensor with
Transistor Output
Vishay Semiconductors
ABSOLUTE MAXIMUM RATINGS
PARAMETER
(1)
TEST CONDITION
SYMBOL
VALUE
UNIT
Collector emitter voltage
VCEO
32
V
Emitter collector voltage
VECO
7
V
IC
50
mA
PV
100
mW
Tj
100
°C
OUTPUT (DETECTOR)
Collector current
Tamb ≤ 25 °C
Power dissipation
Junction temperature
Note
Tamb = 25 °C, unless otherwise specified
(1)
ABSOLUTE MAXIMUM RATINGS
P - Power Dissipation (mW)
300
Coupled device
200
Phototransistor
100
IR - diode
0
25
0
95 11071
75
50
100
Tamb - Ambient Temperature (°C)
Fig. 1 - Power Dissipation vs. Ambient Temperature
BASIC CHARACTERISTICS
(1)
PARAMETER
TEST CONDITION
SYMBOL
MIN.
TYP.
Collector current
VCE = 5 V, IF = 20 mA,
d = 0.3 mm (figure 1)
IC (2)
0.3
1.0
Cross talk current
VCE = 5 V, IF = 20 mA, (figure 2)
ICX (3)
600
nA
IF = 20 mA, IC = 0.1 mA,
d = 0.3 mm (figure 1)
VCEsat (2)
0.3
V
Forward voltage
IF = 50 mA
VF
Radiant intensity
IF = 50 mA, tp = 20 ms
Ie
IF = 100 mA
λP
Method: 63 % encircled energy
d
MAX.
UNIT
COUPLER
Collector emitter saturation
voltage
mA
INPUT (EMITTER)
Peak wavelength
Virtual source diameter
1.25
1.6
V
7.5
mW/sr
940
nm
1.2
mm
OUTPUT (DETECTOR)
Collector emitter voltage
IC = 1 mA
VCEO
32
V
Emitter collector voltage
IE = 100 µA
VECO
5
V
VCE = 20 V, IF = 0 A, E = 0 lx
ICEO
Collector dark current
200
nA
Notes
(1) T
amb = 25 °C, unless otherwise specified
(2) Measured with the "Kodak neutral test card", white side with 90 % diffuse reflectance
(3) Measured without reflecting medium
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For technical questions, contact:
Document Number: 83751
Rev. 1.7, 17-Aug-09
CNY70
Reflective Optical Sensor with
Transistor Output
Vishay Semiconductors
Reflecting medium
(Kodak neutral test card)
~
~~
~
~~
d
Detector
Emitter
A
C
C
E
95 10808
Fig. 2 - Pulse diagram
BASIC CHARACTERISTICS
Tamb = 25 °C, unless otherwise specified
10
IC - Collector Current (mA)
IF - Forward Current (mA)
1000
100
10
1
1
0.1
0.01
0.1
0.001
0.1
0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
VF - Forward Voltage (V)
96 11862
I F = 20 mA
1.2
d = 0.3 mm
I C - Collector Current (mA)
CTR rel - Relative Current Transfer Ratio
10
1.3
1.1
1.0
0.9
0.8
0.7
10
100
Fig. 5 - Collector Current vs. Forward Current
1.5
VCE = 5 V
1
I F - Forward Current (mA)
95 11065
Fig. 3 - Forward Current vs. Forward Voltage
1.4
Kodak neutral card
(white side)
d = 0.3 mm
VCE = 5 V
Kodak neutral card
(white side)
d = 0.3 mm
I F = 50 mA
20 mA
1
10 mA
5 mA
0.1
2 mA
0.01
1 mA
0.6
0.5
- 30 - 20 -10 0 10 20 30 40 50 60 70 80
96 11913
Tamb - Ambient Temperature (°C)
Fig. 4 - Relative Current Transfer Ratio vs. Ambient Temperature
Document Number: 83751
Rev. 1.7, 17-Aug-09
For technical questions, contact:
0.001
0.1
95 11066
1
10
100
VCE - Collector Emitter Voltage (V)
Fig. 6 - Collector Current vs. Collector Emitter Voltage
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CNY70
Reflective Optical Sensor with
Transistor Output
Vishay Semiconductors
10
Kodak neutral card
(white side)
d = 0.3 mm
V CE = 5 V
I C - Collector Current (mA)
CTR - Current Transfer Ratio (%)
100
10
1
0.1
0.1
1
10
d
0.1
V CE = 5 V
I F = 20 mA
0.001
0
100
I F - Forward Current (mA)
96 11914
1
2
Fig. 7 - Current Transfer Ratio vs. Forward Current
6
4
8
10
d - Distance (mm)
95 11069
Fig. 9 - Collector Current vs. Distance
0°
10°
20°
I erel - Relative Radiant Intensity
Icrel - Relative Collector Current
CTR - Current Transfer Ratio (%)
10
I F = 50 mA
1 mA
20 mA
10 mA
1
5 mA
2 mA
Kodak neutral card
(white side)
d = 0.3 mm
30°
40°
1.0
0.9
50°
0.8
60°
70°
0.7
80°
0.1
0.1
0.6
100
1
10
V CE - Collector Emitter Voltage (V)
96 12001
0.2
0.4
0
0.2
0.4
0.6
95 11063
Fig. 8 - Current Transfer Ratio vs. Collector Emitter Voltage
Fig. 10 - Relative Radiant Intensity/Collector Current vs.
Angular Displacement
I Crel - Relative Collector Current
1.0
0.9
E
d = 5 mm
4 mm
3 mm
2 mm
1 mm
0
0.7
0.6
0.5
0.4
s
D
5 mm
10 mm
d
0
E
s
5 mm
D
0.3
10 mm
VCE = 5 V
I F = 20 mA
0.2
0.1
0.0
0
96 11915
0
1.5
0.8
1
2
3
4
5
6
7
8
9
10
11
s - Displacement (mm)
Fig. 11 - Relative Collector Current vs. Displacement
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For technical questions, contact:
Document Number: 83751
Rev. 1.7, 17-Aug-09
CNY70
Reflective Optical Sensor with
Transistor Output
Vishay Semiconductors
PACKAGE DIMENSIONS in millimeters
95 11345
TUBE DIMENSIONS in millimeters
20291
Document Number: 83751
Rev. 1.7, 17-Aug-09
For technical questions, contact: sensorstechsupport@vishay.com
www.vishay.com
5
Legal Disclaimer Notice
Vishay
Disclaimer
All product specifications and data are subject to change without notice.
Vishay Intertechnology, Inc., its affiliates, agents, and employees, and all persons acting on its or their behalf
(collectively, “Vishay”), disclaim any and all liability for any errors, inaccuracies or incompleteness contained herein
or in any other disclosure relating to any product.
Vishay disclaims any and all liability arising out of the use or application of any product described herein or of any
information provided herein to the maximum extent permitted by law. The product specifications do not expand or
otherwise modify Vishay’s terms and conditions of purchase, including but not limited to the warranty expressed
therein, which apply to these products.
No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted by this
document or by any conduct of Vishay.
The products shown herein are not designed for use in medical, life-saving, or life-sustaining applications unless
otherwise expressly indicated. Customers using or selling Vishay products not expressly indicated for use in such
applications do so entirely at their own risk and agree to fully indemnify Vishay for any damages arising or resulting
from such use or sale. Please contact authorized Vishay personnel to obtain written terms and conditions regarding
products designed for such applications.
Product names and markings noted herein may be trademarks of their respective owners.
Document Number: 91000
Revision: 18-Jul-08
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1
Page 1
Vishay Semiconductors
Application of Optical Reflex Sensors
TCRT1000, TCRT5000, CNY70
Vishay Semiconductor optoelectronic sensors contain infrared-emitting diodes as a radiation source and
phototransistors as detectors.
Typical applications include:
Copying machines
Printers
Video recorders
Object counters
Proximity switch
Industrial control
Vending machines
Special features:
Compact design
Ambient light protected
Operation range 0 to 20 mm
Cut-off frequency up to 40 kHz
High sensitivity
High quality level, ISO 9000
Low dark current
Automated high-volume production
Minimized crosstalk
These sensors present the quality of perfected products. The components are based on Vishay Semiconductor’s
many years’ experience as one of Europe’s largest producers of optoelectronic components.
Sensor Drawings
94 9442
94 9318
TCRT1000
TCRT5000
94 9320
CNY70
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1
Document Number 80107
02-02
Page 1
Page 2
Vishay Semiconductors
Optoelectronic Sensors
In many applications, optoelectronic transmitters and
receivers are used in pairs and linked together
optically. Manufacturers fabricate them in suitable
forms. They are available for a wide range of
applications as ready-to-use components known as
couplers, transmissive sensors (or interrupters),
reflex couplers and reflex sensors. Increased
automation in industry in particular has heightened the
demand for these components and stimulated the
development of new types.
General Principles
The operating principles of reflex sensors are similar
to those of transmissive sensors. Basically, the light
emitted by the transmitter is influenced by an object or
a medium on its way to the detector. The change in the
light signal caused by the interaction with the object
then produces a change in the electrical signal in the
optoelectronic receiver.
The main difference between reflex couplers and
transmissive sensors is in the relative position of the
transmitter and detector with respect to each other. In
the case of the transmissive sensor, the receiver is
opposite the transmitter in the same optical axis, giving
a direct light coupling between the two. In the case of
the reflex sensor, the detector is positioned next to the
transmitter, avoiding a direct light coupling.
The transmissive sensor is used in most applications
for small distances and narrow objects. The reflex
sensor, however, is used for a wide range of distances
as well as for materials and objects of different shapes.
In the following chapters, we will deal with reflex
sensors placing particular emphasis on their
practical use. The components TCRT1000,
TCRT5000 and CNY70 are used as examples.
However, references made to these components and
their use apply to all sensors of a similar design.
The reflex sensors TCRT1000, TCRT5000 and
CNY70 contain IR-emitting diodes as transmitters and
phototransistors as receivers. The transmitters emit
radiation of a wavelength of 950 nm. The spectral
sensitivity of the phototransistors are optimized at this
wavelength.
There are no focusing elements in the sensors
described, though lenses are incorporated inside the
TCRT5000 in both active parts (emitter and detector).
The angular characteristics of both are divergent. This
is necessary to realize a position-independent function
for easy practical use with different reflecting objects.
In the case of TCRT5000, the concentration of the
beam pattern to an angle of 16° for the emitter and 30°
for the detector results in operation at an increased
range with optimized resolution. The emitting and
acceptance angles in the other reflex sensors are
about 45°. This is an advantage in short distance
operation.
The main difference between the sensor types is the
mechanical outline (as shown in the figures, see
previous page ), resulting in various electrical
parameters and optical properties. A specialization for
certain applications is necessary. Measurements and
statements on the data of the reflex sensors are made
relative to a reference surface with defined properties
and precisely known reflecting properties. This
reference medium is the diffusely reflecting Kodak
neutral card, also known as gray card (KODAK neutral
test card; KODAK publi-cation No. Q-13,
CAT 1527654). It is also used here as the reference
medium for all details. The reflection factor of the white
side of the card is 90% and that of the gray side is 18%.
Table 1 shows the measured reflection of a number of
materials which are important for the practical use of
sensors. The values of the collector current given are
relative and correspond to the reflection of the various
surfaces with regard to the sensor’s receiver. They
were measured at a transmitter current of IF = 20 mA
and at a distance of the maximum light coupling.
These values apply to all reflex sensors. The
‘black-on-white paper’ section stands out in table 1.
Although all surfaces appear black to the ‘naked eye’,
the black surfaces emit quite different reflections at a
wavelength of 950 nm. It is particularly important to
account for this fact when using reflex sensors. The
reflection of the various body surfaces in the infrared
range can deviate significantly from that in the visible
range.
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Document Number 80107
02-02
Page 2
Page 3
Vishay Semiconductors
Table 1. Relative collector current (or coupling factor) of the reflex sensors for reflection on various materials.
Reference is the white side of the Kodak neutral card. The sensor is positioned perpendicular to the surface.
The wavelength is 950 nm.
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ÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁÁ
Kodak neutral card
White side (reference medium)
Gray side
Paper
Typewriting paper
Drawing card, white (Schoeller Durex)
Card, light gray
Envelope (beige)
Packing card (light brown)
Newspaper paper
Pergament paper
Black on white typewriting paper
Drawing ink (Higgins, Pelikan, Rotring)
Foil ink (Rotring)
Fiber-tip pen (Edding 400)
Fiber-tip pen, black (Stabilo)
Photocopy
Plotter pen
HP fiber–tip pen (0.3 mm)
Black 24 needle printer (EPSON
LQ-500)
Ink (Pelikan)
Pencil, HB
100%
20%
94%
100%
67%
100%
84%
97%
30-42%
4-6%
50%
10%
76%
7%
84%
28%
100%
26%
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Plastics, glass
White PVC
Gray PVC
Blue, green, yellow, red PVC
White polyethylene
White polystyrene
Gray partinax
Fiber glass board material
Without copper coating
With copper coating on the reverse
side
Glass, 1 mm thick
Plexiglass, 1 mm thick
Metals
Aluminum, bright
Aluminum, black anodized
Cast aluminum, matt
Copper, matt (not oxidized)
Brass, bright
Gold plating, matt
Textiles
White cotton
Black velvet
90%
11%
40-80%
90%
120%
9%
12-19%
30%
9%
10%
110%
60%
45%
110%
160%
150%
110%
1.5%
Parameters and Practical Use of the Reflex Sensors
A reflex sensor is used in order to receive a reflected
signal from an object. This signal gives information on
the position, movement, size or condition (e.g. coding)
of the object in question. The parameter that describes
the function of the optical coupling precisely is the
so-called optical transfer function (OT) of the sensor.
It is the ratio of the received to the emitted radiant
power.
OT r
e
Additional parameters of the sensor, such as operating
range, the resolution of optical distance of the object,
the sensitivity and the switching point in the case of local changes in the reflection, are directly related to this
optical transfer function.
In the case of reflex sensors with phototransistors as
receivers, the ratio Ic/IF (the ratio of collector current Ic
to the forward current IF) of the diode emitter is preferred to the optical transfer function. As with
optocouplers, Ic/IF is generally known as the coupling
factor, k. The following approximate relationship exists
between k and OT:
k = Ic/ IF = [(S B)/h] r/e
where B is the current amplification, S = Ib/Φr
(phototransistor’s spectral sensitivity), and h = IF/Φe
(proportionality factor between IF and Φe of the
transmitter).
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Document Number 80107
02-02
Page 3
Page 4
Vishay Semiconductors
In figures7 and 8, the curves of the radiant intensity, Ie,
of the transmitter to the forward current, IF, and the
sensitivity of the detector to the irradiance, Ee, are
shown respectively. The gradients of both are equal to
unity slope.
This represents a measure of the deviation of the
curves from the ideal linearity of the parameters. There
is a good proportionality between Ie and IF and
between Ic and Ee where the curves are parallel to the
unity gradient.
Greater proportionality improves the relationship
between the coupling factor, k, and the optical transfer
function.
surface and the frequency that is, the speed of
reflection change.
For all reflex sensors, the curve of the coupling factor
as a function of the transmitter current, IF, has a flat
maximum at approximately 30 mA (figure 9). As
shown in the figure, the curve of the coupling factor
follows that of the current amplification, B, of the
phototransistor. The influence of temperature on the
coupling factor is relatively small and changes
approximately –10% in the range of –10 to +70°C
(figure 10). This fairly favorable temperature
compensation is attributable to the opposing
temperature coefficient of the IR diode and the
phototransistor.
The maximum speed of a reflection change that is
detectable by the sensor as a signal is dependent
either on the switching times or the threshold
frequency, fc, of the component. The threshold
frequency and the switching times of the reflex
sensors TCRT1000, TCRT5000, and CNY70 are
determined by the slowest component in the system in this case the phototransistor. As usual, the threshold
frequency, fc, is defined as the frequency at which the
value of the coupling factor has fallen by 3 dB
(approximately 30%) of its initial value. As the
frequency increases, f > fc, the coupling factor
decreases.
Figure 7. Radiant intensity, Ie = f (IF), of the IR transmitter
Coupling Factor, k
In the case of reflex couplers, the specification of the
coupling factor is only useful by a defined reflection
and distance. Its value is given as a percentage and
refers here to the diffuse reflection (90%) of the white
side of Kodak neutral card at the distance of the
maximum light coupling. Apart from the transmitter
current, IF, and the temperature, the coupling factor
also depends on the distance from the reflecting
Figure 8. Sensitivity of the reflex sensors’ detector
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Document Number 80107
02-02
Page 4
Page 5
Vishay Semiconductors
from the diagram fc as a function of the load
resistance, RL.
Working Diagram
The dependence of the phototransistor collector
current on the distance, A, of the reflecting medium is
shown in figures 12 and 13 for the reflex sensor
TCRT1000.
Figure 9. Coupling factor k = f (IF) of the reflex sensors
The data were recorded for the Kodak neutral card
with 90% diffuse reflection serving as the reflecting
surface, arranged perpendicular to the sensor. The
distance, A, was measured from the surface of the
reflex sensor.
The emitter current, IF, was held constant during the
measurement. Therefore, this curve also shows the
course of the coupling factor and the optical transfer
function over distance. It is called the working diagram
of the reflex sensor.
The working diagrams of all sensors (figure 12) shows
a maximum at a certain distance, Ao. Here the optical
coupling is the strongest. For larger distances, the
collector current falls in accordance with the square
law. When the amplitude, I, has fallen not more than
50% of its maximum value, the operation range is at its
optimum.
Figure 10. Change of the coupling factor, k,
with temperature, T
As a consequence, the reflection change is no longer
easily identified.
Figure 11 illustrates the change of the cut-off
frequency at collector emitter voltages of 5, 10 and
20 V and various load resistances. Higher voltages
and low load resistances significantly increase the
cut-off frequency.
The cut-off frequencies of all Vishay Semiconductor
reflex sensors are high enough (with 30 to 50 kHz) to
recognize extremely fast mechanical events.
In practice, it is not recommended to use a large load
resistance to obtain a large signal, dependent on the
speed of the reflection change. Instead, the opposite
effect takes place, since the signal amplitude is
markedly reduced by the decrease in the cut-off
frequency. In practice, the better approach is to use the
given data of the application (such as the type of
mechanical movement or the number of markings on
the reflective medium). With these given data, the
maximum speed at which the reflection changes can
be determined, thus allowing the maximum frequency
occurring to be calculated. The maximum permissible
load resistance can then be selected for this frequency
Figure 11. Cut-off frequency, fc
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a) TCRT5000
c) TCRT1000
b) CNY 70
Figure 12. Working diagram of reflex sensors TCRT5000, CNY70 and TCRT1000
Resolution, Trip Point
The behavior of the sensors with respect to abrupt
changes in the reflection over a displacement path is
determined by two parameters: the resolution and the
trip point.
If a reflex sensor is guided over a reflecting surface
with a reflection surge, the radiation reflected back to
the detector changes gradually, not abruptly. This is
depicted in figure 13a. The surface, g, seen jointly by
the transmitter and detector, determines the radiation
received by the sensor. During the movement, this
surface is gradually covered by the dark reflection
range. In accordance with the curve of the radiation
detected, the change in collector current is not abrupt,
but undergoes a wide, gradual transition from the
higher to the lower value.
As illustrated in figure 13b, the collector current falls to
the value Ic2, which corresponds to the reflection of the
dark range, not at the point Xo, but at the points
Xo + Xd/2, displaced by Xd/2.
The displacement of the signal corresponds to an
uncertainty when recording the position of the
reflection change, and it determines the resolution and
the trip point of the sensor.
The trip point is the position at which the sensor has
completely recorded the light/ dark transition, that is,
the range between the points Xo + Xd/2 and Xo – Xd/2
around Xo. The displacement, Xd, therefore,
corresponds to the width or the tolerance of the trip
point. In practice, the section lying between 10 and
90% of the difference Ic = Ic1 – Ic2 is taken as Xd. This
corresponds to the rise time of the generated signal
since there is both movement and speed. Analogous
to switching time, displacement, Xd, is described as a
switching distance.
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The resolution is the sensor’s capability to recognize
small structures. Figure 13 illustrates the example of
the curve of the reflection and current signal for a black
line measuring d in width on a light background (e.g.
on a sheet of paper). The line has two light/ dark transitions the switching distance Xd/2 is, therefore,
effective twice.
Reflection
R1
Line
d = line width
R2
d
Collector current
X
IC1
Xd < line width
IC2
a)
Collector current
Xd
X
IC1
IC2
Xd > line width
g
Xd
X
Figure 14. Reflection of a line of width d and
corresponding curve of the collector current Ic
b)
Figure 14 shows the dependence of the switching
distance, Xd, on the distance A with the sensors placed
in two different positions with respect to the separation
line of the light/ dark transition.
The curves marked position 1 in the diagrams
correspond to the first position. The transmitter/
detector axis of the sensor was perpendicular to the
separation line of the transition. In the second position
(curve 2), the transmitter/ detector axis was parallel to
the transition.
In the first position (1) all reflex sensors have a better
resolution (smaller switching distances) than in
position 2. It can recognize lines smaller than half a
millimeter at a distance below 0.5 mm.
Figure 13. Abrupt reflection change with
associated Ic curve
The line is clearly recognized as long as the line width
is d Xd. If the width is less than Xd, the collector
current change, Ic1 – Ic2, that is the processable signal,
becomes increasingly small and recognition increasingly uncertain. The switching distance or better its
inverse can therefore be taken as a resolution of the
sensor.
The switching distance, Xd, is predominantly dependent on the mechanical/ optical design of the sensor
and the distance to the reflecting surface. It is also influenced by the relative position of the transmitter/
detector axis.
It should be remarked that the diagram of TCRT5000
is scaled up to 10 cm. It shows best resolution
between 2 and 10 cm.
All sensors show the peculiarity that the maximum
resolution is not at the point of maximum light coupling,
Ao, but at shorter distances.
In many cases, a reflex sensor is used to detect an
object that moves at a distance in front of a
background, such as a sheet of paper, a band or a
plate. In contrast to the examples examined above, the
distances of the object surface and background from
the sensor vary.
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Since the radiation received by the sensor’s detector
depends greatly on the distance, the case may arise
when the difference between the radiation reflected by
the object on the background is completely equalized
by the distance despite varying reflectance factors.
Even if the sensor has sufficient resolution, it will no
longer supply a processable signal due to the low reflection difference. In such applications it is necessary
to examine whether there is a sufficient contrast. This
is performed with the help of the working diagram of
the sensor and the reflectance factors of the materials.
a) TCRT5000
c) TCRT1000
b) CNY70
Figure 15. The switching distance as a function of the distance A for the reflex sensors TCRT5000,
CNY70 andTCRT1000
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Sensitivity, Dark Current and Crosstalk
The lowest photoelectric current that can be
processed as a useful signal in the sensor’s detector
determines the weakest usable reflection and defines
the sensitivity of the reflex sensor. This is determined
by two parameters – the dark current of the
phototransistor and the crosstalk.
The phototransistor as receiver exhibits a small dark
current, ICEO, of a few nA at 25°C. However, it is
dependent on the applied collector-emitter voltage,
VCE, and to a much greater extent on the temperature,
T (see figure 16). The crosstalk between the
transmitter and detector of the reflex sensor is given
with the current, Icx. Icx is the collector current of the
photoelectric transistor measured at normal IR
transmitter operating conditions without a reflecting
medium.
For design and optical reasons, the transmitter and
detector are mounted very close to each other.
Electrical interference signals can be generated in the
detector when the transmitter is operated with a pulsed
or modulated signal. The transfer capability of the
interference increases strongly with the frequency.
Steep pulse edges in the transmitter’s current are
particularly effective here since they possess a large
portion of high frequencies. For all Vishay Semiconductor sensors, the ac crosstalk, Icxac, does not
become effective until frequencies of 4 MHz upwards
with a transmission of approximately 3 dB between the
transmitter and detector.
The dark current and the dc - and ac crosstalk form the
overall collector fault current, Icf. It must be observed
that the dc-crosstalk current, Icxdc, also contains the
dark current, ICEO, of the phototransistor.
Icf = Icxdc + Icxac
This current determines the sensitivity of the reflex
sensor. The collector current caused by a reflection
change should always be at least twice as high as the
fault current so that a processable signal can be reliably identified by the sensor.
Ambient Light
Ambient light is another feature that can impair the
sensitivity and, in some circumstances, the entire
function of the reflex sensor. However, this is not an
artifact of the component, but an application specific
characteristic.
Figure 16. Temperature-dependence of the collector dark
current
It is ensured that no (ambient) light falls onto the
photoelectric transistor. This determines how far it is
possible to guarantee avoiding a direct optical
connection between the transmitter and detector of the
sensor.
At IF = 20 mA, the current Icx is approximately 15 nA
for the CNY70, TCRT1000 and TCRT5000.
Icx can also be manifested dynamically. In this case,
the origin of the crosstalk is electrical rather than
optical.
The effect of ambient light falling directly on the
detector is always very troublesome. Weak steady
light reduces the sensor’s sensitivity. Strong steady
light can, depending on the dimensioning (RL, VC),
saturate the photoelectric transistor. The sensor is
‘blind’ in this condition. It can no longer recognize any
reflection change. Chopped ambient light gives rise to
incorrect signals and feigns non-existent reflection
changes.
Indirect ambient light, that is ambient light falling onto
the reflecting objects, mainly reduces the contrast
between the object and background or the feature and
surroundings. The interference caused by ambient
light is predominantly determined by the various
reflection properties of the material which in turn are
dependent on the wavelength.
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If the ambient light has wavelengths for which the ratio
of the reflection factors of the object and background
is the same or similar, its influence on the sensor’s
function is small. Its effect can be ignored for
intensities that are not excessively large. On the other
hand, the object/ background reflection factors can
differ from each other in such a way that, for example,
the background reflects the ambient light much more
than the object. In this case, the contrast disappears
and the object cannot be detected. It is also possible
that an uninteresting object or feature is detected by
the sensor because it reflects the ambient light much
more than its surroundings.
determine the ambient light and its effects precisely.
Therefore, an attempt to keep its influence to a
minimum is made from the outset by using a suitable
mechanical design and optical filters. The detectors of
the sensors are equipped with optical filters to block
such visible light. Furthermore, the mechanical design
of these components is such that it is not possible for
ambient light to fall directly or sideways onto the
detector for object distances of up to 2 mm.
In practice, ambient light stems most frequently from
filament, fluorescent or energysaving lamps. Table 2
gives a few approximate values of the irradiance of
these sources. The values apply to a distance of
approximately 50 cm, the spectral range to a distance
of 850 to 1050 nm. The values of table 2 are only
intended as guidelines for estimating the expected
ambient radiation.
AC operation of the reflex sensors offers the most
effective protection against ambient light. Pulsed
operation is also helpful in some cases.
In practical applications, it is generally rather difficult to
If the ambient light source is known and is relatively
weak, in most cases it is enough to estimate the
expected power of this light on the irradiated area and
to consider the result when dimensioning the circuit.
Compared with dc operation, the advantages are
greater transmitter power and at the same time
significantly greater protection against faults. The only
disadvantage is the greater circuit complexity, which is
necessary in this case. The circuit in figure 20 is an
example of operation with chopped light.
Table 2. Examples for the irradiance of ambient light sources
Light source (at 50 cm distance)
Irradiance Ee (µW/cm 2)
850 to 1050 nm
Steady light
AC light (peak value)
500
25
30
14
16
Frequency (Hz)
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Filament lamp (60 W)
Fluorescent lamp OSRAM (65 W)
Economy lamp OSRAM DULUX (11 W)
100
100
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Vishay Semiconductors
Application Examples, Circuits
The most important characteristics of the Vishay
Semiconductor reflex sensors are summarized in
table 3. The task of this table is to give a quick
comparison of data for choosing the right sensor for a
given application.
Application Example with Dimensioning
As shown in figure 17, the coupling factor is at its
maximum. In addition, the degradation (i.e. the
reduction of the transmitted IR output with aging) is
minimum for currents under 40 mA (< 10% for
10000 h) and the self heating is low due to the power
loss (approximately 50 mW at 40 mA).
With a simple application example, the dimensioning
of the reflex sensor can be shown in the basic circuit
with the aid of the component data and considering the
boundary conditions of the application.
The reflex sensor is used for speed control. An
aluminum disk with radial strips as markings fitted to
the motor shaft forms the reflecting object and is
located approximately 3 mm in front of the sensor. The
sensor signal is sent to a logic gate for further
processing.
Dimensioning is based on dc operation, due to the
simplified circuitry.
The optimum transmitter current. IF, for dc operation is
between 20 and 40 mA. IF = 20 mA is selected in this
case.
+5 V
TCRT5000
74HCTXX
Q
RS
180
RE
15 k
GND
Figure 17. Reflex sensor - basic circuit
Table 3.
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Parameter
Distance of optimum coupling
Distance of best resolution
Coupling factor
Switching distance (min.)
Optimum working distance
Operating range
Symbol
A0
Ar
k
xd
Xor
Aor
CNY70
0.3 mm
0.2 mm
5%
1.5 mm
0.2 to 3 mm
9 mm
Reflex Sensor Type
TCRT1000
1 mm
0.8 mm
5%
0.7 mm
0.4 to 2.2 mm
8 mm
TCRT5000
2 mm
1.5 mm
6%
1.9 mm
0.2 to 6.5 mm
> 20 mm
Table 4.
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Application Data
Aluminum disk
Diameter 50 mm, distance from the sensor 3 mm, markings printed on the aluminum
Markings
8 radial black stripes and 8 spacings, the width of the stripes and spacings in front of
the sensor is approximately = 4 mm (in a diameter of 20 mm)
Motor speed
1000 to 3000 rpm
Temperature range 10 to 60°C
Ambient light
60 W fluorescent lamp, approximate distance 2 m
Power supply
5 V ± 5%
Position of the
Position 1, sensor/ detector connecting line perpendicular to the strips
sensor
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Special attention must also be made to the
downstream logic gate. Only components with a low
input offset current may be used. In the case of the TTL
gate and the LS-TTL gate, the ILH current can be
applied to the sensor output in the low condition. At
–1.6 mA or –400 µA, this is above the signal current of
the sensor. A transistor or an operational amplifier
should be connected at the output of the sensor when
TTL or LS-TTL components are used. A gate from the
74HCTxx family is used.
According to the data sheet, its fault current ILH is
approximately 1 µA.
The expected collector current for the minimum and
maximum reflection is now estimated.
According to the working diagram in figure 12a, it
follows that when A = 3 mm
Ic = 0.95 Icmax
Icmax is determined from the coupling factor, k,
for IF = 10 mA.
Icmax = k IF
In addition, 1 µA, the fault current of the 74HCTxx
gate, is also added
Ic2 = 49.5 µA
The effect of the indirect incident ambient light can
most easily be seen by comparing the radiant powers
produced by the ambient light and the sensor’s
transmitter on 1 mm2 of the reflecting surface. The
ambient light is then taken into account as a
percentage in accordance with the ratio of the powers.
From table 2:
Ee (0.5 m) = 40 µW/ cm2 (dc + ac/ 2)
Ee (2 m) = Ee(0.5 m) (0.5/ 2)2
(Square of the distance law)
Ee (2 m) = 2.5 µW/ cm2
sf = 0.025 W
The radiant power (Φsf = 0.025 µW)
therefore falls on 1 mm2.
At IF = 10 mA, the typical value
k = 2.8%
When IF = 10 mA, the sensor’s transmitter has the
radiant intensity:
is obtained for k from figure 9.
However, this value applies to the Kodak neutral card
or the reference surface. The coupling factor has a
different value for the surfaces used (typewriting paper
and black-fiber tip pen). The valid value for these
material surfaces can be found in table 1:
k1 = 94% k = 4.7% for typing paper and
k2 = 10% k = 0.5% for black-tip pen (Edding)
Therefore:
Ic2. Crosstalk with only a few nA for the TCRT5000 is
ignored. However, the dark current can increase up to
1 µA at a temperature of 70°C and should be taken into
account.
Ic1 = 0.95 k1 IF = 446.5 µA
Ic2 = 0.95 k2 IF = 47.5 µA
Ie e
0.25 Wsr
(see figure 7)
The solid angle for 1 mm2 surface at a distance of
3 mm is
2
1 mm 2 1 sr
9
(3 mm)
It therefore follows for the radiant power that:
Temperature and aging reduce the collector current.
They are therefore important to Ic1 and are subtracted
from it.
Figure 10 shows a change in the collector current of
approximately 10% for 70°C. Another 10% is deducted
from Ic1 for aging
Ic1 = 263 µA – (20% 263 µA) = 357.2 µA
The fault current Icf (from crosstalk and collector dark
current) increases the signal current and is added to
e = Ie = ca. 27.8 mW
The power of 0.025 µW produced by the ambient light
is therefore negligibly low compared with the
corresponding power (approximately 28 µW) of the
transmitter.
The currents Ic1, Ic2 would result in full reflecting
surfaces, that is, if the sensor’s visual field only
measures white or black typing paper. However, this
is not the case. The reflecting surfaces exist in the form
of stripes.
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The signal can be markedly reduced by the limited
resolution of the sensor if the stripes are narrow. The
suitable stripe width for a given distance should
therefore be selected from figure 15. In this case, the
minimum permissible stripe width is approximately
2.5 mm for a distance of 3 mm (position 1, figure 15a).
The markings measuring 4 mm in width were
expediently selected in this case. For this width, a
signal reduction of about 20% can be permitted with
relatively great certainty, so that 10% of the difference
(Ic1 – Ic2) can be subtracted from Ic1 and added to Ic2.
Ic1 = 357.2 A – 30.8 A = 326.4 A
Ic2 = 49.5 A + 30.8 A = 80.3 A
The suitable load resistance, RE, at the emitter of the
photo-transistor is then determined from the low and
high levels 0.8 V and 2.0 V for the 74HCTxx gate.
RE < 0.8 V/ Ic2 and RE > 2.0 V/ Ic1,
i.e., 6.1 k < RE < 9.96 k
6.8 k is selected for RE
The corresponding levels for determining RE must be
used if a Schmitt trigger of the 74HCTxx family is
employed.
The frequency limit of the reflex sensor is then
determined with RE = 6.8 k and compared with the
maximum operating frequency in order to check
whether signal damping attributable to the frequency
can occur.
Figure 11 shows for Vs = 5 V and RE = 6.8 k
approximately, for the TCRT5000, fc = 3.0 kHz.
Sixteen black/ white stripes appear in front of the
sensor in each revolution. This produces a maximum
signal frequency of approximately 400 Hz for the
maximum speed of 3000 rpm up to 50 rps. This is
significantly less than the fc of the sensor, which
means there is no risk of signal damping.
In the circuit in figure 17, a resistor, Rc, can be used on
the collector of the photoelectric transistor instead of
RE. In this case, an inverted signal and somewhat
modified dimensioning results. The current Ic1 now
determines the low signal level and the current Ic2 the
high. The voltages (Vs – 2 V) and (Vs – 0.8 V) and not
the high level and low level 2 V and 0.8 V, are now
decisive for determining the resistance, Rc.
Circuits with Reflex Sensors
The couple factor of the reflex sensors is relatively
small. Even in the case of good reflecting surfaces, it
is less than 10%. Therefore, the photocurrents are in
practice only in the region of a few µA. As this is not
enough to process the signals any further, an
additional amplifier is necessary at the sensor output.
Figure 18 shows two simple circuits with sensors and
follow-up operational amplifiers.
The circuit in figure 18b is a transimpedance which
offers in addition to the amplification the advantage of
a higher cut-off frequency for the whole layout.
Two similar amplification circuits
transistors are shown in figure 19.
incorporating
The circuit in figure 20 is a simple example for
operating the reflex sensors with chopped light. It uses
a pulse generator constructed with a timer IC. This
pulse generator operates with the pulse duty factor of
approximately 1. The frequency is set to
approximately 22 kHz. On the receiver side, a
conventional LC resonance circuit (fo = 22 kHz) filters
the fundamental wave out of the received pulses and
delievers it to an operational amplifier via the capacitor,
Ck. The LC resonance circuit simultaneously
represents the photo transistor’s load resistance. For
direct current, the photo transistor’s load resistance is
very low in this case approximately 0.4, which
means that the photo transistor is practically shorted
for dc ambient light.
At resonance frequencies below 5 kHz, the necessary
coils and capacitors for the oscillator become unwieldy
and expensive. Therefore, active filters, made up with
operational amplifiers or transistors, are more suitable
(figures 21 and 22). It is not possible to obtain the
quality characteristics of passive filters. In addition the
load resistance on the emitter of the photo transistor
has remarkably higher values than the dc resistance
of a coil. On the other hand, the construction with
active filters is more compact and cheaper. The
smaller the resonance frequency becomes, the
greater the advantages of active filters compared to
LC resonant circuits.
In some cases, reflex sensors are used to count steps
or objects, while at the same time recognition of a
change in the direction of rotation (= movement
direction) is necessary. The circuit shown in figure 23
is suitable for such applications. The circuit is
composed of two independent channels with reflex
sensors. The sensor signals are formed via the
Schmitt trigger into TTL impulses with step slopes,
which are supplied to the pulse inputs of the binary
counter 74LS393. The outputs of the 74LS393 are
coupled to the reset inputs. This is made in such a way
that the first output, whose condition changes from
‘low’ to ‘high’, sets the directly connected counter. In
this way, the counter of the other channel is deleted
and blocked. The outputs of the active counter can be
displaced or connected to more electronics for
evaluation.
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It should be mentioned that such a circuit is only suited
to evenly distributed objects and constant
movements. If this is not the case, the channels must
be close to each other, so that the movement of both
sensors are collected successively. The circuit also
works perfectly if the last mentioned condition is
fulfilled. Figure 24 shows a pulse circuit combining
analog with digital components and offering the
possibility of temporary storage of the signal delivered
by the reflex sensor. A timer IC is used as the pulse
generator.
The negative pulse at the timer’s output triggers the
clock input of the 74HCT74 flip-flop and, at the same
a)
time, the reflex sensor’s transmitter via a driver
transistor. The flip-flop can be positively triggered, so
that the condition of the data input at this point can be
received as the edge of the pulse rises. This then
remains stored until the next rising edge.
The reflex sensor is therefore only active for the
duration of the negative pulse and can only detect
reflection changes within this time period. During the
time of negative impulses, electrical and optical
interferences are suppressed. A sample and hold
circuit can also be employed instead of the flip-flop.
This is switched on via an analog switch at the sensor
output as the pulse rises.
b)
+10 V
+10 V
RF
IF = 20 mA Reflex sensor
IF = 20 mA Reflex sensor
220 k
7
TLC271
6
2
2
3
RS
390
7
3
Output
Output
4
RE
1k
TLC271
6
4
RF
RE
1k
RS
390
220 k
RI
1k
RI
1k
GND
GND
Figure 18. Circuits with operational amplifier
a)
b)
+10 V
+10 V
RC
1k
RE
220
RL
1k
IF = 20 mA
Reflex sensor
BC178B
PNP
IF = 20 mA
CK
Reflex sensor
RS
390
RF
Output
2.2 F
RL
10 k
RS
390
220 k
Output
BC108B
PNP
RE
1k
GND
GND
Figure 19. Circuits with transistor amplifier
www.vishay.com
14
Document Number 80107
02-02
Page 14
Page 15
Vishay Semiconductors
VS = +5 V
82
1.2 k
Reflex sensor
8
7
2.7 k
6
3
Q
CV 5
DIS
R
THR
TR
2
4
555
CK
GND
100 nF
1
10 nF
100 nF
C
62 nF
7
TLC 271
6
3
Output
2
4
RF
L
0.86 mH
10 k
100
GND
Figure 20. AC operation with oscillating circuit to suppress ambient light
+VS (10 V)
RS
220
Reflex sensor
CF
1 nF
RA
9.1 k
4 Timer
8
7
RQ 3
DIS
RB
5.1 k
6
2
C
R
33 k
33 k
2
7
6
1 F
3
THR
TR
CK
R
CV
5
GND
1
RE
510
555
100 nF
R1
1k
Cq
4
Output
TLC 271
(CA3160)
22 nF
100 nF
GND
GND
Timer dimensions:
Active filter :
tp (pulse width) = 0.8 RC = 400 s
T (period)
= 0.8 (RA + RB) C = 1 ms
C
Cf Cq
Q
fo 1(6.28 C R)
Cq
Cf
V uo 2 R Q 2
RE
Figure 21. AC operation with active filter made up of an operational amplifier, circuit and dimensions
www.vishay.com
15
Document Number 80107
02-02
Page 15
Page 16
Vishay Semiconductors
+VS (10 V)
RV
220
CF
RA
9.1 k
Reflex sensor
4 Timer
8
7
RB
5.1 k
6
THR
2
TR
CK
RQ 3
DIS
1
CK
51 k
51 k
1 F
555
100 nF
Output
NPN
Cq
RE
1.8 k
GND
C
R
1.5 nF
R
1 F
5
CV
RC
1k
33 nF
100 nF
GND
GND
Timer dimensions:
tp (pulse width) = 0.8 RC = 400 ms
T (period)
= 0.8 (RA + RB) C = 1 ms
C
Active filter :
Cf C q
Q
Cq
Cf
V uo 2 R Q 2
RE
fo 1(6.28 C R)
Figure 22. AC operation with transistor amplifier as active filter
Left
A
+5 V
Reflex sensor
Display system
QA
QB
QC
QD
CLK
CLR
B
CLK QA
QB
QC
CLR QD
LS393
A
A
RE
RD
CLK
Q
D
74HCT14
15 k
or report
LS393
Q
+5 V
3.3 k
SD
GND
Reset
+5 V
Q SD
D
CLK
Reflex sensor
B
Q
RE
100
15 k
74HCT14
GND
B7474
A
CLK
RV
RD
CLR
QA
QB
QC
QD
LS393
Display system
B
CLK
CLR
GND
Right
QA
QB
QC
QD
or report
LS393
Figure 23. Circuit for objects count and recognition of movement direction
www.vishay.com
16
Document Number 80107
02-02
Page 16
Page 17
Vishay Semiconductors
VS (+5 V)
82
RA
RC
3.3k
R1
PNP
PNP
4
CK
8
7
RB
DIS
RQ
6
THR
2
CV
TR GND
555
C
100
4
Reflex
sensor
3
2
3
1
SD Q 5
CLK
D
RD
Q
6
74HCT74
5
R2
1
100 nF
GND
Figure 24. Pulse circuit with buffer storage
www.vishay.com
17
Document Number 80107
02-02
Page 17
Freescale Semiconductor, Inc.
On-Chip Peripheral Systems
Page 18
Freescale Semiconductor, Inc...
A Practical Motor Control Example
In this section, we will develop a practical application by expanding some
of the software developed in this book. The example will add some
external hardware to the MC68HC705K1 so that we can observe the
effects of our software on the world outside the microcontroller. We will
use a slightly modified version of the PWM routine that was developed
in this chapter to control the speed of a small permanent-magnet direct
current (DC) motor. In addition, we will use the concepts developed in
the chapter titled On-Chip Peripheral Systems that allow the CPU to
read the state of switches connected to the MCU’s general-purpose I/O
pins.
Theory
DC motors are often the best choice for variable-speed motor
applications. Brush DC motors are the easiest to control electronically.
Electronic control of brushless DC, stepper, AC induction, and switched
reluctance motors all require more-complex control circuits in addition to
more power-switching devices. Small, low-cost brush DC motors are
available off the shelf for many low-volume applications where custom
designs would be too expensive. The reliability of brush motors is
adequate for most applications. However, eventually, the brushes will
wear out and need to be replaced.
To vary the speed of a brush DC motor, we must vary the voltage that is
applied to the motor. Several approaches can be used to accomplish
this. We will examine several of the methods, explaining the major
advantages and disadvantages of each.
The first and most obvious approach to varying the voltage applied to a
motor might be to place a variable resistor in series with the motor and
the power source, as shown in Figure 47. While this approach is very
simple, it has some serious disadvantages. First, the resistor’s power
dissipation capabilities must be matched to the power requirements of
the motor. For very small fractional-horsepower DC motors, the size of
the variable resistor will be quite modest. However, as the size of the
motor increases, the motor’s power requirement increases and the size
and cost of the variable resistor will increase.
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
198
On-Chip Peripheral
Page 18 Systems
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Freescale Semiconductor, Inc.
Page 19
On-Chip Peripheral Systems
A Practical Motor Control Example
M
MOTOR
Figure 47. Motor Speed Controlled by a Variable Resistor
Freescale Semiconductor, Inc...
The second major disadvantage of this type of speed control is the
inability to automatically adjust the speed of the motor to compensate for
varying loads. This is a primary disadvantage for applications that
require precise speed control under varying mechanical loads.
An electronic variation of the variable resistor form of speed control is
shown in Figure 48. In this arrangement, we have replaced the variable
resistor with a transistor. Here, the transistor is operated in its linear
mode. When a transistor operates in this mode, it essentially behaves as
an electrically controlled variable resistor. By applying a proportional
analog control signal to the transistor, the "resistivity" of the transistor
can be varied, which will in turn vary the speed of the motor. By using a
transistor to control the speed of the motor in this manner, the magnitude
of the control signal is reduced to much lower voltage and current levels
that can be readily generated by electronic circuity.
M
MOTOR
RB
VBB
Figure 48. Motor Speed Controlled by a Transistor
Unfortunately, using a transistor in its linear mode still retains a major
disadvantage of using a variable resistor. Like a variable resistor, a
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
MOTOROLA
On-Chip Peripheral
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Freescale Semiconductor, Inc.
Page 20
On-Chip Peripheral Systems
power transistor operating in its linear region will have to dissipate large
amounts of power under varying speed and load conditions. Even
though power transistors capable of handling high power levels are
widely available at relatively modest prices, the power dissipated by the
transistor will usually require a large heat sink to prevent the device from
destroying itself.
Freescale Semiconductor, Inc...
In addition to being operated as a linear device, transistors also may be
operated as electronic switches. By applying the proper control signal to
a transistor, the device will either be turned on or turned off. As shown in
Figure 49, when the transistor is turned on, it will essentially behave as
a mechanical switch allowing electric current to pass through it and its
load virtually unimpeded. When turned off, no current passes through
the transistor or its load. Because the transistor dissipates very little
power when it is fully turned on or saturated, the device operates in an
efficient manner.
M
VCC
IC
RB
VBB
M
VCC
VCE = VCC
RB
VCE ≅ 0 VOLTS
TRANSISTOR “ON”
IC = 0
VBB = 0
TRANSISTOR “OFF”
Figure 49. Transistor Used as an Electronic Switch
It would seem that, when using a transistor to control the speed of a DC
motor, we are stuck using the device in its inefficient linear mode if we
want a motor to operate at something other than full speed. Fortunately,
there is an alternative method of controlling the speed of a DC motor
using a transistor. By using the transistor as an electronically controlled
switch and applying a PWM control signal of sufficient frequency, we can
control the speed of the motor. To help understand how turning a motor
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
200
On-Chip Peripheral
Page 20 Systems
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MOTOROLA
Freescale Semiconductor, Inc.
Page 21
On-Chip Peripheral Systems
A Practical Motor Control Example
fully on and then fully off can control its speed, consider the PWM
waveforms in Figure 50.
T1
T2
+5 VOLTS
0 VOLTS
Freescale Semiconductor, Inc...
a) DUTY CYCLE = T2/T1 = 50%
T1
T2
+5 VOLTS
0 VOLTS
b) DUTY CYCLE = T2/T1 = 80%
Figure 50. PWM Waveforms with 50 and 80 Percent Duty Cycles
Figure 50(a) shows a single cycle of a 50 percent duty cycle PWM
waveform that is 5 volts during the first half of its period and at 0 volts
during the second half. If we integrate (or average) the voltage of the
PWM waveform in Figure 50(a) over its period, T1, the average DC
voltage is 50 percent of 5 volts or 2.5 volts. Correspondingly, the
average DC voltage of the PWM waveform in Figure 50(b), which has a
duty cycle of 80 percent, is 80 percent of 5 volts or 4.5 volts. By using a
PWM singal to switch a motor on and off in this manner, it will produce
the same effect as applying a continuous or average DC voltage at
varying levels to the motor. The frequency of the PWM signal must be
sufficiently high so that the rotational inertia of the motor integrates the
on/off pulses and causes the motor to run smoothly.
Motor Control
Circuit
As mentioned earlier, we will be using a slightly modified version of our
PWM routine to control the speed of a small motor. However, before
discussing the software involved, we need to take a look at the hardware
components required to drive the motor.
Figure 51 is a schematic diagram of the power section of our motor
control circuit. There are a number of differences between this
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
MOTOROLA
On-Chip Peripheral
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201
Freescale Semiconductor, Inc.
On-Chip Peripheral Systems
Page 22
schematic and the conceptual ones used in Figure 48 and Figure 49.
We will describe these differences in the following paragraphs.
Freescale Semiconductor, Inc...
The most noticeable difference is the schematic symbol for the power
transistor that will be used as an electronic switch. This device is a power
MOSFET. Unlike the bipolar transistor shown in Figure 48 and
Figure 49, this special type of transistor is controlled by the magnitude
of a voltage applied to its gate. Additionally, this particular power
MOSFET, the MTP3055EL, may be completely saturated with only 5
volts applied to its gate. These two characteristics allow this device to be
controlled directly by a microcontroller’s output pin for many
applications.
Because the input iimpedance of a power MOSFET is very high (greater
than 40 megaohms), a 10 KΩ resistor is placed between the MOSFET
gate and ground to prevent erratic operation of the motor should the
connection between the microcontroller and the gate ever become cut.
The 15-volt zener diode is placed in parallel with the resistor to protect
the gate of the MOSFET from possible damage from high voltage
transients that may be generated in the system. The 1N4001 diode in
parallel with the motor is used to snub the inductive kick of the motor
each time the MOSFET is turned off. The 0.1-µf capacitor in parallel with
the motor is used to reduce the electrical noise generated by the motor’s
brushes.
For further information on designing with power MOSFETs, it is
suggested that the reader study the Theory and Applications section of
the Motorola Power MOSFET Transistor Data Book (DL153).
Figure 52 is a schematic diagram of the microprocessor section of the
circuit that we will be using in this example. In addition to generating a
PWM output, the MC68HC705K1 is reading three momentary
pushbutton switches connected to its I/O pins. As the schematic shows,
a single switch turns the motor on and off while two switches set the
speed of the motor.
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
202
On-Chip Peripheral
Page 22 Systems
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MOTOROLA
Freescale Semiconductor, Inc.
Page 23
On-Chip Peripheral Systems
A Practical Motor Control Example
+5 V
1N4001
M
0.1 F
FROM PA7 OF
MC68HC705K1
Freescale Semiconductor, Inc...
10 k
MTP3055EL
15 V
Figure 51. Power Section of the Motor Speed Control Circuit
+5 V
MC68HC705K1
27 pF
10 k
(5)
MOTOR
CONTROL
SWITCHES
OSC1
10 M
4 MHz
RESET
OSC2
27 pF
IRQ
ON/OFF
PA0
SPEED
DOWN
PA1
SPEED
UP
PA2
TO GATE OF
MTP3055EL
PA7
+5 V
VSS
VDD
0.1 F
Figure 52. Microcontroller Section
of the Motor Speed Control Circuit
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
MOTOROLA
On-Chip Peripheral
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On-Chip Peripheral Systems
Page 24
Freescale Semiconductor, Inc...
One side of each switch is connected to circuit ground, while the other
side of the switch is connected to an I/O pin on the MC68HC705K1
microcontroller. Each of the input pins on the microcontroller is "pulled
up" through a 10-kΩ resistor to +5 volts. These 10-kΩ pullup resistors
keep each of the three input pins at a logic 1 when the pushbutton
switches are not pressed.
In this exampel circuit, the switch controls will operate in the following
manner. The motor on/off switch operates as an alternate-action control.
Each time the switch is pushed and released, the motor will alternately
be turned on or off. When the motor is turned on, its speed will be set to
the speed it was going the last time the motor was on.
The speed up and speed down switches increase or decrease motor
speed, respectively. To increase or decrease the speed of the motor, the
respective switch must be pressed and held. The motor speed PWM will
be increased or decreased at a rate of approximately 0.4 percent every
24 ms. This "ramp" rate will allow the motor speed to be adjusted across
its entire speed range in approximately six seconds.
Motor Control
Software
Figure 53 shows a flowchart that describes the new RTI interrupt
software. The only functional change to the PWM routine developed
earlier in this chapter is the addition of one instruction at the beginning
of the RTI interrupt service routine. This instruction decrements the
variable RTIDlyCnt. This variable is used by the three routines that read
the input switches to develop a switch debounce delay.
As mentioned in the Programming chapter, there are usually many
ways to perform a specific task using the microcontroller’s instruction
set. To demonstrate this, one part of the revised RTI interrupt routine has
been impmlemented in a slightly different manner. Remember, looking
at Listing 6. Speed Control Program Listing, that we had to split the
variable DesiredPWM into two parts, PWMFine and PWMCoarse. To do
this, we used a combination of shifts and rotates to place the upper four
bits of the A accumulator (DesiredPWM) into the lower four bits of the X
register (PWMCoarse) and the lower four bits of A into the upper four bits
of A (PWMFine). This method required nine bytes of program memory
and 26 CPU cycles.
M68HC05 Family — Understanding Small Microcontrollers — Rev. 2.0
204
On-Chip Peripheral
Page 24 Systems
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MOTOROLA
Page 25
$GYDQFHG 3RZHU 026)(7
IRL510A
FEATURES
BVDSS = 100 V
♦ Logic-Level Gate Drive
♦ Avalanche Rugged Technology
RDS(on) = 0.44Ω
♦ Rugged Gate Oxide Technology
♦ Lower Input Capacitance
ID = 5.6 A
♦ Improved Gate Charge
♦ Extended Safe Operating Area
TO-220
♦ Lower Leakage Current: 10µA (Max.) @ VDS = 100V
♦ Lower RDS(ON): 0.336Ω (Typ.)
1
2
3
1.Gate 2. Drain 3. Source
Absolute Maximum Ratings
Symbol
VDSS
ID
Value
Units
Drain-to-Source Voltage
Characteristic
100
V
Continuous Drain Current (TC=25°C)
5.6
Continuous Drain Current (TC=100°C)
4.0
A
IDM
Drain Current-Pulsed
VGS
Gate-to-Source Voltage
±20
V
EAS
Single Pulsed Avalanche Energy
(2)
62
mJ
A
(1)
A
20
IAR
Avalanche Current
(1)
5.6
EAR
Repetitive Avalanche Energy
(1)
3.7
mJ
dv/dt
Peak Diode Recovery dv/dt
(3)
6.5
V/ns
37
W
PD
TJ , TSTG
TL
Total Power Dissipation (TC=25°C)
Linear Derating Factor
°C
0.25
Operating Junction and
- 55 to +175
Storage Temperature Range
°C
Maximum Lead Temp. for Soldering
300
Purposes, 1/8 from case for 5-seconds
Thermal Resistance
Symbol
Characteristic
Typ.
Max.
RθJC
Junction-to-Case
--
4.1
RθCS
Case-to-Sink
0.5
--
RθJA
Junction-to-Ambient
--
62.5
Units
°C/W
Rev. B
©1999 Fairchild Semiconductor Corporation
Page 25
1
1&+$11(/
32:(5 026)(7
Page 26
IRL510A
Electrical Characteristics (TC=25°C unless otherwise specified)
Symbol
Characteristic
BVDSS
Drain-Source Breakdown Voltage
100
--
--
∆BV/∆TJ
Breakdown Voltage Temp. Coeff.
--
0.1
--
IGSS
IDSS
RDS(on)
gfs
Gate Threshold Voltage
See Fig 7
VDS=5V,ID=250µA
--
2.0
--
100
Gate-Source Leakage , Reverse
--
--
-100
--
--
10
--
--
100
VDS=100V
µA V =80V,T =150°C
DS
C
--
--
0.44
Ω
VGS=5V,ID=2.8A
(4)
VDS=40V,ID=2.8A
(4)
Drain-to-Source Leakage Current
Static Drain-Source
On-State Resistance
3.2
--
Input Capacitance
--
180
235
Coss
Output Capacitance
--
50
65
Crss
Reverse Transfer Capacitance
--
20
25
td(on)
Turn-On Delay Time
--
8
25
Rise Time
--
10
30
Turn-Off Delay Time
--
17
45
Fall Time
--
8
25
5.5
8
Qg
V
--
--
tf
V/°C ID=250µA
1.0
Forward Transconductance
td(off)
VGS=0V,ID=250µA
Gate-Source Leakage , Forward
Ciss
tr
V
Test Condition
Ω
VGS(th)
Min. Typ. Max. Units
Total Gate Charge
--
Qgs
Gate-Source Charge
--
0.9
--
Qgd
Gate-Drain ( Miller ) Charge
--
3.5
--
nA
pF
VGS=20V
VGS=-20V
VGS=0V,VDS=25V,f =1MHz
See Fig 5
VDD=50V,ID=5.6A,
ns
RG=12Ω
See Fig 13
(4) (5)
VDS=80V,VGS=5V,
nC
ID=5.6A
See Fig 6 & Fig 12 (4) (5)
Source-Drain Diode Ratings and Characteristics
Symbol
Characteristic
IS
Continuous Source Current
Min. Typ. Max. Units
--
--
5.6
--
20
ISM
Pulsed-Source Current
(1)
--
VSD
Diode Forward Voltage
(4)
--
--
trr
Reverse Recovery Time
--
85
Qrr
Reverse Recovery Charge
--
0.23
A
Test Condition
Integral reverse pn-diode
in the MOSFET
1.5
V
TJ=25°C, IS=5.6A,VGS=0V
--
ns
TJ=25°C, IF=5.6A
--
µC
diF/dt=100A/µs
Notes;
(1) Repetitive Rating: Pulse Width Limited by Maximum Junction Temperature
(2) L=3mH, IAS=5.6A, VDD=25V, RG=27Ω, Starting TJ =25°C
(3) ISD ≤ 5.6A, di/dt ≤ 250A/µs, VDD ≤ BVDSS, Starting TJ =25°C
(4) Pulse Test: Pulse Width = 250µs, Duty Cycle ≤ 2%
(5) Essentially Independent of Operating Temperature
Page 26
2
Page 27
1&+$11(/
32:(5 026)(7
IRL510A
Fig 1. Output Characteristics
Fig 2. Transfer Characteristics
VGS
Top :
7.0 V
6.0 V
5.5 V
5.0 V
4.5 V
4.0 V
3.5 V
Bottom : 3.0 V
100
@ Notes :
1. 250 µs Pulse Test
2. TC = 25 oC
10-1 -1
10
0.8
RDS(on) , [ Ω ]
Drain-Source On-Resistance
ID , Drain Current [A]
101
100
175 oC
100
25 oC
@ Notes :
1. VGS = 0 V
2. VDS = 40 V
3. 250 µs Pulse Test
- 55 oC
10-1
101
0
2
4
6
8
10
VDS , Drain-Source Voltage [V]
VGS , Gate-Source Voltage [V]
Fig 3. On-Resistance vs. Drain Current
Fig 4. Source-Drain Diode Forward Voltage
IDR , Reverse Drain Current [A]
ID , Drain Current [A]
101
VGS = 5 V
0.6
0.4
VGS = 10 V
0.2
@ Note : TJ = 25 oC
101
100
@ Notes :
1. VGS = 0 V
2. 250 µs Pulse Test
175 oC
25 oC
-1
0.0
0
5
10
15
20
10
0.4
ID , Drain Current [A]
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
VSD , Source-Drain Voltage [V]
Fig 5. Capacitance vs. Drain-Source Voltage
Fig 6. Gate Charge vs. Gate-Source Voltage
350
Capacitance [pF]
280
C iss
210
C oss
140
@ Notes :
1. VGS = 0 V
2. f = 1 MHz
C rss
70
0
100
6
VGS , Gate-Source Voltage [V]
Ciss= Cgs+ Cgd ( Cds= shorted )
Coss= Cds+ Cgd
Crss= Cgd
VDS = 20 V
VDS = 50 V
VDS = 80 V
4
2
@ Notes : ID = 5.6 A
0
0
101
2
4
6
QG , Total Gate Charge [nC]
VDS , Drain-Source Voltage [V]
Page 27
3
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1&+$11(/
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IRL510A
Fig 7. Breakdown Voltage vs. Temperature
Fig 8. On-Resistance vs. Temperature
3.0
1.1
1.0
0.9
2.5
RDS(on) , (Normalized)
Drain-Source On-Resistance
BVDSS , (Normalized)
Drain-Source Breakdown Voltage
1.2
@ Notes :
1. VGS = 0 V
2.0
1.5
1.0
2. ID = 250 µA
0.8
-75
-50
-25
0
25
50
75
100
125
150
175
0.0
-75
200
-25
0
25
50
75
100
125
150
175
200
TJ , Junction Temperature [oC]
Fig 9. Max. Safe Operating Area
Fig 10. Max. Drain Current vs. Case Temperature
6
ID , Drain Current [A]
Operation in This Area
is Limited by R DS(on)
100 µs
101
1 ms
10 ms
DC
100
-50
TJ , Junction Temperature [oC]
102
@ Notes :
1. TC = 25 oC
2. TJ = 175 oC
5
4
3
2
1
3. Single Pulse
10-1 0
10
101
0
25
102
50
75
100
125
150
175
T , Case Temperature [oC]
VDS , Drain-Source Voltage [V]
c
Thermal Response
Fig 11. Thermal Response
D=0.5
0
10
10- 1
0.2
0.1
@ Notes :
1. Z J C (t)=4.1 o C/W Max.
0.05
2. Duty Factor, D=t /t
0.02
3. TJ M -TC =PD M *Z
θ
1
θ JC
2
(t)
PDM
0.01
t1
single pulse
θJC
Z (t) ,
ID , Drain Current [A]
@ Notes :
1. VGS = 5 V
2. ID = 2.8 A
0.5
t2
10- 5
10- 4
t
1
10- 3
10- 2
10- 1
, Square Wave Pulse Duration
Page 28
100
101
[sec]
4
Page 29
1&+$11(/
32:(5 026)(7
IRL510A
Fig 12. Gate Charge Test Circuit & Waveform
Current Regulator
VGS
Same Type
as DUT
50kΩ
Qg
200nF
12V
5V
300nF
VDS
Qgs
VGS
Qgd
DUT
3mA
R1
R2
Current Sampling (IG)
Resistor
Charge
Current Sampling (ID)
Resistor
Fig 13. Resistive Switching Test Circuit & Waveforms
RL
Vout
Vout
90%
VDD
Vin
( 0.5 rated VDS )
RG
DUT
Vin
10%
5V
tr
td(on)
td(off)
t on
tf
t off
Fig 14. Unclamped Inductive Switching Test Circuit & Waveforms
BVDSS
1
EAS = ---- LL IAS2 -------------------2
BVDSS -- VDD
LL
VDS
Vary tp to obtain
required peak ID
BVDSS
IAS
ID
RG
C
ID (t)
VDD
DUT
VDS (t)
VDD
5V
tp
tp
Page 29
Time
5
Page 30
1&+$11(/
32:(5 026)(7
IRL510A
Fig 15. Peak Diode Recovery dv/dt Test Circuit & Waveforms
DUT
+
VDS
--
IS
L
Driver
VGS
RG
VGS
VGS
( Driver )
Same Type
as DUT
VDD
dv/dt controlled by RG
IS controlled by Duty Factor
D
Gate Pulse Width
D = -------------------------Gate Pulse Period
5V
IFM , Body Diode Forward Current
IS
( DUT )
di/dt
IRM
Body Diode Reverse Current
VDS
( DUT )
Body Diode Recovery dv/dt
Vf
VDD
Body Diode
Forward Voltage Drop
Page 30
6
Page 31
L7800
SERIES
POSITIVE VOLTAGE REGULATORS
■
■
■
■
■
OUTPUT CURRENT TO 1.5A
OUTPUT VOLTAGES OF 5; 5.2; 6; 8; 8.5; 9;
12; 15; 18; 24V
THERMAL OVERLOAD PROTECTION
SHORT CIRCUIT PROTECTION
OUTPUT TRANSITION SOA PROTECTION
DESCRIPTION
The L7800 series of three-terminal positive
regulators is available in TO-220, TO-220FP,
TO-220FM, TO-3 and D2PAK packages and
several fixed output voltages, making it useful in a
wide range of applications. These regulators can
provide local on-card regulation, eliminating the
distribution problems associated with single point
regulation. Each type employs internal current
limiting, thermal shut-down and safe area
protection, making it essentially indestructible. If
adequate heat sinking is provided, they can
deliver over 1A output current. Although designed
primarily as fixed voltage regulators, these
devices can be used with external components to
obtain adjustable voltage and currents.
TO-220
D2PAK
TO-220FP
TO-220FM
TO-3
SCHEMATIC DIAGRAM
April 2004
1/33
Page 31
Page 32
L7800 SERIES
ABSOLUTE MAXIMUM RATINGS
Symbol
VI
Parameter
DC Input Voltage
Value
for VO= 5 to 18V
35
for VO= 20, 24V
40
Unit
V
Output Current
Internally Limited
Ptot
Power Dissipation
Internally Limited
Tstg
Storage Temperature Range
-65 to 150
°C
Top
Operating Junction Temperature for L7800
Range
for L7800C
-55 to 150
0 to 150
°C
IO
Absolute Maximum Ratings are those values beyond which damage to the device may occur. Functional operation under these condition is
not implied.
THERMAL DATA
Symbol
Parameter
Rthj-case Thermal Resistance Junction-case Max
Rthj-amb Thermal Resistance Junction-ambient
Max
D2PAK
TO-220
3
5
5
62.5
50
60
SCHEMATIC DIAGRAM
2/33
Page 32
TO-220FP TO-220FM
TO-3
Unit
5
4
°C/W
60
35
°C/W
Page 33
L7800 SERIES
CONNECTION DIAGRAM (top view)
TO-220 (Any Type)
TO-220FP/TO-220FM
D2PAK (Any Type)
TO-3
ORDERING CODES
TYPE
L7805
L7805C
L7852C
L7806
L7806C
L7808
L7808C
L7885C
L7809C
L7812
L7812C
L7815
L7815C
L7818
L7818C
L7820
L7820C
L7824
L7824C
TO-220
(A Type)
TO-220
(C Type)
L7805CV
L7852CV
TO-220
(E Type)
D2PAK
(A Type) (*)
D2PAK
(C Type)
(T & R)
TO-220FP
TO-220FM
L7805C-V L7805CV1 L7805CD2T L7805C-D2TR
L7852CD2T
L7805CP
L7852CP
L7805CF
L7852CF
L7806CV
L7806C-V
L7806CD2T
L7806CP
L7806CF
L7808CV
L7885CV
L7809CV
L7808C-V
L7809C-V
L7808CD2T
L7885CD2T
L7809CD2T
L7808CP
L7885CP
L7809CP
L7808CF
L7885CF
L7809CF
L7812CV
L7812C-V
L7812CD2T
L7812CP
L7812CF
L7815CV
L7815C-V
L7815CD2T
L7815CP
L7815CF
L7818CV
L7818CD2T
L7818CP
L7818CF
L7820CV
L7820CD2T
L7820CP
L7820CF
L7824CV
L7824CD2T
L7824CP
L7824CF
TO-3
L7805T
L7805CT
L7852CT
L7806T
L7806CT
L7808T
L7808CT
L7885CT
L7809CT
L7812T
L7812CT
L7815T
L7815CT
L7818T
L7818CT
L7820T
L7820CT
L7824T
L7824CT
(*) Available in Tape & Reel with the suffix "-TR".
3/33
Page 33
Page 34
L7800 SERIES
APPLICATION CIRCUITS
TEST CIRCUITS
Figure 1 : DC Parameter
Figure 2 : Load Regulation
4/33
Page 34
Page 35
L7800 SERIES
Figure 3 : Ripple Rejection
ELECTRICAL CHARACTERISTICS OF L7805 (refer to the test circuits, TJ = -55 to 150°C, VI = 10V,
IO = 500 mA, CI = 0.33 µF, CO = 0.1 µF unless otherwise specified).
Symbol
Parameter
Test Conditions
Min.
Typ.
Max.
4.8
5
5.2
V
PO ≤ 15W
4.65
5
5.35
V
mV
VO
Output Voltage
TJ = 25°C
VO
Output Voltage
IO = 5 mA to 1 A
VI = 8 to 20 V
∆VO(*)
Line Regulation
VI = 7 to 25 V
TJ = 25°C
3
50
VI = 8 to 12 V
TJ = 25°C
1
25
IO = 5 mA to 1.5 A
TJ = 25°C
100
IO = 250 to 750 mA
TJ = 25°C
25
∆VO(*)
Id
∆Id
Load Regulation
SVR
mV
Quiescent Current
TJ = 25°C
6
mA
Quiescent Current Change
IO = 5 mA to 1 A
0.5
mA
VI = 8 to 25 V
0.8
∆VO/∆T Output Voltage Drift
eN
Unit
IO = 5 mA
0.6
Output Noise Voltage
B =10Hz to 100KHz
Supply Voltage Rejection
VI = 8 to 18 V
Vd
Dropout Voltage
IO = 1 A
RO
Output Resistance
f = 1 KHz
Isc
Short Circuit Current
VI = 35 V
Iscp
Short Circuit Peak Current
TJ = 25°C
TJ = 25°C
f = 120Hz
mV/°C
40
68
µV/VO
dB
TJ = 25°C
2
2.5
TJ = 25°C
0.75
1.2
A
2.2
3.3
A
17
1.3
V
mΩ
(*) Load and line regulation are specified at constant junction temperature. Changes in VO due to heating effects must be taken into account
separately. Pulse testing with low duty cycle is used.
5/33
Page 35
Page 36
L7800 SERIES
Figure 4 : Dropout Voltage vs Junction
Temperature
Figure 7 : Output Voltage vs Junction
Temperature
Figure 5 : Peak Output Current vs Input/output
Differential Voltage
Figure 8 : Output Impedance vs Frequency
Figure 6 : Supply Voltage Rejection vs
Frequency
Figure 9 : Quiescent Current vs Junction
Temperature
15/33
Page 36
Page 37
L7800 SERIES
Figure 10 : Load Transient Response
Figure 12 : Quiescent Current vs Input Voltage
Figure 11 : Line Transient Response
Figure 13 : Fixed Output Regulator
NOTE:
1. To specify an output voltage, substitute voltage value for "XX".
2. Although no output capacitor is need for stability, it does improve transient response.
3. Required if regulator is locate an appreciable distance from power supply filter.
16/33
Page 37
Page 38
L7800 SERIES
Figure 14 : Current Regulator
Vxx
IO = 
+ Id
R1
Figure 15 : Circuit for Increasing Output Voltage
IR1 ≥ 5 Id
R2
VO = VXX (1+  ) + Id R2
R1
Figure 16 : Adjustable Output Regulator (7 to 30V)
17/33
Page 38
Page 39
L7800 SERIES
Figure 17 : 0.5 to 10V Regulator
R4
VO = V xx 
R1
Figure 18 : High Current Voltage Regulator
VBEQ1
R1 = 
IQ1
IREQ - 
βQ1
VBEQ1
IO = IREG + Q1 (IREG )
R1
Figure 19 : High Output Current with Short Circuit Protection
VBEQ2
RSC = 
ISC
18/33
Page 39
Page 40
L7800 SERIES
Figure 20 : Tracking Voltage Regulator
Figure 21 : Split Power Supply (± 15V - 1 A)
* Against potential latch-up problems.
Figure 22 : Negative Output Voltage Circuit
19/33
Page 40
Page 41
L7800 SERIES
Figure 23 : Switching Regulator
Figure 24 : High Input Voltage Circuit
VIN = VI - (VZ + VBE)
Figure 25 : High Input Voltage Circuit
Figure 26 : High Output Voltage Regulator
20/33
Page 41
Page 42
L7800 SERIES
Figure 27 : High Input and Output Voltage
VO = VXX + VZ1
Figure 28 : Reducing Power Dissipation with Dropping Resistor
VI(min) - VXX - VDROP(max)
R = 
IO(max) + Id(max)
Figure 29 : Remote Shutdown
21/33
Page 42
Page 43
L7800 SERIES
Figure 30 : Power AM Modulator (unity voltage gain, IO ≤ 0.5)
NOTE: The circuit performs well up to 100 KHz.
Figure 31 : Adjustable Output Voltage with Temperature Compensation
R2
VO = VXX (1+ )
+ V BE
R1
NOTE: Q2 is connected as a diode in order to compensate the variation of the Q1 VBE with the temperature. C allows a slow rise time of the VO.
Figure 32 : Light Controllers (VOmin = VXX + VBE)
VO rises when the light goes up
VO falls when the light goes up
22/33
Page 43
Page 44
L7800 SERIES
Figure 33 : Protection against Input Short-Circuit with High Capacitance Loads
Application with high capacitance loads and an output voltage greater than 6 volts need an external diode (see fig. 33) to protect the device
against input short circuit. In this case the input voltage falls rapidly while the output voltage decrease slowly. The capacitance discharges by
means of the Base-Emitter junction of the series pass transistor in the regulator. If the energy is sufficiently high, the transistor may be destroyed. The external diode by-passes the current from the IC to ground.
23/33
Page 44