COMPACT WIDEBAND TAG ANTENNA
FOR UHF RFID
Sang Ho Lim, Young Cheol Oh, Ho Lim, and Noh Hoon Myung
Department of Electrical Engineering and Computer Science, Korea
Advanced Institute of Science and Technology (KAIST), 373-1,
Kuseong Dong, Yuseong Gu, Daejeon, Korea; Corresponding author:
limsangho@kaist.ac.kr
Received 21 August 2008
ABSTRACT: A UHF radio frequency identification tag antenna using an inductively coupled feeding method and meander lines is proposed. The antenna has a compact size (diameter: 0.08␭, 26.4 mm),
wide bandwidth, and simple matching technique between a tag antenna and a tag chip. To get a physical operation of the proposed
antenna, a simplified structure is presented and analyzed. An equivalent circuit model with a simplified structure has good results compared with a simulation. © 2009 Wiley Periodicals, Inc. Microwave
Opt Technol Lett 51: 1291–1294, 2009; Published online in Wiley
InterScience (www.interscience.wiley.com). DOI 10.1002/mop.24295
Key words: tag antennas; compact; wideband; meander line; radio frequency identification (RFID)
1. INTRODUCTION
Radio frequency identification (RFID) systems can be grouped by
the size of objects for fixed applications, i.e., pallet level, box
level, and item level. For many applications, pallet and box level
RFID systems can be successfully used with a high recognition
rate. However, item level RFID systems need to be comparatively
more accurate and stable. In recent years, the requirements imposed on small object identifications have rapidly increased; thus,
compact tag antenna analysis and design have become mandatory
to obtain good performance from an item level RFID system. For
UHF ranges, the dipole-like tag antennas, which are used in many
Figure 2 Simplified structure and operation principle of the proposed
antenna. [Color figure can be viewed in the online issue, which is available
at www.interscience.wiley.com]
applications, are too big to attach to a small object. Therefore, a
high number of studies on small tag antennas have been performed.
Andrenko et al. [1] proposed a custom design for tag antennas
for stacked CDs. Even though these antennas have a small size
(diameter: 0.1␭, 33.2 mm) and an easy way to control the tag
antenna impedance, they do not have sufficient bandwidth or a
long read range. Circular loop antennas that use a short stub have
been suggested [2]. Such antennas have a small size (diameter:
0.12␭, 40 mm), a wide bandwidth, and an omni-directional radiation pattern. However, these antennas have been designed to
match 50 ⍀, so it is necessary to use another matching network
between a tag antenna and a tag chip. Such an adjustment might
cause reduction of the bandwidth, increase the fabrication cost, and
increase the structure complexity.
In this article, we present a compact wideband tag antenna for
UHF RFID, as shown in Figure 1. Using an inductively coupled
feeding method, a wide bandwidth and a simple matching technique between a tag antenna and a tag chip can be obtained without
any kind of matching network. A compact size can be achieved
with meander lines. To understand the physical operation of the
proposed tag antenna, we analyze and present an equivalent circuit
model with a simplified structure, as illustrated in Figure 2. Then,
we check the validation of the model through comparison between
the simulation and calculation of an equivalent circuit. Finally, we
suggest a compact (diameter: 0.08␭, 26.4 mm) wideband tag
antenna for a plastic lid.
2. ANALYSIS AND EQUIVALENT CIRCUIT MODEL
Figure 1 Geometry of the proposed antenna. [Color figure can be
viewed in the online issue, which is available at www.interscience.wiley.
com]
DOI 10.1002/mop
Figure 2 shows the simplified structure and operation principle of
the proposed antenna. Even though the structure does not have
meander lines, it will suffice to express the basic operation of the
proposed antenna. The structure consists of a loop shaped feeding
part with a tag chip and a split ring shaped radiating part. The
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 5, May 2009
1291
The solution to the mutual inductance of the simplified structure is
not known. Therefore, we analyzed the problem and validated the
results. In general, a mutual inductance can be expressed using (6),
where Ifp is a current on the feeding part, Bfp is the magnetic flux
density, which is created by Ifp, and Srp is the area of the radiating
part. If the feeding part, which has a radius FRs, is placed in the x-y
plane, centered at the origin, and carrying a constant current Ifp, the
current density has only a ␸ component, as in (7). The magnetic
vector potential, A, which can be derived from J, can be expressed
by using a complete elliptic integral of first kind and second kind
[5, 6].
M⫽
1
I fp
冕
srp
Bfp 䡠 dsrp共H兲
Figure 3 Equivalent circuit model of the simplified structure
(6)
␦共r⬘ ⫺ FRs兲
FRs
J ␾ ⫽ I fpsin␪⬘␦共cos␪⬘兲
circumference of the radiating part should be close to ␭/2 for the
purpose of effective radiation into free space.
The operation principle can be explained as follows. It is
possible to assume constant current flows on the feeding part,
because the feeding part is much smaller than the wavelength of
UHF frequency ranges. If the current flows in a counterclockwise
direction, it produces magnetic fields around the feeding loop. To
cancel the magnetic fields, new magnetic fields are induced near
the radiating part. Therefore, a current that flows in a clockwise
direction is also induced on the radiating part. It is the source to
radiate into free space. The simplified structure can be expressed as
an equivalent circuit model in Figure 3 [3]. The capacitance Ccouple
between the feeding part and the radiating part is negligible,
because the distance between the two parts is large. Therefore,
only inductive coupling survives. The input resistance and reactance of the inductively coupled feeding structure can be expressed
as (1) and (2) near the resonance frequency, where M is the mutual
inductance between the two parts, and Rrp,o and Qrp are the
radiation resistance and the quality factor of the radiating part at
resonance frequency, respectively. Lfp is the self inductance of the
feeding part [3]. The unknown variables (Lfp, M, Rrp,o, and Qrp) of
the equivalent circuit are analyzed and designed.
1
共2 ␲ fM兲 2
Ra ⫽
R rp,o 1 ⫹ 关Qrp共 f/fo ⫺ fo/f 兲兴2
X a ⫽ 2 ␲ fL fp ⫺
共2␲fM兲2 Qrp共 f/fo ⫺ fo/f 兲
Rrp,o 1 ⫹ 关Qrp共 f/fo ⫺ fo/f 兲兴2
(1)
(2)
The inductance of a circular loop which has radius “a” and a wire
radius “b” can be calculated by (3) [4]. However, the feeding part
of the simplified structure is a planar type, which has a radius FRs
and width FWs, as shown in Figure 2. Therefore, (3) needs to be
modified. If most currents flow on the surface of the wire, from (4),
the circumference of the wire is equal to the feeding part width of
the simplified structure. Substituting (4) in (3), we obtain the
inductance of the feeding part by (5).
冋冉冊 册
L loop ⫽ ␮oa ln
8a
⫺2
b
2 ␲ b ⬇ FWs 3 b ⬇ FWs/2␲
冋冉
冊 册
16␲FRs
⫺2
L fp ⬇ ␮oFRs ln
FWs
1292
K共k兲 ⫽
再 冉冊 冉 冊
冉
冊
再 冉冊 冉 冊
冉
冊
␲
1
1⫹
2
2
k2 ⫹
1䡠3
2䡠4
E共k兲 ⫽
␲
1
1⫺
2
2
2
k2 ⫺
1䡠3
2䡠4
A ␾共r, ␪ 兲 ⫽
k4 ⫹ 䡠 䡠 䡠
k 2n ⫹ 䡠 䡠 䡠
冎
(8)
k 2n
⫹ 䡠 䡠 䡠
2n ⫺ 1
冎
(9)
2
2
k4 ⫺ 䡠 䡠 䡠
共2n ⫺ 1兲!!
2 nn!
⫺
k2 ⫽
2
共2n ⫺ 1兲!!
2 nn!
⫹
2
4FRs rsin␪
FR ⫹ r2 ⫹ 2FRsrsin␪
(10)
2
s
␮0
4I fpFRs
4 ␲ 共FRs2 ⫹ r2 ⫹ 2FRsrsin␪ 兲1/ 2
⫻
冋
册
共2 ⫺ k2 兲 K共k兲 ⫺ 2E共k兲
k2
(11)
The magnetic flux density, B, can be calculated using (12). Since
the loop lies in the x-y plane (at ␪ ⫽ 90°), Br becomes zero and
only the B␪ component survives. Finally, substituting (12.2) in (6),
we can get the mutual inductance, M, between the feeding part and
the radiating part, as shown in (13). Because the current flow
direction of the feeding part is opposite to that of the radiating part,
the mutual inductance has a minus sign. The radiation resistance
and the reactance of a radiating part can be calculated by using
ADS Momentum. The quality factor of the split ring shaped
radiating part can be also calculated by using (14).
Br ⫽
1 ⭸
共sin␪A␾ 兲
rsin␪⭸␪
B␪ ⫽ ⫺
(3)
(4)
2
(7)
1⭸
共rA ␾兲
r ⭸r
B␾ ⫽ 0
M⫽
(12.1)
(12.2)
(12.3)
冋
册
共2 ⫺ k2 兲 K共k兲 ⫺ 2E共k兲
⫺ 2 ␮ 0FRsRRs
2
1/ 2 ⫻
共FR ⫹ RRs ⫹ 2FRsRRs兲
k2
2
s
(13)
(5)
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 5, May 2009
DOI 10.1002/mop
Figure 4 Input resistance and reactance of the simplified structure
Q rp ⫽
␻ dXr
␻o X2 ⫺ X1
兩␻ ⫽ ␻ o ⬇
2Rrp,o d␻
2Rrp,o␻2 ⫺ ␻1
(14)
The specific dimensions were calculated to use the simplified
structure in the UHF range. The results of the computation were
RRs ⫽ 23.8 mm, RWs ⫽ 2 mm, RGs ⫽ 2 mm, FRs ⫽ 5.2 mm, FWs
⫽ 1.5 mm, TagGs ⫽ 0.6 mm, and ds ⫽ 17.1 mm. The unknown
variables were determined to be Lfp ⫽ 20.6 (nH), M ⫽ ⫺2.37
(nH), Rrp,o ⫽ 9.85 (⍀), and Qrp ⫽ 58. To confirm the validation,
the input impedance of the equivalent circuit with the calculated
variables was compared with the input impedance obtained from
high frequency structural simulator (HFSS) simulation. Figure 4
shows good agreement; hence, the availability of the inductance of
the feeding part from (5) and the mutual inductance M from (13)
was confirmed.
3. ANTENNA DESIGN
A compact wideband tag antenna with meander lines was designed
and prototyped for an RFID tag chip with an input impedance of
6.2-j127 (⍀) [3], as shown in Figure 5. Indeed, there are parasitic
capacitances between the gaps of the meander lines. However, the
basic principle of operation is nearly the same as that of the
simplified structure. The design parameters of the prototype antenna are RRp ⫽ 8.4 mm, RWp ⫽ 4.8 mm, RGp ⫽ 0.5 mm, FRp
⫽ 2.2 mm, FWp ⫽ 2 mm, FGp ⫽ 0.3 mm, TagGp ⫽ 1 mm, and
dp ⫽ 4.2 mm (refer to Fig. 1). The antenna was printed on a thin
substrate of polytetrafluoroethylene (␧r ⫽ 3.5, tan ␦ ⫽ 0.001) using
copper traces (␴ ⫽ 5.8 ⫻ 107 S/m) with a thickness of 35 ␮m. The
proposed antenna is a kind of balanced symmetric type antenna.
Thus, the antenna input impedance was measured by measuring
half of the antenna over a 400 ⫻ 400 mm2 ground plane, as shown
in Figure 6, and multiplying the measured impedance by two.
Figure 7 shows the simulated return loss, which was obtained
using HFSS and the measured return loss of the antenna. The
simulation result at the reference level of 3 dB return loss was 102
MHz (862–964 MHz), whereas the measured result at the same
reference level was 102 MHz (856 –958 MHz). Both results approximately cover the UHF RFID frequency bandwidth (860 –960
MHz). Therefore, the prototype antenna can be operated continually in the UHF RFID frequency range.
Figures 8 and 9 show the simulation results for the input
impedance characteristic of the proposed tag antenna with a varying feeding part width, FWp, and a varying distance, dp, between
the radiating and the feeding parts. Figure 8 shows the results of
DOI 10.1002/mop
Figure 5 Prototype of the proposed antenna mounted on a bottle cap.
[Color figure can be viewed in the online issue, which is available at
www.interscience.wiley.com]
varying the feeding part width FWp without changing other parameters. The feeding part width FWp controls the input reactance
of the tag antenna. The reactance of the tag antenna is dependent
on the tag chip due to the maximization of the power delivery to
the tag antenna. The width FWp was 2 mm, which gives a complex
conjugate value for the chip impedance. As the width FWp increases, the reactance of the antenna increases. Therefore, it is easy
to control the reactance of the tag antenna.
Figure 9 shows the results when the distance dp between the
radiating and the feeding parts is varied. As seen in Figure 9, the
input resistance of the tag antenna is dependent on the distance dp
between the radiating and feeding parts. The inductively coupled
feeding method can control the input resistance of the antenna by
varying the mutual coupling between the radiating and feeding
Figure 6 Input impedance measurement of the proposed antenna. [Color
figure can be viewed in the online issue, which is available at www.interscience.wiley.com]
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 5, May 2009
1293
Figure 7 Return loss of the proposed antenna
Figure 9 Input resistance as a function of distance between the two parts
parts. The input resistance of the tag antenna increases in proportion to the squared mutual coupling in (1). As the distance dp
between the parts increases, the mutual coupling decreases. As the
mutual coupling decreases, the resistance of the antenna decreases.
Therefore, the input resistance of the tag antenna can be easily
adjusted.
A number of tag chips are fabricated with various impedances
according to the chip providers. Therefore, this method is convenient when designing tag antennas.
4. CONCLUSION
In this article, to understand the physical operation of the proposed
tag antenna, we have presented an analysis and an equivalent
circuit model of the simplified structure. We have validated the
model through a comparison between simulation and calculation
of the equivalent circuit. For use in plastic lids, we have presented
a compact (diameter: 0.08␭, 26.4 mm) wideband (102 MHz) tag
antenna for UHF RFID. Using an inductively coupled feeding
method, a wide bandwidth and a simple matching technique can be
obtained, while size reduction can also be achieved using meander
lines. The prototype antenna can be operated continually in the
UHF RFID frequency range.
ACKNOWLEDGMENT
This work was supported by KAIST BK 21(Brain Korea 21),
Agency for Defense Development (ADD) through the Radiowave
Detection Research Center (RDRC) at KAIST, and the Samsung
Advanced Institute of Technology (SAIT).
REFERENCES
1. A.S. Andrenko, M. Kai, T. Maniwa, and T. Yamagajo, Compact printed
on CD UHF RFID tag antennas, IEEE Antennas and Propagation
Society International Symposium, 2007, Honolulu, HI, pp. 5455–5458.
2. H.-K. Ryu and J.-M. Woo, Miniaturisation of circular loop antenna
using short stub for RFID system, Electron Lett 42 (2006), 955–956.
3. H.W. Son and C.S. Pyo, Design of RFID tag antennas using an inductively coupled feed, Electron Lett 41 (2005), 994 –996.
4. C.A. Balanis, Antenna theory, 3rd ed., Wiley Interscience, Hoboken,
NJ, 2005.
5. J.D. Jackson, Classical electrodynamics, Wiley, New York, NY, 1998.
6. I.S. Gradshteyn and I.M. Ryzhik, Table of integrals, series, and products, Academic press, Orlando, FL, 1980.
© 2009 Wiley Periodicals, Inc.
CASCADED DUAL-MODE SQUARELOOP RESONATORS USING HILBERTCURVE PERTURBATION FOR WIDE
BANDPASS FILTER APPLICATIONS
Ji-Chyun Liu,1 Chiung-Hung Li,1 Ching-Pin Kuei,2 and
Bing-Hao Zeng3
1
Department of Electrical Engineering, Ching Yun University, ChungLi, Tao-yuan 32097, Taiwan, Republic of China; Corresponding
author: jichyun@cyu.edu.tw
2
Department of Electronics Engineering, Ching Yun University,
Chung-Li, Tao-yuan 32097, Taiwan, Republic of China
3
Department of Communication Engineering, Yuan Ze University,
Chung-Li, Tao-yuan 32003, Taiwan, Republic of China
Received 26 August 2008
Figure 8
1294
Input reactance as a function of feeding part width
ABSTRACT: The cascaded dual-mode square-loop resonator
(DMSLR) with Hilbert-curve perturbation is introduced to design
wide band-pass filter in this article. To obtain low insertion loss
(⫺0.85 dB), high out-of-band rejection level (⫺51.82 dB), and wider
band (BW 62.3%) responses, using Hilbert-curve perturbation and
MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 5, May 2009
DOI 10.1002/mop