Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/1
06.09.16
Receiver Architectures
Markku Renfors
Department of Electronics and
Communications Engineering
Tampere University of Technology, Finland
markku.renfors@tut.fi
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/2
06.09.16
Topics
1. Main analog components for receivers
- amplifiers
- filters
- mixers
- oscillators
2. Receiver architectures and their properties
- superheterodyne principle
- direct conversion
- DC offsets as a challenging problem
- low IF, Weaver
- effects of I/Q imbalance
3. Non-idealities and performance meatrics of the
analog front-end modules
- sensitivity, dynamic range
- noise figure
- intermodulation distortion, IP3
- system calculation principles
- leakage, spurious frequencies
- phase noise
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/3
06.09.16
PART 1
What is needed in the receiver front-end?
· Amplification to compensate for transmission losses
· Selectivity to separate the desired signal from others
· Tunability to select the desired signal
· Conversion to digital domain
In the following we examine different receiver architectures
and non-idealities affecting in the different building blocks
used in the receivers.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/4
06.09.16
Main Analog Components for Receivers - 1
· Amplifiers
- Low-noise amplifiers (LNAs) in the first stages.
- Automatic gain control (AGC) needed to cope with
different signal levels.
· Filters
- Impossible to achieve sufficient selectivity by
tunable RF filters (operating in the RF frequency
band of the modulated signal) to separate the
desired signal from others.
- Sufficient selectivity can be achieved by fixed (RF or IF)
filters based on special technologies (SAW,
Surface Acoustic Wave, ceramic, crystal,
mechanical)
or active analog filters operating on basedband or low
bandpass center frequencies
or multirate digital filters up to some hundreds of MHz
range.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/5
06.09.16
Main Analog Components for Receivers - 2
· Filters (continued)
- Special complex filters, phase splitters (related also
to Hilbert transformers) can be used to suppress
certain frequency range from the negative part of
the frequency axis. Such filters find application in
certain special receiver architectures.
- Real bandpass signal and the corresponding ideal analytic
bandpass signal (obtained ideally through Hilbert
transform):
f
FILTERING
0
ß
W
f
0
- Other side of the desired signal spectrum suppressed by a
practical phase splitter with finite attenuation:
f
0
ß
0
fc
f
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/6
06.09.16
Main Analog Components for Receivers - 3
· Mixers
- Complex (I/Q, quadrature) mixer: pure frequency
translation by the local oscillator frequency:
Special case with
real input signal:
e
jwL O t
I
I
cos( wLOt)
f
fc
sin(w LOt)
f
fc+fLO
Q
I
- Real mixer produces the combination of frequency
translations in both directions:
-fc
-fc-fLO
0
-fc+fLO 0 fc-fLO
f
fc
fc +fLO
f
cos(w LO t)
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/7
06.09.16
Main Analog Components for Receivers - 4
· Oscillators
- Voltage (or current or digitally) controlled oscillators
(VCO, ICO, DCO) are used to generate the local
oscillator (LO) signals in a tunable manner.
- In a communications transceiver (receiver+transmitter,
RX+TX), the frequency synthesizer is one of the main
blocks. It is used for generating all the needed LO
signals in a controllable manner.
· Analog-to-digital interface
- Various ADC technologies
- Key performance metrics:
o Number of bits / Dynamic range / SNR
o Sampling jitter effects
- Main bottleneck in advanced DSP-based architectures.
- Room for innovative solutions
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/8
06.09.16
PART 2
Classical Receiver Architecture:
The Superheterodyne
AGC
RF-STAGES
MIXER
IF-STAGES
LO
RF-FILTER
fRF
0
fLO fRF +2fIF
-fLO
+fLO
IF-FILTER
0
fIF=fLO-fRF
Example: One common choice in GSM900 receivers has
been 1st IF = 71 MHz, 2nd IF = 13 MHz
Majority of all the receivers have been based on the
superheterodyne principle.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/9
06.09.16
Filtering Requirements in Superheterodynes
Selectivity is achieved at the IF stage(s) working at fixed
center frequency using special filter technologies.
· The RF filter should provide sufficient attenuation for the
image band at the distance of 2xfIF in frequency.
· The final IF stage should have sufficient selectivity to
suppress the neighbouring channels sufficiently.
· In case of double (or triple) super heterodyne, the first
(and second) IF stage should provide enough attenuation
at twice the next IF frequency.
· Image reject (I/Q-) mixer is one possibility to reduce the
RF filter requirements (but not sufficient as the only
solution)
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/10
06.09.16
Alternatives in Superheterodynes
Down-conversion rx
Upconversion rx
- fIF << fRF
- easier to get good
selectivity at first IF
- fIF > fRF
- easier to get good
image rejection
Low-side LO injection High-side LO injection
- fLO < fRF
- image band below fRF
- fLO > fRF
- image band above fRF
- spectrum inverted
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/11
06.09.16
Drawbacks of the Superheterodyne
Architecture
Some parts are difficult to integrate
· IF-filter
· RF-filter
· Oscillators
Power consumption high
· External components => parasitics
· Several submodules => low impedance (e.g., 50 W) levels
used for matching the modules
Complicated structure
è There is great interest for simpler architectures
which could be integrated more easily.
Spurious responses
· LO and IF signals and harmonics and mixtures leaking to
different places may cause problems.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/12
06.09.16
Direct-Conversion
Receiver Architecture
TUNED
TO RF
I
LO
LOWPASS
CHANNEL
FILTERS
90°
Q
ZERO IF
“Zero IF” -principle
Advantages
· No image bands => RF-filtering not so critical
· Not so much spurious responses
· Simple structure, no IF filters
Problem: Difficulties in implementation: dc offsets, leakage
between RX and TX in full duplex operation
Direct-conversion principle has become quite popular
in recent years in mobile terminals!
· This is mostly due to the availability of effective digital
calibration/compensation methods.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/13
06.09.16
DC-Offsets in Direct-Conversion Receivers
DC-offsets appear mainly due to LO leakage:
SELF-MIXING
OF INTERFERER
LO SELF-MIXING
LO
Constant DC-offset can be compensated by measuring it
without signal and then subtracting it during reception.
In TDMA systems, different channels/bursts may have
different signal levels and different AGC-values and hence
different DC-offsets => compensation is difficult.
Also 1/f -type of noise appearing in active components may
be a problem. This effect also appears at DC and low
frequencies.
Other issues
Signal leakage from antenna to the surroundings takes
place more easily than in superhet.
Sensitivity to 2nd-order intermodulation is another drawback
that will be discussed later.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/14
06.09.16
Low-IF Receiver Architecture
The idea is to use quadrature down-conversion and a low IF
frequency which is just high enough to cope with the DC-offset
problem (e.g., 250 kHz in case of GSM).
f
0
-fLO
-fIF
f
0
fIF
A COMPLEX
BANDPASS
FILTER
The channel filtering and final quadrature down-conversion can
be done after A/D-conversion in the DSP domain.
As we shall see on the next pages, quadrature down-conversion
in the analog domain may not, in practice, provide sufficient
attenuation for the image band.
More attenuation can be obtained by using a phase splitter
attenuating the image band on the negative part of the frequency
axis.
Even more attenuation could possibly be achieved through
baseband digital signal processing.
The low-IF concept is facilitated by the fact that system
specifications (like GSM) don't allow the maximum signal level to
appear in the nearest adjacent channels in case of a very low
desired signal level.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/15
06.09.16
Low-IF Receiver by Steyert et al.
I
PASSIVE
POLYPHASE
FILTER
MIRROR
SIGNAL
SUPPRESSION
+
FINAL
DOWNCONVERSION
Q
Q
I
HF LO
+
I in
+
Q in
-
I in
-
Q in
+
I out
+
Q out
-
I out
-
Q out
In this architecture, the image is suppressed
25 … 30 dB by the phase splitter implemented as a
polyphase RC network, and another 25 … 30 dB by
the quadrature downconversion approach.
I
Q
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/16
06.09.16
Weaver Receiver Architecture
LPF
RF
INPUT
sinw1t
sinw2t
cosw1 t
cosw2 t
LPF
Here the first LO frequency is fixed (using e.g., an IF
frequency of about 200 MHz in a DECT example) and the
second LO is used for channel selection.
In this architecture, the requirements for RF and IF filtering
are mild, and the channel selectivity is implemented at
baseband.
DC-offset problems of direct conversion receiver can be
reduced because there can be more amplification before
the second mixer.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/17
06.09.16
Oscillator Phase Quadrature,
Gain and Phase Imbalance in I/Q Systems
Quadrature downconversion is trying to produce a pure
frequency translation which would suppress the image band
completely.
In practice, there is some missmatch (imbalance) of gain
and/or phase in the components involved (oscillator,
amplifiers, mixers).
Consequently, the image suppression is, in practice, far
from complete.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/18
06.09.16
Image Rejection as a Function of Gain and
Phase Imbalance
Assuming that g is the gain imbalance ratio and f is the
phase difference due to imbalance, we can write:
y (t ) = x(t ) éëcos wct + jg sin (wct + f )ùû
é e jwct + e- jwct
e j (wct +f ) - e- ( jwct +f ) ù
= x (t ) ê
+g
ú
2
2
êë
úû
- jf ù
é jw t 1 + ge jf
- jwct 1 - ge
c
= x (t ) ê e
+e
ú
2
2
êë
úû
From the latter form, we can identify the strength of the two
spectral components produced by the two frequency
translations. The ratio of the image and desired signal
powers is obtained as
2
R =
1 - ge - jf
2
1 + ge
2
2
jf 2
=
1 + g 2 - 2 g cos f
1 + g 2 + 2 g cos f
This result can be applied to all cases of quadrature mixing
where gain and/or phase imbalance appears.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/19
06.09.16
Image Rejection as a Function of Gain and
Phase Imbalance
10
Amplitude imbalance in dB
0
Image attenuation [dB]
-10
2
-20
1
-30
0.5
0.2
-40
0.1
-50
0.05
0
-60
-70
-1
10
0
10
1
10
Phase imbalance in degrees
2
10
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/20
06.09.16
Frequency-Dependant Imbalance Model
Especially in wide-band systems, the imbalance parameters
may be frequency dependant and can be written as g(f) and
f(f).
Then also the image suppression ratio is frequencydependant:
2
R( f ) =
1 + g ( f ) 2 - 2 g ( f )cos f ( f )
1 + g ( f ) 2 + 2 g ( f )cos f ( f )
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/21
06.09.16
Effects of Gain and Phase Imbalance
In case of direct conversion-receiver (or final demodulation of an
I/Q-signal), gain mismatch and phase errors cause “self images”:
f
0
-fLO
+fLO
f
0
This is usually not a problem with low-order modulations, for
which an image attenuation of, e.g., 20 dB doesn’t essentially
effect the system performance. For high-order modulations, this
issue is more critical.
In other cases of quadrature down-conversion, like the low-IF
receiver, the image signal may be at a considerably stronger level
(up to 100 dB!!) than the desired signal, and I/Q imbalance is
very critical:
f
0
+fLO
-fLO
A COMPLEX
BANDPASS
FILTER
-fIF
f
0
fIF
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/22
06.09.16
Selectivity Requirements in System
Specifications
Radio system specifications (e.g., for cellular systems) don't
allow a strong adjacent channel signal to be present when a
weak desired signal is to be received. (Radio Resource
Management functionalities of wireless systems take care
of this!)
For example, the GSM specifications give:
· The maximum levels for the 3 adjacent channels on both
sides (at 200, 400 and 600 kHz from the carrier) in case
of a GSM interferer and desired signal 20 dB above the
reference sensitivity level of -102 dBm.
· Maximum levels for more distant signals (>600 kHz from
carrier), blocking signals, in case of a sinusoidal interferer
and desired signal 3 dB above the reference sensitivity
level.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/23
06.09.16
GSM Interference Mask
Note: This is a simplified picture. The related Invocom
demo gives better idea of the test cases.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/24
06.09.16
Summary 1:
Techniques for Providing Image Rejection in
Different Architectures
1. RF, first IF filters
- Challenges to get sufficient performance in integrated
solutions.
- Not applicable in low-IF or direct conversion cases.
2. Quadrature down-conversion
- Mostly for low-IF and direct conversion cases, but utilized
sometimes also in superheterodynes ("image reject mixer")
to simplify the RF filter requirements.
- Rejection limited by gain and phase imbalance.
- Produces complex signal and the consecutive signal
processing blocks must be duplicated, i.e., implemented for
both for I and Q branches.
3. Phase splitter (passive polyphase RC network)
- Mostly for low-IF and direct conversion cases, but can be
used also in superhets.
- Produces complex signal and the consecutive signal
processing blocks must be duplicated.
4. Advanced baseband digital signal processing
- If the imbalance parameters can be estimated, the effects
can be compensated fairly well. In recent years, various
effective and practical solutions have been developed for this
purpose.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/25
06.09.16
Summary 2:
Selectivity Tradeoffs
1. Selectivity at IF (superhet)
- discrete, high-cost IF filters
- less demands for analog circuits after IF
- simple A/D converter
2. Selectivity by analog baseband processing
(direct conversion case)
- no costly IF filters
- more RF gain needed
=> RF has to very linear to avoid intermodulation effects
- simple A/D converter
3. Selectivity by baseband digital filtering
(direct conversion and low-IF cases)
- no costly IF filters
- more flexible than case 2: selectivity can be easily
adapted to different systems
- more RF gain with high linearity needed
- high dynamic range A/D-converters (14…17 bits)
4. Selectivity by digital filtering after wideband IF
sampling
- simplified IF filter
- high flexibility
- suitable for multi-channel receivers with common
analog parts
- high dynamic range A/D-converters (14…17 bits)
- very strict demands for low jitter sampling clock
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/26
06.09.16
PART 3
Non-Idealities and Performance Metrics of the
Analog Front-End Modules
· Sensitivity of the receiver is mainly determined by the
noise produced by the receiver front-end components. It
determines the minimum detectable signal in noiselimited situation. In general:
Receiver sensitivity = thermal noise power (dBm)
+ noise figure (dB)
+ implementation loss (dB)
+ required SNR (dB)
In room temperature:
Thermal noise power (dBm) = -174 dBm+10 log10B
where B is the equivalent noise bandwidth of the RX.
For example in GSM, minimum S/N is 9 dB, B=200 kHz,
and required sensitivity is -102 dBm => NF<10 dB.
· With low received signal levels, the receiver front-end
components can be assumed to be linear. With higher
signal levels, the nonlinearity of the amplifiers and other
components produce harmful intermodulation products.
In this way the nonlinearity limits the dynamic range of
received signals from above.
Intermodulation is measured by 1 dB compression point
or the so-called IP3 figure.
In RF circuit design, there are always tradeoffs between
noise figure, linearity, and power consumption.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/27
06.09.16
Non-Idealities and Performance Measures of
the Analog Front-End Components (continued)
· Frequency accuracy and stability are determined by
the local oscillators of the receiver. In systems like GSM,
the receiver is locked to the network, and the frequency
stability is very good. However, to guarantee that the
receiver is able to synchronize to the network, certain
frequency accuracy, stability, and settling time
requirements are set for the components.
The short term instability of the oscillators appear as
phase noise, and it is very critical for the RF performance
of the system.
· Leakage effect means that strong signals, especially
local oscillator signals are connected, e.g., through
spurious capacitances to places where they are not
supposed to be connected. This means that various
harmonics, subharmonics, and mixtures of the local
oscillator frequencies are usually added to the signal.
In receiver design, the frequencies of the strongest
spurious frequencies can be calculated. By selecting the
local oscillator frequencies properly, most of the spurious
frequencies can be placed outside the desired frequency
bands at RF and IF.
· In the case of analog I/Q signal processing, amplitude
and phase responses of the I and Q branches are never
exactly the same. The effects of the gain and phase
imbalance depend greatly on the receiver architecture.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/28
06.09.16
Noise Figure
The noise factor of an amplifier stage (or some other
component) is determined by the ratio of S/N
ratios at the input and output:
SNRin
F=
SNRout
The noise figure is:
NF = 10 log10 F .
In this context only thermal noise in the input is taken into
account, i.e.,
Sin
SNRin =
N th
where Nth is the thermal noise power and Sin is the input
signal power. Assuming that g is the power gain of the
stage, the SNR at the output is:
SNRout
Sout gSin
=
=
N out N out
It follows that F = N out gN th
For SNR calculations, it is convenient to assume a model
with a single noise source at the receiver input:
N equiv = N out / g = N th + ( F - 1) N th
The latter part is the contribution of the considered stage.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/29
06.09.16
Overall Noise Figure
For a cascade of n stages, the overall noise factor is
Fn - 1
F - 1 F3 - 1
FT = F1 + 2
+
+L+
g1
g1g 2
g1g 2 L g n -1
where Fi and gi are the noise factor and power gain of
stage i .
This result can be easily derived using the previous results,
since the contribution of each stage to the equivalent input
noise source is
( Fi - 1) N th
g1 g 2 L gi -1
A passive component at the front-end (e.g., duplexer or RF
filter) doesn’t usually produce significant noise, but its
attenuation enhances the effects of the following noisy
amplifier stages. The noise figure of a passive component
is equal to its attenuation in dB. This is because the thermal
noise is present both at the input and output, and the noise
factor can be written as:
Fpassive
Sin / N th
1
=
= , with g < 1
gSin / N th g
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/30
06.09.16
Intermodulation
Consider a test where there are two nearby frequencies f 1 and
f 2 in the system frequency band (like in the neighbouring
channels).
In general, intermodulation produces new frequency components:
k1 f1 + k2 f 2
where k1 , k2 are integers. The order of an intermodulation product
is k1 + k2 .
Third-order intermodulation produces (among some others)
frequencies 2 f1 - f 2 , 2 f 2 - f1 which may fall in the desired signal
band.
2f1 -f2
f
f1
f2
Second-order intermodulation produces frequencies
f1 + f 2 , f 2 - f1 .
In general, with low-enough signal levels, the levels of secondand third-order intermodulation products are proportional to the
2nd and 3rd power of the fundamental signal level, respectively.
Low-order polynomial transfer characteristic can be used for
modelling memoryless nonlinearities. For example, a third-order
model produces only second- and third-order intermodulation. If
the signal transfer characteristic is antisymmetric with respect to
origin, only odd-order terms appear in the polynomial model, and
only odd-order intermodulation products appear. Differential
analog circuitry helps in this direction.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/31
06.09.16
IP3
The device-specific IP3-values can be used to determine
the strength of third-order intermodulation products, in
reference to the fundamental signal component.
IP3
MP
RO
DU
CT
RD
ER
I
INPUT-REFERRED
INTERCEPT POINT (IP3)
INPUT LEVEL, dBm
COMPRESSION
POINT
3R
D-O
N
FU
L
TA
N
ME
DA
SPURIOUS-FREE
DYNAMIC RANGE
OUTPUT LEVEL, dBm
1dB
In general, if the fundamental signal level (at a certain point
in the receiver chain) is x [dBm] and we know the IP3-value
calculated from the preceding stages, then third-order
intermodulation products are 2(IP3-x) [dB] below the
fundamental signal level (assuming, of course, that
x<<IP3).
These calculations are mostly relevant in the receiver
stages before the channel selection filtering, where there
are strong adjacent channel/blocking components. In the
calculations, the fundamental signal level is determined by
the worst-case blocker level.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/32
06.09.16
IP3 for a Cascade of Amplifiers
The overall output-referred IP3 can be estimated from the
output-referred IP3-values and power gains of the stages.
There are two alternative formulas:
1
1
=å
IP3 i IP3i g i +1 g i + 2 L g n
1
1
=å
IP32 i ( IP3i g i +1 g i + 2L g n )2
(IP3 in power units)
The reasoning behind these formulas is that the significant
signal components in each of the involved receiver stages
are at the same frequencies, only the levels are changed by
frequency-independent factors as the overall signal gets
amplified. Then, also the intermodulation products appear
at the same frequencies in all the stages.
The first formula assumes coherent summation of these
distortion products, the latter one assumes non-coherent
summation.
The latter formula is said to give more realistic results, but
the first one is safer and it is widely used in dimensioning
the system.
As mentioned, the above formulas are applicable in cases
where there is no significant frequency selectivity affecting
on the interesting signal components (i.e., the desired
signal and significant blockers).
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/33
06.09.16
IP2 for a Cascade of Amplifiers
As discussed below, 3rd-order intermodulation effects are
significant in traditional superheterodyne receiver design,
whereas 2nd-order intermodulation effects are less critical.
However, with direct-conversion and low-IF receivers, also
2nd-order effects are very important.
IP2 is defined in a similar way as IP3. In general, the
second-order intermodulation products are (IP2-x) [dB]
below the fundamental signal level x (assuming again that
x<<IP2). For amplifiers and other active components, the
IP2 values are typically somewhat higher than the IP3
values but, anyway, 2nd-order products tend to be stronger
than 3rd-order effects.
The overall output-referred IP2 can be estimated from the
output-referred IP2-values and power gains of the stages.
There are two alternative formulas:
1
1
=å
IP 2 i IP 2i gi +1gi + 2 L g n
1
1
=å
IP 2 i IP 2i gi +1gi + 2Lg n
(IP2 in power units)
Again, the first formula assumes coherent summation of
these distortion products, the latter one assumes noncoherent summation.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/34
06.09.16
Effects of 2nd and 3rd-Order Intermodulation
For equal IP2 and IP3 values, 2nd-order intermodulation
products are clearly stronger than 3rd-order products
=> If 2nd-order intermodulation products of some
strong signals (blocking signals) appear on the
signal band, then better linearity is required.
· In typical superheterodyne receivers, only 3rd-order
intermodulation is a problem, because signals causing
2nd-order products on the signal band are attenuated by
the RF filter.
· In wideband superhet with relatively low IF, also the 2ndorder intermodulation may become a problem.
· 2nd-order intermodulation is always a problem in directconversion and low-IF receivers.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/35
06.09.16
Example [2]: Analysing the Dynamic Range (1)
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/36
06.09.16
Example [2]: Analysing the Dynamic Range (2)
Small-signal case: AGC is set so that the the amplifiers
have maximum gain.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/37
06.09.16
Example [2]: Analysing the Dynamic Range (3)
Large-signal case: AGC is set so that the the
amplifiers have minimum gain.
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/38
06.09.16
Non-Idealities in Oscillators
In-band effects (i.e., they may exist even if no adjacent
channels and blockers are present):
· Constant phase error rotates the constellation; This can be
corrected by baseband processing afterwards
· Phase noise: random fluctuations in the instantaneous
phase/frequency of the oscillator cause random constellation
rotations, reducing the noise margin and increasing BER:
4
3
2
1
Q
0
-1
-2
-3
-4
-4
-3
-2
-1
0
1
2
3
4
I
The RF effects of phase noise tend to be far more critical
than in-band effects:
·
Mixing products of the phase noise spectrum and strong
adjacent channel signals (reciprocal mixing) may produce
spurious signals which overlap the desired signal. The following
figure shows the noisy LO spectrum, RF-spectrum, and
spectrum after mixing with practical LO with phase noise.
f
fLO
f
fc
f
fIF
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/39
06.09.16
Example of Phase Noise Calculations
VCO phase noise in a GSM 900 terminal
· wanted signal -102 dBm + 3 dB = -99 dBm
· blocking signal -43 dBm @ 600 kHz
· Minimum S/N=9 dB
· Noise bandwidth 200 kHz
Assume that the phase noise spectrum is flat within the
noise bandwidth with x dBc/Hz (i.e., the noise power in 1
Hz bandwidth is x dB in reference to the power of the VCO
at the LO frequency)
ð noise power at noise bandwidth:
-43 dBm+x+10 log10 200000 < -99 - 9 dBm
ð x < -118 dBc/Hz (@600 kHz)
f
fIF
fIF+600 kHz
Receiver Architectures
M. Renfors, TUT/ELT/WICO
ELT-44007/RxArch/40
06.09.16
Further reading
Superheterodyne architecture: [1], [2], [3] , [11]
· [1] is comprehensive discussion of the superheterodyne architecture and circuit design aspects
for the needed components.
Direct conversion architecture: [2], [3], [5]-[7], [10], [11]
Low-IF architecture: [3], [8], [9], [11]
Weaver architecture: [5] - [7], [11]
Noise figure and intermodulation analysis: [1], [2a], [3]
Phase noise: [1], [6]
I/Q-mismatch: [2], [8], [9]
Literature
[1] U.L. Rohde, J.C. Whitaker, T.T.N. Bucher, Communications Receivers: Principles and Design,
2nd Edition. McGraw-Hill 1997.
[2] L.E. Larson (Ed.), RF and Microwave Circuit Design for Wireless Communications. Artech
House 1996. (Chapters 2 and 3)
[2a] J.Y.C. Cheah, "Introduction to wireless communications applications and circuit
design," pp. 17-41.
[2b] A.A. Abidi, "Low-power radio-frequency ICs for portable communications," pp. 43-98.
(Also in Proc. IEEE, April 1995.)
[3] C. Chien, Digital Radio Systems on a Chip, A Systems Approach. Kluwer 2001.
[4] S. Sheng, R. Brodersen, Low-Power CMOS Wireless Communications - A Wideband CDMA
System Design. Kluwer 1998.
[5] B. Razavi, "Architecture and circuits for RF CMOS receivers," in Proc. IEEE 1998 Custom
Integrated Circuits Conference, pp. 393-400.
[6] B. Razavi, "Challenges in portable RF transceiver design," IEEE Circuits and Devices
Magazine, Sept. 1996, pp. 12-25.
[7] B. Razavi, "Design considerations for direct-conversion receivers, " IEEE Trans Circuits Syst. II, vol. 44, pp. 428-435, June 1997.
[8] J. Crols, M. Steyaert, CMOS Wireless Transceiver Design. Kluwer 1997.
[9] J. Crols, M.S. Steyaert, "Low-IF topologies for high-performance analog front ends of fully
integrated receievrs," IEEE Trans Circuits Syst. -II, vol. 45, pp. 269-282, March 1998.
[10] A. Loke, F. Ali, "Direct conversion radio for digital mobile phones - Design issues, status, and
trends," IEEE Trans. Microwave Theory and Techniques, vol 50, no. 11, pp. 2422-2435, Nov.
2002.
[11] P.-I. Mak, S.-P U, and R. P. Martins, “Transceiver architecture selection: Review, State-of-theArt Survey and Case Study”, IEEE Circuits and Systems Magazine, vol. 7, 2nd quarter 2007, pp. 625.