LED Lighting Illuminates
Buck Regulator Design
By John Betten, Applications Engineer, and Robert Kollman,
Senior Applications Manager, Texas Instruments, Dallas
When driving high-brightness LEDs with switching regulators, LED requirements will influence
many details of regulator design including the
methods used for closing the feedback loop and
for LED dimming.
A
s their production costs fall, LEDs are being
used more often in applications ranging from
handheld devices to automotive and architectural lighting. Their high reliability (operational
lifetimes of greater than 50,000 hours), good
efficiency (greater than 120 Lumens/W) and nearly instantaneous response make them very attractive light sources.
LEDs produce light in as little as 5 ns, compared to the 200ms response time of an incandescent bulb. Consequently,
they have been embraced by the automotive industry in
brake lights.
Additionally, LEDs are increasingly being used as the
primary light source in digital light projectors (DLP) and
television applications, where they replace the white-arc lamp
and mechanical color-wheel assembly. In DLP applications,
LEDs are switched on and off at rapid rates, generating the
red, green and blue color components as needed.
Driving LEDs is not without challenges, however. A controlled brightness requires driving the LED with a constant
current, which must be maintained regardless of input voltage. This can be far more challenging than driving an incan-
descent bulb connected to a battery. A common approach is
the use of a buck regulator to provide the LED drive current
from a dc input voltage. However, specific LED performance
requirements dictate several aspects of switching regulator
design, including the method used to close the feedback loop
and the choice of dimming technique.
LED I-V Characteristics
An LED has a forward I-V characteristic curve shape
similar to a diode. Below the LED turn-on threshold, which
is approximately 3.5 V for a white LED, very little current
flows through it. Above that threshold, current flow increases
exponentially for each small incremental increase in forward
voltage. This allows the LED to be modeled in SPICE as a
voltage source with a series resistance, but with one caveat:
The model is valid only at a single operating dc current.
If the dc current in the LED is changed, then the resistance of the model should be changed to reflect the new
operating current. Fig. 1 shows the measured impedance of
a 1-W white LED. The graph in Fig. 1 displays the change
in forward-voltage drop divided by the change in current.
This is the slope of the LED’s I-V curve, which effectively
represents the LED’s dynamic impedance for a specific drive
current. Note that a 1-W LED illuminates at currents as low
as 1 mA, although not very brightly.
Additionally, at large forward currents, the LED operates
at a high-power level that begins to heat the device. This increases the forward-voltage drop and, therefore, the dynamic
impedance. It is critical to consider the thermal environment
once the LED impedance has been determined.
Ripple current in the LED can increase its power dissipation, leading to increased junction temperatures. This
increase in temperature has a major impact on the life of
the LED. Fig. 2 shows the relative light output from an LED
as functions of operating time and junction temperature. If
we establish an 80% limit on light output as the useful life
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Fig. 1. This dynamic impedance measurement for a 1-W LED reflects the
series-resistor/voltage-source model for the device.
Power Electronics Technology October 2007
38
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Fig. 2. Minor reductions in junction temperature greatly extend the
operating life of a high-brightness white LED.
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of the LED, the lifetime is extended from about 10,000 hours
at 74°C to 25,000 hours at 63°C.
When an LED is driven by a buck regulator, the LED often
conducts the ac output-ripple current in addition to the dc
current, depending on the output filter arrangement chosen.
Fig. 3 quantifies the increased LED power dissipation due
to the ripple-current content. Since the ripple frequency is
high compared to the thermal time constant of the LED, the
high peak power dissipation due to the high ripple current
does not instantaneously impact the junction temperature.
Instead, the junction temperature is determined by the
average power.
Much of the LED’s voltage drop is like a voltage source.
Even at large ripple currents, there is no significant impact
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Fig. 3. While average forward current is the greatest contributor to heat
generation in an LED, ripple current becomes significant beyond peakto-average ratios of 50%.
on power dissipation (Fig. 3). For instance, 50% ripple current (IPK-PK = IOUTMAX) adds less than 10% to the total power
loss. Much above this level, ac ripple current from the supply
must be reduced to limit junction temperatures in order to
extend the semiconductor’s rated useful life. This is because
there is a resistive component to the voltage drop, and the
total power dissipated by the LED is determined by:
PLED = RLED  ILEDRMS + VLEDFORWARD  ILEDAVG.
The power dissipation can be used together with the total
thermal resistance and ambient temperature to calculate the
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LED LIGHTING
junction temperature of the LED. A useful rule of thumb is
that the semiconductor’s rated useful life doubles for every
10°C the junction temperature is reduced.
Also, most designs tend toward much lower ripple currents because of inductor constraints. Most inductors are
designed with a much lower ripple-current ratio (IPK-PK/IOUT
< 20%). Additionally, peak current in the LED should not
exceed the manufacturer’s specified maximum safe operational rating.
respective control characteristics. The switching action of
the buck power FET and diode was modeled as a voltagecontrolled voltage source having a gain of 10, and the LEDs
were modeled as a 3- resistor in series with a 6-V source.
Between the LEDs and ground, a 1- resistor was added to
sense the current. Fig. 5 shows the results.
In circuit A, the response is that of a first-order system,
which is inherently stable. The dc gain is set by the voltagecontrolled voltage source, the divider formed by the LED
resistance and the current-sense resistor. The pole of the
system is set by the output inductor and the series resistances
in the circuit. The compensator design is straightforward,
using a type-two amplifier.
Circuit B has a second-order response caused by the
presence of the output capacitor. This capacitor might be
required if a significant amount of LED ripple current was
unacceptable either due to electromagnetic interference
(EMI) or heating concerns. The dc gain is the same as the
first circuit; however, there is a pair of complex poles at the
resonant frequency of the output inductor and capacitor.
The filter’s total phase shift is 180 degrees, which could lead
to an unstable system if care is not taken when designing the
Closing the Control Loop
Fortunately, the process of closing the current loop on
an LED supply to meet these restrictions can be simpler
than closing a voltage loop on a conventional power supply.
This is because loop complexity can be controlled by the
designer and is determined by the output filter configuration. Three possibilities for these configurations are shown
in Fig. 4. These configurations are a simple inductor-only
filter (A), a typical power-supply filter (B) and a modified
filter (C) design.
A simple PSPICE model was built for each of the three
configurations to illustrate the differences among their
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Fig. 4. Potential output-filter configurations provide various options for speed and loop-compensation simplicity.
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Fig. 5. The gain and phase plots for the filter designs in Fig. 4 show that circuit C has high stability similar to that of circuit A, while providing the
high gain found in circuit B.
Power Electronics Technology October 2007
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LED LIGHTING
compensation circuit. However, the compensation circuit
design is similar to a conventional voltage-mode power supply and requires a type-three amplifier. Compared to circuit
A, circuit B has three additional components, including the
output capacitor.
In circuit C, the output capacitor has been repositioned
to make the circuit easier to compensate. The ripple voltage
across the LEDs is similar to circuit B. However, the inductor
ripple current flows through the current-sense resistor R105
in circuit C. This must be accounted for when calculating
the power dissipation. Circuit C has one zero and a pair of
poles, and is nearly as easy to compensate as circuit A. It has
the same dc gain as the first two circuits.
The zero is introduced by the capacitor and the LED series
resistance. One of the poles is set by the output capacitor and
current-sense resistor. The other pole is set by the currentsense resistor and output inductor. At high frequencies, the
response is the same as circuit A in Fig. 5.
Dimming
Quite often, LED applications require a dimming capability. For instance, it may be desirable to dim a display or
dim architectural lights. There are two ways to accomplish
this. One way is to reduce the LED current, which reduces
the intensity of the emitted light. The other way is to quickly
turn the LED on and off, which is perceived as a steady but
dimmer light compared to an LED that is always on. Furthermore, changing the ratio of on time to the total switching-cycle time, referred to as the duty cycle, produces linear
changes in the apparent light intensity. This technique is
known as pulse-width modulation (PWM).
Between these two methods, the least-effective way is
reducing the current, because the emitted light intensity is
not completely linear with current. Additionally, the LED
color spectrum tends to shift at currents below full rating.
Furthermore, human perception of brightness is exponential, so dimming can require the LED current to change by
large percentages. The potential impact of this relationship
between drive current and emitted light intensity on circuit
design is obvious.
For example, a given magnitude for ripple in the LED
drive current may only cause a 3% regulation error when
the LED drive current is 100% of full scale. However, this
same ripple-current magnitude produces a regulation error
of 30% when the LED drive current is reduced to 10% of
full scale.
Therefore, dimming the current through PWM is more
accurate, but it must be performed at switching speeds that
exceed the response time of the human eye in order to work
properly. Specifically in lighting and display applications,
PWM switching should be greater than 100 Hz so the human
eye does not perceive flicker.
To support this high-speed operation, the power supply
driving the LED must also operate at high frequencies. For
example, a 10% duty cycle in the millisecond range requires
the power supply to have a bandwidth greater than 10 kHz.
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Power Electronics Technology October 2007
LED LIGHTING
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Fig. 6. In this circuit, Q1 dims the LED by diverting current and also allows the control loop to maintain continuous operation.
Fig. 6 provides an example of a buck power stage with PWM
dimming using a filter stage similar to that shown in circuit
A of Fig. 4. In this circuit, the LED is simply switched in and
out of the circuit by FET Q1. In this manner, the control
loop is always active, resulting in an extremely fast transient
response (Fig. 7).
An alternate dimming method uses the controller’s enable function, or soft-start capacitor, to turn the controller
on and off. The current waveforms of a circuit using this
method are shown 9_07
in Fig. 9/27/07
8. Cycling 2:30
the controller
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Although this approach is somewhat slow when compared to
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This method is also efficient, because all output current is
delivered to the LED, or an LED string, during the on time,
and the circuit is shut down during the off time.
Once commanded to turn on, the controller has an
inherent startup delay before current begins to flow. The
controller’s internally programmed rise time, or the external
soft-start capacitor value on some controllers, sets the loadcurrent rise time. This approach will have a minimum dutycycle limit that is partially determined by the controller, the
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Fig. 8. Using the soft-start feature of a power supply to perform LED
drive current switching is a slower method than using a FET to short
the LEDs, but it is also more efficient.
Fig. 7. The extremely fast switching speed of an LED in a switching converter can be harnessed by a shorting FET that performs the function
of high-frequency current gating through the LED.
soft-start capacitor value and the output-filter values.
Often, this limit on the minimum duty cycle can be as
high as 10% to 30%, and this range of operation can suffer
from nonlinear brightness control. This may be unacceptable
for certain lighting applications where brightness gradients
are critical such as televisions, but may be usable for lessdemanding applications such as automotive taillights.
Of course, the final selection for the LED-dimming scheme
is determined not by the application, but by the design en-
gineer, who must weigh all design goals. If high-frequency
operation is desired, placing a FET in parallel with the LEDs
provides high-speed dimming, but this method is expensive
and introduces switching losses. Alternatively, the controller’s
enable pin and soft-start capacitor can be used to perform the
dimming function with greater efficiency, but this method
is relatively slow. Additionally, regardless of the dimming
scheme selected, the final output-filter selection of an LED
driver also impacts size, cost and EMI susceptibility. PETech
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