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HOW TO DRIVE MOSFETs AND IGBTs INTO
THE 21ST CENTURY
By
Mr. Abhijit D. Pathak and Mr. Ralph E. Locher,
IXYS Corporation
Santa Clara, CA 95054
ABSTRACT
As the industry pushes for higher power levels
and higher switching frequencies, power
supplies, which use MOSFETs/IGBTs for power
conversion, are being designed to their ultimate
attainable efficiency, smallest footprint, size and
weight. The subject of driving MOSFETs/IGBTs
to their highest possible frequencies at the
highest possible power levels, still providing
protection, is multi-dimensional and requires
knowledge of MOSFETs and IGBTs as well as
circuit theory and how it applies to this discipline.
Many topologies in high voltage converter,
inverter and matrix converter applications
require special techniques to drive MOSFETs
and IGBTs. The paper deals with this subject
and explains with examples how to face the
challenges of tomorrow today.
1. INTRODUCTION
Modern Power Electronics makes liberal use of
MOSFETs and IGBTs in many applications and,
if the present trend is any indication, future will
see more and more applications making use of
MOSFETs and IGBTs. Although sufficient
literature is available on characteristics of
MOSFETs and IGBTs, practical aspects of
driving them in specific circuit configurations at
different power levels and at different
frequencies require that design engineers pay
attention to a number of aspects.
1.1 MOSFET AND IGBT TECHNOLOGY
Due to the absence of minority carrier transport,
MOSFETs can be switched at much higher
frequencies. The limit on this is imposed by two
factors: transit time of electrons across the drift
region and the time required to charge and
discharge the input Gate and ‘Miller’
capacitances.
IGBT derives its advantages from MOSFET and
BJT. It operates as a MOSFET with an injecting
region on its Drain side to provide for conductivity
modulation of the Drain drift region so that on-state
losses are reduced, especially when compared to an
equally rated high voltage MOSFET. Advances and
breakthroughs continue to improve performances of
both devices. Lower parasitic capacitances, lower
RDS (on), RGint , Qg, Rthjc and faster switching times
are being achieved in newer MOSFETs. Third
generation IGBTs have lower forward voltage drop,
lower gate Charge, lower trr of inverse parallel diode,
shorter and lesser tail current, lower tf , smaller Eon
and Eoff, improved current sharing amongst parallel
devices and better Rthjc .
1.2 POWER LOSSES IN DRIVERS AND DRIVEN
MOSFET/IGBT
For determining the power loss in a Driver while
driving a power MOSFET, the best way is to refer to
the Gate Charge Qg vs. V GS curve for different
values of V DS(off).
PGATE=VCC*Qg*fsw
Eq. 1.1
Wherein, Vcc is the Driver’s supply voltage, Qg is
the total Gate Charge of the MOSFET being driven
and fsw is the switching frequency. It is prudent then
to choose a MOSFET with lower value of Qg and it
is here that of low Gate Charge MOSFETs are
preferred because they as well as the drivers incur
lower losses.
As far as switching losses in a MOSFET are
concerned, there are some short time-intervals,
during which finite VDS and finite ID coexist, albeit
momentarily. When this happens during turn-on, the
actual integration:
∫
VDS(t) ID(t) dt
Eq. 1.2
Is defined as Turn-On switching energy loss.
Likewise, during turn-off, when finite values of ID and
VDS coexist, integration of:
∫
VDS(t) ID(t) dt
Eq.1.3
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Is called Turn-off switching energy loss in a
MOSFET. Amongst the responsible parameters
determining these switching energy losses, Ciss,
Coss and Crss affect the turn-on and turn-off
delays as well as turn-on and turn-off times.
For an IGBT, it would be similarly shown that:
∫
VCE(t)IC(t)dt
Eq. 1.4
This equation represents switching energy loss.
Needless to emphasize that the time interval for
these integrals would be the appropriate time
during which finite values of ID and VDS or VCE
and IC coexist in a MOSFET or IGBT
respectively. Thus average switching energy lost
in the device can be computed as follows:
MOSFET: Ps = 1/2*VDS *ID *fsw*(t on+t off ) Eq.1.5
IGBT:
Ps = 1/2*VCE *IC *fsw *(t on+t off ) Eq.1.6
Main emphasis in modern Power Electronics is
on reducing total losses dissipated in devices
and subsystems for achieving higher operating
efficiency and more compact designs, reducing
volume and weight of resultant systems. Thus,
operation at higher and higher switching
frequencies is now a necessity, and as a result,
switching losses predominate in power-lossbudget in semiconductor switches. Reducing
switching losses then becomes the single most
crucial goal. Keeping this goal in mind, drive
circuits should be so designed as to yield ultra
fast rise (t r) and fall times (t f ). Assuming sum of
tr and tf be no more than 2% of the PWM period
in hard switching applications, a Table:1 is
prepared of 1000 Volts rated MOSFETs of three
types: standard, Low Gate Charge types and FClass, which are optimised for megahertz
switching. Table: 1 shows, for a given fsw, peak
current required to drive it, recommended
external gate resistor, total power loss in
driver+gate resistor and the driver IC used - All
for SMPS designs of 500W, 1KW and 2 KW
ratings. While standard MOSFETs are optimally
usable up to about 400 kHz, Low Gate Charge
MOSFETs give adequate performance up to 800
kHz in hard switching SMPS applications.
Designing above 800 kHz in hard switching
mode, the levels of RFI/EMI noise and switching
losses are excessive and hence soft switching
(resonant mode, in which sum of tr and tf should
be no more than 10% of the total one cycle
PWM period) is preferred. The third section of
rows in Table:1, shows how F-Class MOSFETs
perform with the same driver ICs at 1MHz, 2 MHz
and 4 MHz in resonant switching mode.
Both, in hard switching and in resonant switching
modes, trying to achieve values of tr and tf less that
what are assumed tend to generate higher electrical
noise and oscillations in drain current, while
increasing them will tend to reduce overall efficiency.
One can appreciate that energy lost in gate drive
system, say, @ fsw =100 kHz for a 500 W SMPS, is
only 0.04% - a negligible value.
Table: 2 and Table: 3 show how a full range of driver
ICs with peak current ratings from 2 Amps to 30
Amps can be used for a full range of MOSFETs with
VDS rating of 60 Volts to 1000 Volts and ID rating of
24 Amps to 340 Amps at different fsw =50 KHz, 100
KHz, 200 KHz, 400KHz and 800 KHz. Please note
that * indicates that the driver IC (in TO-263 and/or
TO-220 package) will need a heat sink. The
availability of a 30 Amp driver IC opens doors to
great many possibilities.
It is very important to understand that when 30
Amps peak current is required to drive a
MOSFET/IGBT, it is never prudent to parallel two 14
Amp driver ICs, as this tends to allow momentary
shoot through current from P-Channel MOSFET
(Totem pole output stage) of one driver into NChannel MOSFET (Totem pole output stage) of
another driver due to mismatch in propagation
delays between two drivers. Thus the importance of
having a 30 Amps driver IC can’t be
overemphasized.
1.3 ADVANCES IN MOSFETs AND IGBTs
Extending the discussion further, we know that
advances in Power MOSFET technology are giving
dividends in the form of lower R DS(on) (that gives
lower conduction losses), lower RGint (that improves
switching speed), lower Qg (that improves dynamic
performance and requires less power from driver),
lower transient thermal impedance (enabling higher
power dissipation) and lower Cissand lower gate to
drain feedback capacitance Crss and lower rise and
fall times (enabling operation at still higher switching
frequency).
Advances in IGBT technology centre around
improving NPT technology by ‘soft punch through’
techniques, called the Third Generation NPT
Technology. The dividends from such advances are
multi-fold: lower cost, 20% lower on-state and 20%
lower switching losses, better parallel current
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sharing, lower thermal impedance, increased
ruggedness
to
overloads,
lower
input
capacitance and faster yet smother turn-on and
turn-off waveforms. It looks like a true win-win
situation! Another path chosen by some
manufacturers is Trench IGBTs, which also have
improved VCE(sat) and switching losses.
current still higher. It allows one to choose different
Turn-On and Turn-Off times by choosing different
values of Rgon and Rgoff . It allows for incorporating –
ve bias for reasons explained above. A pair of 18 V
Zener diodes with their cathodes connected together
protects the Gate-Emitter Junction of IGBT from
voltage spikes.
2. TYPES OF DRIVERS
2.3 TECHNIQUES TO GENERATE –Ve BIAS
DURING TURN-OFF
2.1 IC DRIVERS
Although there are many ways to drive
MOSFET/IGBTs using hard-wired electronic
circuits, IC Drivers offer convenience and
features that attract designers. The foremost
advantage
is
compactness.
IC
Drivers
intrinsically offer lower propagation delay. As all
important parameters are specified in an IC
Driver, designers need not go through time
consuming process of defining, designing and
testing circuits to drive MOSFET/IGBTs. Another
advantage is repeatability and predictability of
performance, which can’t be easily achieved in
hardwired driver circuits.
2.2 TECHNIQUES AVAILABLE TO BOOST
CURRENT OUTPUTS
Totem pole stage with N-Channel and PChannel MOSFETs can be used to boost the
output from an IC Driver. The disadvantage is
that the signal is inverted and also there exists
momentary shoot through when common gate
voltage is in transition.
Totem pole arrangement using matched NPNPNP transistors; on the other hand, offer many
advantages, while boosting the output currents
from IC Drivers. Shoot through phenomenon is
absent in this case. The pair of transistors
protects each other’s base-emitter junctions and
handle current surges quite well. One such
arrangement is shown in Fig. (10). Here Q1 is a
NPN transistor, while Q2 is a matched PNP
transistor with appropriate collector current
rating and switching speed to satisfy Drive
requirement for the High Power IGBT. Another
feature added is –ve bias for guaranteed fast
switch-off even in electrically noisy environment.
This is done, by utilizing power supply with +15
and –5 Volts output, whose common ground is
connected to the IGBT emitter.
The arrangement shown in Fig. (10) does a few
more things in addition to boosting the output
The importance of –ve bias during turn-off for
practically all semiconductor switches cannot be
overemphasized, as one may recall from the days of
bipolar transistors. -Ve bias helps to quickly remove
any charge on the C GS and CGD in the case of
MOSFETs
and
IGBTs,
thus
accelerating the turn-off mechanism.
considerably
It is important to understand that turn-on speed of a
MOSFET or IGBT can be increased only up to a
level matched by the reverse recovery of rectifiers or
diodes in a power supply, because in an inductive
clamped load (most common), turn-on of a MOSFET
or IGBT coincides with turn-off (or reverse recovery
completion) of the rectifier diode. Any turn-on faster
than this does not help. Too fast a turn-on could also
cause oscillation in the Drain or Collector current.
However, it is always beneficial to have a Driver with
intrinsic low turn-on time and then be able to tailor
this with a series gate resistor. Turn-off
phenomenon, on the other hand, does not have to
wait for any other component in the subsystem. It is
here that any enhancement technique can best be
utilized. Although many IC drivers themselves
feature extremely low turn-on and turn-off times,
arrangement to provide –ve bias during turn-off
helps still faster turn-off and prevents false turn on
even in electrically noisy environment.
Fig.(10) demonstrates one way of generating –ve
bias during turn-off. Fig. (14) shows how to generate
–ve bias in a transformer coupled Drive circuit
arrangement. Here Zener diode can be chosen of
appropriate voltage for giving that much –ve bias
(plus one diode drop) during turn-off. Another unique
feature of circuit in Fig. (14) is its ability to maintain
exact pulse wave shape across Gate and source. In
Fig. (12) a method of using isolated DC to DC
converter with outputs of +15 and –5 V is used to
power IXD_414, while by connecting isolated ground
of this DC to DC Converter to the emitter of the IGBT
being driven, –5 V of –ve bias during turn-off is
ensured.
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2.4
NEED
PROTECTION
FOR
UNDERVOLTAGE
In Transfer Characteristics (I D vs. VGS ) of a
MOSFET, one can see that for values of VGS
below VGS(th), the drain current is negligible, but
in this vicinity, the device is in its linear (Ohmic)
region and concurrent application of large values
of V DS could cause considerable amount of
localized heating of the junction. In short, when
a MOSFET is being used as a switch, any
operation in its linear region could cause
overheating or device failure. Bringing the
MOSFET quickly into its saturation from its offstate is the Driver’s job. And, if Vcc is below the
minimum required value, linear operation can
ensue to the detriment of MOSFET. I hasten to
add, however, that most PWM ICs, controller
ICs and microcomputer ICs have this protection
feature built-in and, if sharing the same Vcc bus,
the Driver IC gets the benefit of this function
being implemented elsewhere in the subsystem.
Nevertheless, Driver ICs having this feature is
an added benefit. The need of UV protection
applies equally well to IGBTs.
2.5OVERLOAD/SHORT CIRCUIT ROTECTION
Any operation of MOSFET/IGBT outside the
Safe Operating Area (SOA) could cause
overheating and eventual device failure and
should be prevented by an electronic active
monitoring and corrective arrangement.
Load or current sensing could be done by either
a Hall Effect Sensor or by a Shunt resistor in
series with source/emitter terminal. The voltage
picked up, which is proportional to current, is low
pass filtered and then compared to a preset
limit. The comparator output could initiate turnoff of MOSFET/IGBT. A circuit to detect
overload/short circuit is shown in Fig. (12),
where the output FAULT will go low when it
occurs.
All IXDD series of Drivers have an ENABLE pin,
which, when driven low, say, by the FAULT
output from this comparator, puts the final NChannel and P-channel MOSFETs of the IXDD
Driver in its TRISTATE mode. This not only
stops any output from the Driver, but also
provides an environment for implementing soft
turn-off. There are two ways of doing this. Just
by connecting a resistor of appropriate value
from Gate to source/emitter, the CGS gets
discharged through this resistor and, depending on
the value of the connected resistor, soft turn-off of
any duration can be achieved. Another way, as
shown in the Fig. (2), is to use a signal MOSFET Q1
to pull down the Gate, when short circuit is detected.
The resistor in series with this signal MOSFET
determines the time duration of this soft turn-off. Soft
turn-off helps protect IGBT/MOSFET from any
voltage transients generated due to L*dIC/dt (or
L*dID/ dt) that could otherwise bring about avalanche
breakdown. The PC board layout and Bill of
Materials for this circuit are shown in Fig. (3).
For an IGBT, Desat detection (Desat = Desaturation
of forward voltage drop) is a method used for short
circuit/overload
protection.
When
short
circuit/overload occurs, the forward voltage drop of
the IGBT (V CE) rises to disproportionately high
values. One must ignore the initial turn-on rise in
VCE, when output from Driver has still not risen to
high enough value. Nevertheless, when V CE rises to
a level of, say, 7 Volts, in presence of sufficient Gate
Drive voltage, it means the collector current IC has
risen to a disproportionately high value, signalling
overload. When a voltage level higher than 6.5 Volts
is detected, Gate signal can be softly turned off,
resulting into soft turn-off of the IGBT. Fig. (12)
shows how Desat feature can be wired into a total
Driver Circuit, using also other features, such as
Opto-isolation and –ve turn-off bias.
3. HIGH SIDE DRIVING TECHNIQUES
3.1
EMPLOYING
CHARGE-PUMP
BOOTSTRAP METHODS
AND
For driving the upper MOSFET/IGBT in a phase leg
employed in a bridge topology, a buck converter or a
2-transitor forward converter, low side drivers cannot
be used directly. This is because the source/emitter
of upper MOSFET/IGBT is not sitting at ground
potential.
Fig. (5) shows how a charge pump creates a higher
Vcc to be used for the driver IC for the Upper
MOSFET/IGBT. Here the pair of N-Channel and PChannel MOSFETs acts as switches, alternately
connecting incoming supply voltage to output
through capacitors and Schottky diodes, isolating it
and almost doubling it. Switching frequency in
several hundred Kilohertz is used and, therefore, low
ripple isolated output voltage is available as DC
Supply for the Driver of Upper MOSFET/IGBT. Fig.
(7) illustrates how one IXD_404 can be used as
charge pump, delivering 350 mA, and one IXD_408
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as a Driver giving +/- 8 Amps, in conjunction with
IXBD4410 and IXBD4411, for driving a phase
leg of two IXFX50N50 MOSFETs. Fig. (8) shows
how a charge pump delivering as much as 500
mA can be constructed using one IXD_404; and
by utilizing one IXD_414, one can boost the
output from IXBD4410 and IXBD4411 to +/14Amps for driving Size 9 high power MOSFETs
and IGBTs or even MOSFET/IGBT module.
Another method is the Bootstrap Technique as
shown in Fig. (6). The basic bootstrap building
elements are the level shift circuit, bootstrap
diode DB, level shift transistor Q1, bootstrap
capacitor CB and IXD_408 or IXD_414. The
bootstrap capacitor, IXD_408/IXD_414 driver
and the gate resistor are the floating, sourcereferenced parts of the bootstrap arrangement.
The disadvantages of this technique are longer
turn-on and turn-off delays and 100% duty cycle
is not possible. Additionally the driver has to
overcome the load impedance and negative
voltage present at the source of the device
during turn-off.
3.2 ACHIEVING GALVANIC ISOLATION BY
USING OPTO-COUPLERS TO DRIVE UPPER
MOSFET/IGBT
For driving high side MOSFET/IGBT in any
topology, opto-couplers can be used with
following advantages:
1. They can be used to give a very high isolation
voltage; 2500 to 5000 Volts of isolation is
achievable by use of properly certified optocouplers.
2. Signals from DC to several MHz can be
handled by opto-couplers.
3. They can be easily interfaced to
Microcomputers, DSPs or other controller ICs
or any PWM IC.
One disadvantage is that the opto-coupler adds
its
own
propagation
delay.
Another
disadvantage of using an opto-coupler is that
separate isolated power supply is required to
feed the output side of the opto-coupler and the
driver connected to it. However, isolated DC-toDC Converters with few thousand Volts of
isolation are readily available. These can be
used to supply isolated and regulated +ve 15 V
and –ve 5V to the output side of the optocoupler and the Driver IC for driving Upper
MOSFET/IGBT as is shown in Fig. (12) and Fig.
(13). As can be seen, identical chain of optocoupler, Driver and DC-to-DC Converters are
used for even lower IGBTs. This is to guarantee
identical propagation delays for all signals so that
their arrival time at the gate of IGBT bears the same
phase relationships with one another as when they
originated in the DSP.
3.3 USE OF TRANSFORMERS TO OBTAIN
GALVANIC ISOLATION IN DRIVING UPPER
MOSFET/IGBT
Using transformers to achieve galvanic isolation is a
frequently used technique. Depending on the range
of frequencies being handled and power rating
(voltage and current ratings and ratios), transformers
can be designed to be quite efficient. The gate drive
transformer carries very small average power but
delivers high peak currents at turn-on and turn-off of
MOSFET/IGBT.
While designing or choosing a Gate Drive
transformer, the following points should be kept in
mind:
1. Average power being handled by the transformer
should be used as a design guideline. Margin of
safety should be taken into account, keeping in
mind maximum volt-second product and allowing
for worst-case transients with maximum duty ratio
and maximum input voltage possible.
2. Employing bifilar winding techniques to eliminate
any net DC current in any winding. This is to avoid
core saturation.
3. If operation in any one quadrant of B-H loop is
chosen, care should be taken for resetting the
core.
Advantages of employing transformers for Gate
Drive are:
1. There is no need for any isolated DC-to-DC
Converter for driving an upper MOSFET/IGBT .
2. There is practically no propagation delay in a
transformer to carry signals from primary side to
the secondary side.
3. Several thousand volts of isolation can be built in
between windings by proper design and layouts.
The disadvantages of using transformers for Gate
Drive are:
1. They can be used only for time varying signals.
2. It is difficult to implement DESAT protection
feature.
Two examples of gate Drive circuits, using
transformers follow. In Fig. (11), a phase shift
controller outputs its signals to the IXD_404 Dual
Drivers, which in-turn, feed the transformers. The
secondary windings of these transformers are
coupled to the Gates of upper and lower MOSFETs
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in an “H” Bridge topology. Fig. (14) shows
another transformer coupled Gate Drive circuit
employing DC restore technique to maintain
same wave shape of original signal with added
feature of -ve bias offered using a Zener in
series with a fast diode across secondary.
4.0 DESIRABLE SPECIFICATIONS AND
FEATURES OF ANY LOW AND HIGH SIDE
DRIVER PAIRS, SUCH AS IXBD4410 AND
IXBD4411
1. Under voltage and over voltage lockout
protection for Vcc ;
2. dV /dt immunity of greater than ± 50 V/ns;
3. Galvanic isolation of 1200 Volts (or greater)
between
low side and high side;
4. On-chip negative gate-drive supply to ensure
MOSFET/IGBT turn-off even in electrically
noisy environments;
5. CMOS/HCMOS compatible inputs with
hysteresis;
6. < 20 ns rise and fall times with 1000 pf load
and <100 ns rise and fall times with 10000 pf
load;
7. <100 ns of propagation delay;
8. >2 Ampere peak output Drive Capability;
9. Automatic shutdown of output in response to
over current and/or short -circuit;
10. Protection against cross conduction between
upper and lower MOSFET/IGBT;
11. Logic compatible fault indication from both
low and high-side drivers.
A suggested wiring diagram, making use of such
a high/low side driver pair is shown in Fig. (4).
This diagram is for a phase leg of MOSFETs or
IGBTs. Likewise, the wiring diagram is to be
repeated for each phase leg and hence one
needs two such circuits for “H” Bridge and three
such circuits for 3-Phase Bridge topologies
As can be seen in this schematic, to obtain
galvanic isolation, it uses one ferrite core
transformer for sending drive signals to
IXBD4411 and another ferrite core transformer
for receiving fault and status signals from
IXBD4411.
T1
represents
both
these
transformers housed in one IC type package. To
avoid saturation, capacitors are placed in series
with each primary winding to which PWM pulse
train is transmitted. 1200 Volts of isolation
barrier is built in. Note the use of high voltage
fast diode with required voltage rating to feed
Vcc to upper driver.
Higher current MOSFET/IGBTs require higher drive
currents, especially for operating them at high
switching frequencies. For these applications, one
can use IXD_408 or IXD_414, either in stand-alone
mode or in conjunction with IXBD4410 and
IXBD4411. It is easy to realize now that one can
easily get all the facilities of feature-rich IXBD4410
and IXBD4411 and when higher drive current
capability is called for, use them in conjunction with
IXD_408 or IXD_409 or IXD_414. In case of a lowside MOSFET/IGBT, it is simple to use IXD_408 or
IXD_414 alone. One such example is given in Fig.
(7).
For driving upper MOSFET/IGBT of a phase leg, one
of the approaches is to employ a charge pump. Two
such application circuit schematics are shown in Fig.
(7) and Fig. (8). In Fig. (7) output from
IXBD4410/4411 is boosted up to ± 8 Amps using
IXD_408 and the charge pump output is boosted to
350 mA, using one driver of IXD_404 for driving
IXFK48N50 (rated at Id=48 Amps and Vd = 500
Volts). In Fig. (8), the output from IXBD4410/4411 is
boosted up to ±14 Amps by IXD_414 to drive a Size
9 MOSFET IXFN80N50 (rated at Id = 80 Amps and
Vd = 500 Volts). IXD_404 can still adequately
provide 500 mA for the charge pump circuit.
4.2 GENERAL REMARKS ABOUT MOSFET/IGBT
IC DRIVERS
The most important strength of MOSFET/IGBT IC
Drivers should be their ability to provide high
currents needed to adequately drive today’s and
tomorrow’s large size MOSFETs and IGBTs.
Another important feature is that there should be no
cross conduction in the output stage of the driver IC,
thus saving transition power dissipation.
In addition, all these Drivers incorporate a unique
facility to disable output by using the ENABLE pin.
With the exception of the IXD_408, the ENABLE pin
is tied high internally. When this pin is driven LOW in
response to detecting an abnormal load current, the
Driver output enters its Tristate (High Impedance
State) mode and a soft turn-off of MOSFET/IGBT
can be achieved. This helps prevent damage that
could occur to the MOSFET/IGBT due to Ldi/dt over
voltage transient, if it were to be switched off
abruptly, “L” representing total inductance in series
with Drain or Collector. A suggested circuit to
accomplish this soft turn off upon detecting overload
or short circuit is shown in Fig. (2). It is also possible
to do this by an independent short circuit/overload
detect circuit, which could be a part of the PWM or
other controller IC. All one needs to do is to take
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output signal (FAULT) from such a circuit and
feed it into the ENABLE pin of Driver. A resistor
R P connected across Gate and Source or Gate
and Emitter (as the case may be) would ensure
soft turn-off of the MOSFET/IGBT, turn-off time
being equal to RP(C GS +CGD)
4.2.1 LOW CURRENT DUAL IC DRIVERS
Fig. (11) shows an interesting application for a 4
Amp Dual driver IC in a Phase Shift PWM
Controller application, in which galvanic isolation
is obtained by using Gate Drive Transformers.
Note that this Controller operates at a fixed
frequency. Turn-off enhancement is achieved by
using local PNP transistors in secondary of gate
drive transformer.
For a vast number of low and medium current
MOSFETs and IGBTs, 2 Amp to 4 Amp dual IC
drivers provide a convenient tool and circuit
simplicity, coupled with performance.
4.2.2 HIGH CURRENT IC DRIVERS
These are eminently suitable for driving higher
current MOSFETs and IGBTs and larger size
MOSFET/IGBT
Modules.
Many
circuit
schematics applying these in various topologies
are possible and some of these are shown in
different figures.
The 5 pin TO -263 surface mount version of
some IC drivers can be soldered directly on to a
copper pad on a printed circuit board for better
heat dissipation. It is possible then to use these
high current drivers for very high frequency
switching application, driving high current
MOSFET
modules
for
a
high
power
converter/inverter.
Fig. (1) shows a basic low side driver
configuration. C1 is used as a bypass capacitor
placed very close to pin No. 1 and 8 of the driver
IC. Fig. (9) shows a method to separately control
the turn on and turn off times of MOSFET/IGBT.
Turn-on time can be adjusted by Rgon, while the
turn-off time can be varied by Rgoff. Schotkky
diodes facilitate this, by virtue of their very low
forward voltage drop and low trr. The 18V,
400mW Zener diodes protect the Gate-Emitter
junction of the IGBT. A careful layout of the
PCB, making shortest possible length between
output pin and IGBT Gate, while providing
generous copper surface for a ground plane,
helps achieve fast turn-on and turn-off times without
creating oscillation in the Drain/Collector current.
Fig. (9) shows another arrangement and includes a
method for faster turn-off using a PNP transistor
placed very close to the MOSFET Gate and Source.
It is a good practice to tie the ENABLE pin of drivers
to Vcc through a 10K resistor. This ensures that the
driver always remains in its ENABLED mode except
when driven low due to a FAULT signal. Again, this
FAULT signal puts these two drivers into their
TRISTATE output mode.
Fig. (10) shows a method to boost output from driver
IC to a much higher level for driving very high power
IGBT module. Here the turn-on and turn-off times
can be varied by choosing different values of
resistors: Rgon and Rgoff. To provide –ve bias of 5
Volts, the IGBT emitter is grounded to the common
of +15V and –5V power supply, which feeds +15V
and –5 v to the IXD_408. Notice that the incoming
signals must also be level shifted.
Fig. (12) shows a high current driver IC driving one
IGBT of a Converter Brake Inverter (CBI) module.
Here all protection features are incorporated. For
High Temperature cut-off, a bridge circuit is used
with the CBI module’s thermistor. Comparator U3
compares voltage drop across the thermistor to the
stable voltage from the Zener diode. P1 can be used
to preset the cut-off point at which the comparator’s
output goes low. This is fed into the DSP as
OVERTEMP signal.
Short circuit protection is provided by continuously
monitoring the voltage drop across a SHUNT. Note
that one end of SHUNT is connected to the power
supply ground GND1. The voltage picked up from
this SHUNT is amplified by a low noise Op Amp and
is then compared to the stable voltage from the
same Zener. When short circuit occurs, the
comparator output (FAULT) goes low. 1% metal film
resistors are used throughout in both these circuits
to ensure precision and stability. C3 and C4 help in
offering low pass filtering to avoid nuisance tripping.
Principle of DESAT sensing for detecting overload
on an IGBT has been explained before in section 2.5
above. In the case of AC Motor Drive, each IGBT
has to be protected from overload using separate
DESAT sensing. Fig. (12) and Fig. (13) show the
connection for each IGBT. DESAT sensing is done
on the isolated side of each opto-coupler, while the
resultant FAULT signal is generated on the common
input side with respect to GND1. Each FAULT signal
is open collector type and hence can be tied
IXAN0009
together with other FAULT signals from other
opto-coupler or from other comparators. The
DSP will stop output drive signals when either
FAULT or OVERTEMP signal goes low. When
this happens, notice that IXD_414 offers a -ve
bias of -5V to guarantee turn-off conditions,
even in presence of electrical noise. -5V is
applied to gate of each IGBT during turn-off
even under normal operating conditions. After
fault is cleared, the DSP can issue a RESET
signal for resuming normal operation.
5.0 PRACTICAL CONSIDERATIONS
When designing and building driver circuits for
MOSFET/IGBT, several practical aspects have
to be taken care of to avoid unpleasant voltage
spikes, oscillation or ringing and false turn-on.
More often than not, these are a result of
improper or inadequate power supply bypassing,
layout and mismatch of driver to the driven
MOSFET/IGBT.
As we understand now, turning MOSFET/IGBT
on and off amounts to charging and discharging
large capacitive loads. Suppose we are trying to
charge a capacitive load of 10,000 pF from 0 to
15 VDC (assuming we are turning on a
MOSFET) in 25 ns, using a 14 amp ultra high
speed driver.
I = VxC / t
Eq. 5.1
-12
-9
I = (15-0)x10000x10 / 25x10 = 6 A
What this equation tells us is that current output
from driver is directly proportional to voltage
swing and/or load capacitance and inversely
proportional to rise time. Actually the charging
current would not be steady, but would peak
around 9.6 Amps, well within the capability of 14
Amp driver. However, driver IC will have to draw
this current from its power supply in just 25 ns.
The best way to guarantee this is by putting a
pair of by-pass capacitors (of at least 50 times
the load capacitance) of complementary
impedance curves in parallel, very close to the
VCC pin of the driver IC. These capacitors
should have the lowest possible ESR
(Equivalent Series Resistance) and ESL
(Equivalent Series Inductance). One good
example of this is high quality surface mount
type monolithic ceramic capacitors. Other
preferred type is SMD Tantalum. One must keep
the capacitor lead lengths to the bare minimum..
A smart way of accomplishing this is to solder
the capacitor across the Vcc and ground pin of driver
IC from the bottom (solder side).
Another very crucial point is proper grounding.
Drivers need a very low impedance path for current
return to ground, avoiding loops. The three paths for
returning current to ground are: 1. Between driver IC
and the logic driving it; 2. between the driver IC and
its own power supply; 3. between the driver IC and
the source/emitter of MOSFET/IGBT being driven.
All these paths should be extremely short in length
to reduce inductance and be as wide as possible to
reduce resistance. Also these ground paths should
be kept distinctly separate to avoid returning ground
current from the load to affect the logic line driving
the driver IC. A good method is to dedicate one
copper plane in a multilayered PCB to provide a
ground surface. All ground points in the circuit
should return to the same physical point to avoid
generating differential ground potentials.
With desired rise and fall times in the range of 25 to
50 ns, extreme care is required to keep lengths of
current carrying conductors to the bare minimum.
Since every inch of length adds approximately 20 nH
of inductance, a di/dt of 240 Amps/microsecond
(used in the example calculation for Eq. 5.1)
generates a transient LdI/dt voltage of 4.8 volts per
inch of wire length, which subtracts from the driver’s
output. The real effect will be a significant increase
in rise time for every tiny increase in conductor
length from output pin of driver to the Gate lead of
MOSFET/IGBT. For example, one extra inch of
conductor length could increase rise time from 20 ns
to 70 ns, in an ultra high-speed gate drive circuit.
Another detrimental effect of longer conductor length
is transmission line effect and resultant RFI/EMI.
This inductance could also resonate with parasitic
capacitances of MOSFET/IGBT, making it difficult to
obtain clean current waveforms at rise and fall.
It is important to also keep in mind the fact that
every MOSFET/IGBT has some inductance
depending on the package style and design. The
lower this value, the better is the switching
performance, as this inductance is, in effect, in
series with the source/emitter and the resulting
negative feedback increases switching times.
While applying driver IC for any application, it is also
necessary to compute power dissipated in the driver
for a worst-case scenario. The total power dissipated
in the driver IC is a sum of the following:
1. Capacitive load power dissipation;
2. Transition power dissipation;
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3. Quiescent power dissipation.
For all IXD_series of drivers, transition power
dissipation is absent due to a unique method
(Patent pending) to drive the output N-Channel
and
P-Channel
MOSFETs,
practically
eliminating cross conduction.
As
described
under
section
1.2,
a
MOSFET/IGBT driver incurs losses. Let us
derive formulae to compute this power loss in a
driver:
PD(on) = D x ROH x Vcc x Qg x fsw Eq. 5.2
ROH + RGext + RGint
PD(off) =(1-D)xROLxVccxQgxfsw
Eq.5.3
ROL + RGext+ RGint
where:
ROH = Output resistance of driver @ output
High
ROL = Output resistance of driver @ output
Low
fsw = Switching frequency
RGext = resistance kept externally in series
with
Gate of MOSFET/IGBT
RGint = Internal mesh resistance of MOSFET/
IGBT
D = Duty Cycle (value between 0.0 to 1.0)
Qg= Gate Charge of MOSFET/IGBT
Total loss PD = PD(on) + PD(off)
Note also that in general, RGint is small and can
be neglected and that ROH = ROL for all IXD_
drivers. Consequently, if the external turn-on
and turn-off gate resistors are identical, the total
driver power dissipation formula simplifies to:
PD=PD(on)+P D(off)= ROHxV cc xQgxfsw Eq.5.4
ROL + RGext
Let us review with some examples:
Assume that we are driving an IXFN200N07 for
a Telecom power supply application or for a
UPS/Inverter application at a switching
frequency of 20 kHz. Furthermore, RGext = 4.7
Ohms and gate supply voltage is 15V.
On page two of the IXD_409 Data sheet, we
read the value of ROH = 1.5 Ohms ( Maximum).
For Qg, refer to Data Sheet of the IXFN200N07
and go to Gate Charge vs. VGS curve and look
for value of Qg at Vcc = 15 V. You can read it as
640 nC. Substituting these values into Eq. 5.4 yields:
PD = 1.5 x 15 x 640 x 20,000 x 10-9
1.5 + 4.7
PD = 46.45 mW
Assuming an ambient of 50 o C in the vicinity of
IXD_409PI, the power dissipation capability of
IXD_409PI must be derated by 7.6mW/oC, which
works out to be 190 mW. The maximum allowable
power dissipation at this temperature becomes 975190=785 mW. However, as calculated above, we will
be dissipating only 46.45 mW so we are well within
the dissipation limit of 785 mW.
If one increases fsw to 500 kHz for a DC-to-DC
Converter application, keeping other parameters the
same as above, now the dissipation would be 1.16
W, which exceeds the specification for IXD_409PI.
So in this case, it is recommended to use either the
IXD_409YI (TO -263 package) or IXD_409CI (TO220) package. Both these packages can dissipate
about 17 W with proper heat sinking arrangement.
The TO-263 is a surface mount package and can be
soldered onto a large pad on a copper surface of a
PCB for achieving good heat transfer. For TO -220
package, a heat sink can be used.
Let us take another example of a boost converter,
using IXFK55N50 at VDS = 250 VDC and at ID =
27.5 Amps. Assuming sf w = 500 kHz, Vcc = 12 V.
From the curve of Gate Charge for IXFK55N50 in
the Data Sheet one can determine that Qg = 370 nC.
Let us set RGext = 1.0 Ohm. We use IXD_414YI or
IXD_414CI here, which can dissipate 12W. Here
typical value of ROH = ROL = 0.6 Ohm. Substituting
the above values in our equation, we compute the
power dissipation to be:
PD = 0.6 x 12 x 370 x 500 kHz x 10-9
0.6 + 1.0 + 0.0
PD = 0.83 W.
With adequate air circulation, one may just be able
to use the PDIP Package. However, it is
recommended to use TO -263 or TO -220 package
for reliable performance.
For the third example, considering driving a large
size MOSFET module VMO 580-02F at fsw = 250
kHz. Let Vcc = 10 V, ROH = ROL = 0.6 Ohm, RGext=
0.0 Ohm. We read that Qg =2750 nC at Vcc = 10 V
off the VMO 580-02F data sheet. Now:
PD = 0.6 x 10 x 2750 x 250kHz x 10 -9
0.6+ 0.0 + 0.0
PD = 6.86 W
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IXD_414YI (TO-263) or IXD_414CI (TO-220)
can easily drive this load provided adequate
heat sinking and proper airflow is maintained.
Comments above for mounting TO-263 and/or
TO-220 packages apply here as well. For
derating use 0.1 W/o C. So for an ambient
temperature of 50 o C, it works out to be 2.5 W.
As the limit of IXD_414YI or IXD_414CI is 12 W,
subtracting 2.5 W from this yields 9.5 W. So
6.86 W is still within limit. Thermal Impedance
(Junction to Case) is 0.55o C/W for TO -263 and
TO-220, hence a rise in case temperature
should be within limit. If we increase Vcc to 15V,
conduction losses in MOSFET could reduce due
to lower RDS(on), but obtaining the same rise and
fall times will incur more power loss in driver due
to increased Vcc and Qg. If that happens,
approach described in Fig. (10) can be
employed. Or possibility of using a 30 Amp
driver can be investigated by power calculation
as shown above.
8. High overall efficiency of gate drive is possible.
6.0
ISOLATED
MOSFET/IGBT
Almost all possible configurations of bi-directional
switches employing IGBTs and FREDs (Fast
Recovery Epitaxial Diodes) are shown in Fig.(15).
When connected in series with the IGBT, the FRED
gives reverse blocking capability to the bi-directional
switch. When connected inversely across the IGBT,
the FRED provides a path for the line current to flow
in the reverse direction. Both techniques prevent
reverse voltage application to the emitter of IGBT.
GATE
DRIVE
FOR
Many applications of MOSFETs and IGBTs
require an isolated Gate Drive Circuit. For
example in H-B ridge and 3 -Phase Bridge
inverters, the upper MOSFETs and IGBTs
require an isolated gate drive, because the
Source/Emitter of upper MOSFETs/IGBTs are
not at the ground potential. Similarly in matrix
converters, all bi-directional switches require
isolated gate drive circuits.
Basically there are two popular techniques
available to implement isolated gate drive.
Fig.15 shows a method to implement an isolated
gate drive, using Gate Drive Transformers.
There are several advantages in using the gate
drive transformers:
1. If properly wound and built, they can give
adequate
galvanic isolation.
2. Depending on the drive current and voltage
required, they give step-up or step-down
facility.
3. They are immune to dv/dt transients
4. They experience no propagation delays.
5. Using modern high permeability cores, tiny
Gate Drive Transformers are now available
that meet most stringent design specifications.
6. There is no need to have an isolated power
supply.
7. Large duty cycle range of pulse widths is
possible, say, from 1% to 99%.
The two disadvantages are:
1. Gate drive transformers are not suitable for DC
and very low frequency signals.
2. It is difficult to implement overload/short circuit
protection for the upper MOSFET/IGBT.
As can be seen in Fig. (15), gate drive transformer’s
primary is fed from the output of a Driver IC. For 1:1
transformer, the peak output current and voltage on
the secondary is the same as the Driver IC’s output
values. As explained in section 5.0 before, knowing
the switching frequency, rise and fall times desired,
gate resistor used, total gate charge for the
MOSFET/IGBT and the voltage swing, one can
calculate the peak currents in the transformer
primary and secondary. Calculating the average
values for these currents will enable one to either
design the gate drive transformer or to enable one to
select the appropriate commercially available gate
drive transformers.
These bi-directional switches form nodes for matrix
converters or for A.C. to D.C. convert ers. Driving
IGBTs in these bi-directional switches can be easily
implemented using the gate drive transformers, as
shown in Fig. (15). As can be appreciated, R1, R2,
R3 and R4 help wave shaping by facilitating core
resetting. Z1, Z2, Z3 and Z4 protect the gate of IGBT
from voltage spikes above 18.7 volts.
Another method uses opto-couplers, as illustrated in
Fig. (16). As can be seen, opto-couplers need
isolated power supply. However, they facilitate D.C.
to several Mbits/sec of pulse rate and do provide
kilovolts of isolation. Keeping the same chain of
opto-coupler and Driver IC in each complementary
signal path will nullify the effect of slight propagation
delay through these opto-couplers and Driver ICs. It
is assumed here that difference in propagation
delays between two same type opto-couplers is
negligible.
Opto-couplers have the following features:
IXAN0009
1.Adequate galvanic isolation is possible. Many
opto-couplers are U/L listed.
2.Most opto-couplers are compatible with
TTL/CMOS/HCMOS inputs. Their outputs
depend on Vcc of isolated power supply. They
do need isolated power supply.
3.They are not immune to severe dv/dt
transients.
4.Signals
experience
propagation
delays
through opto-couplers.
5.Large duty cycle range of pulse width is
possible.
6.High overall efficiency of gate drive is possible.
7.Overload and short circuit protection can be
easily implemented using either DESAT
concept or current sense method. This is
explained in section 4.2.2, using Fig. (12) and
Fig. (13).
7. MATRIX CONVERTERS
Matrix converters afford great ease and
convenience in AC-AC conversion, without the
need for bulky energy storage components
required in AC-DC-AC conversion. In addition,
they allow regenerative energy to be fed back to
Mains and feature sinusoidal input and output
currents
and
controllable
input
current
displacement factor.
A basic 3 phase to 3 phase matrix converter
uses nine bi-directional switches as shown in Fig
(17). Several different configurations of
achieving bi-directional switches using IGBTs
and FREDs are shown in Fig.(15) and Fig.(16).
Each IGBT used in these bi-directional switches
will require isolated gate drive and it is here that
the real value and convenience of a gate drive
transformer can be appreciated. It is difficult to
imagine, use of opto-couplers, chiefly because
of the need for isolated power supply.
Tiny gate drive transformers come really handy
in this application, with all its attendant
advantages as listed above.
8. CONCLUSION
With proliferating applications of modern power
electronics worldwide, faster, more efficient and
more compact MOSFETs and IGBTs are
replacing older solid state and mechanical
devices. The design of newer and more efficient
techniques to turn these solid state devices on
and off is a subject that requires thorough study
and understanding of the internal structure and
dynamic processes involved in the working of
MOSFET/IGBTs.
Main emphasis in modern Power Electronics is to
reduce total losses dissipated in devices and
subsystems for higher operating efficiency and
achieving more compact designs, reducing volume
and weight of resultant systems. Thus, operation at
higher and higher switching frequencies is now a
necessity, and as a result, switching losses
predominate
in
the
power-loss-budget
in
semiconductor switches. Reducing switching losses
then becomes the single most crucial goal. Keeping
this goal in mind, the MOSFET/IGBT Drivers should
be designed so as to yield ultra fast rise and fall
times, matching or exceeding that of the driven
MOSFET/IGBTs. Propagation delay time should be
extremely low, facilitating implementation of fast
overload/short circuit protection. Most semiconductor
switches that can withstand momentary overloads
specify 10 us as the maximum allowable duration.
From the instance of sensing overload/short circuit
to the removal of gate signals, time elapsed should
be less than 80% of the rated SCSOA time.
With the advent of IC Drivers for these fast
MOSFET/IGBTs, the designer is relieved of the
tedious task of designing elaborate driver circuits.
Nevertheless, understanding these newer ICs, their
strengths and limitations, is of paramount
importance. Different configurations for particular
topologies call for specific application knowledge.
Illustrations are the best way to explain theory and
applications of these Driver ICs.
REFERENCES
1.
B. Jayant Baliga, “Power Semiconductor
Devices”,
PWS Publishing Company, Boston,
MA (1996)
2. Ned Mohan, Tore M. Undeland, William P.
Robbins: “POWER ELECTRONICS: Converters,
Applications
and Design”, John Wiley & Sons,
New York (1994)
3. Power Supply Design Seminar - 2001 series,
Unitrode Products from Texas Instruments.
4. Sam Ochi, “Driving your MOSFETs wild to obtain
greater
efficiencies, power densities, and lower
overall costs”, Power Electronics Europe, MayJune 2002
IXAN0009
RECOMMENDED FURTHER READING
1. George J. Krausse, “Gate Driver Design for
Switch-Mode Applications and the DE-SERIES
MOSFET Transistor”, Directed Energy,Inc.
Application
Note
available
from
<www.directedenergy.com>.
2000 he has been with IXYS Corporation as a
Senior Applications Engineer. He has written a
number of Application Reports on the subject of
Power Electronics. He is with IXYS Corporation,
3540 Bassett Street, Santa Clara, CA 95054, U.S.A.
Email: a.pathak@ixys.net
DISCLAIMER
Although information furnished here is believed
to be accurate and reliable, authors assume no
responsibility for the consequences of its use;
nor for any infringement of patents or other
rights of third parties, which may result from its
use.
AUTHORS BIOGRAPHIES
Mr. Ralph E. Locher earned his BS (Physics) from
Rensselaer, Troy, NY in 1961 and his MS (EE) from
Case-Western Reserve University, Cleveland, Ohio
in 1965. He worked for General Electric for many
years and was a contributing author of GE “SCR
Manual”. He has published many papers, while
working
for
Fairchild
Electronics,
National
Semiconductor, Power Integration, and IXYS
Corporation. He is currently the Director of
Applications Engineering at IXYS Corporation, Santa
Clara, CA. 95054, USA. Email: r.locher@ixys.net
Mr. Abhijit D. Pathak received his B. E.
(Electrical Engineering) from M. S. University,
Baroda, India in 1966. He obtained his MS (EE)
degree from University of Rhode Island,
Kingston, R. I., U.S.A. in 1971. From 1968 to
1973 he worked in USA in the areas of
development of D.C. Drives, high frequency high
Voltage Inverters, precision instrumentation and
measurement systems and their applications in
industry. In 1973, he returned to India and joined
Indian Space Research Organization (Govt. of
India) as a Senior Scientist/Engineer. At I.S.R.O.
he project engineered development of a number
of large electronic systems for the first time in
the country under sponsorship of Dept. of
Electronics, Govt. of India. He wrote a number of
technical
papers,
while
at
I.S.R.O.
Subsequently, in 1981, he started working in
private sector and developed a number of
industrial systems, including AC Variable
Frequency Drives, UPS Systems, Inverters,
Temperature Controllers, I/P (Current to
Pressure) Converters and Float-cum-boost
chargers. He also taught Final Year Degree
program at the University level for four years as
a Visiting Professor in the Department of
Instrumentation & Controls in India. Since May
IXAN0009
Table: 1
Driving Standard IXYS MOSFETs
fPWM= 100KHz
Peak
Standard Qg in Ig in
Mosfet
nC
fPWM = 200KHz
Rg in Total Pg Gate Peak Ig in
Amp Ohm
fPWM =400KHz
Rg in Total Pg
Gate
Peak Ig in Rg in Total Pg
Gate
Rcmd Pwr
W
Drvr
Amp
Ohm
W
Drvr
Amp
Ohm
W
Drvr
Out
IXFH6N100 88
IXFH12N100 122
0.9
1.2
11.4
8.2
0.1
0.2
Dx402
Dx402
1.8
2.4
5.7
4.1
0.3
0.4
Dx404
Dx404
3.5
4.9
2.8
2
0.5
0.7
Dx404
DD408
500W
1KW
IXTK21N100 250
2.5
4
0.4
Dx404
5
2
0.8
DD408
10
1
1.5
Dx414*
2KW
Gate
Rcmd Pwr
Drvr
Out
Driving Q-Class MOSFETs
fPWM= 200KHz
Q-Class
Mosfet
Peak
Qg in Ig in
nC
fPWM = 400KHz
Rg in Total Pg Gate Peak Ig in
fPWM =800KHz
Rg in Total Pg
Gate
Peak Ig in Rg in Total Pg
Drvr
Amp
Ohm
W
Amp Ohm
W
Drvr
Amp
Ohm
W
IXFH6N100Q 48
0.96
10.4
0.1
Dx402
1.9
5.2
0.28
Dx402
3.84
2.6
0.57
Dx404
500W
IXFH12N100Q 90
IXFH21N100Q 170
1.8
3.4
5.6
2.9
0.3
0.5
Dx404
Dx408*
3.6
6.8
2.8
1.5
0.5
1
DD408
Dx409*
7.2
13.6
1.4
0.7
1.1
2
Dx409*
Dx414*
1KW
2KW
Gate
Rcmd Pwr
Driving F-Class MOSFETs
fPWM= 1MHz
Q-Class
Qg in
Mosfet
nC
IXFH6N100F 54
IXFH12N100F 77
IXFK21N100F 160
Peak
Ig in
fPWM =2MHz
Rg in Total Pg Gate Peak Ig in
fPWM =4MHz
Rg in Total Pg
Gate
Peak Ig in Rg in Total Pg
Amp Ohm
W
Drvr
Amp
Ohm
W
Drvr
Amp
Ohm
W
Drvr
Out
1.08
1.54
3.2
0.8
1.2
2.4
Dx402
DD408*
DD408*
2.16
3.08
6.4
4.6
3.2
1.6
1.6
2.3
4.8
DD408*
DD408*
Dx409*
4.32
6.16
12.8
2.3
1.6
0.8
3.2
4.6
9.6
DD408*
Dx409*
Dx414*
500W
1KW
2KW
9.3
6.5
3.1
IXAN0009
Table: 2
tr + tf in % of tPWM = ~ 1 %
*Note:Gate Drivers Dx409* and Dx414* require heat sinks. All others without (*) don't require them.
Rg listed below are max. acceptable values for application; lower Rg values may be used.
Gate Driver Selection Table for HiPerFETs
Power Discretes/N-channel Power
HiPerFET Power MOSFETS
fPWM = 50 KHz
Device
Type
V DSS
Max .
IXFR180N06
IXFN340N06
60
60
180*
340*
0.005
0.003
420
600
2.1
5
0.32 Dx404
3
3
IXFH76N07-11
IXFH76N07-12
70
70
76
76
0.011
0.012
240
240
1.2
8
1.2
IXFK180N07
IXFK180N07
70
70
180*
180*
0.006
0.006
420
420
2.1
2.1
IXFK180N07
70
180*
0.006
420
IXFN180N07
IXFR180N07
70
70
180*
180*
0.007
0.006
480
420
IXFX180N07
IXFN200N07
70
70
180*
200*
0.006
0.006
IXFN280N07
IXFN340N07
70
70
280*
340*
0.006
0.004
IXFR180N085
IXFX180N085
85
85
180*
180*
IXFH280N095
85
IXFE180N10
IXFK180N10
V
ID
RDS(on) QG(on)
Amps Ohms
nC
fPWM= 100 KHz
fPWM = 200 KHz
fPWM = 400 KHz
fPWM= 800 KHz
Ig
Rg
Pg
Gate
Ig
Rg
Pg
Gate
Ig
Rg
Pg
Gate
Ig
Rg
Pg
Gate
Ig
Rg
Pg
Gate
Peak
Max
Max
Driver
Peak
Max
Max
Driver
Peak
Max
Max
Driver
Peak
Max
Max
Driver
Peak
Max
Max
Driver
Fmly
Amps
A m p s Ohms W a t t s
Fmly
A m p s O h m s Watts
4.2
2.4
0.45 Dx404
6
1.7
0.18 Dx402
2.4
4.2
8
0.18 Dx402
2.4
4.2
5
5
0.32 Dx404
0.32 Dx404
4.2
4.2
2.4
2.4
2.1
5
0.32 Dx404
4.2
2.4
2.1
4
5
0.36 Dx404
0.32 Dx404
4.8
4.2
420
480
2.1
2.4
5
4
0.32 Dx404
0.36 Dx404
420
600
2.1
3
5
3
0.32 Dx404
0.45 Dx404
0.007
0.007
320
320
1.6
6
1.6
6
280*
0.0044
600
3
100
100
180*
180*
0.008
0.008
360
360
IXFN180N10
IXFR180N10
100
100
180*
180*
0.008
0.008
IXFX180N10
IXFN230N10
100
100
180*
230*
0.008
0.006
IXFC80N10
100
80*
IXFC80N10
100
80*
IXFC80N10
IXFH80N10
100
100
IXFH80N10
IXFT80N10
Fmly
0.63 Dx409
A m p s Ohms W a t t s
8.4
1.2
0.9 Dx409
12
0.36 Dx404
4.8
0.36 Dx404
0.63 Dx409
0.63 Dx409
2.4
2.1
2.4
4.2
4.8
4.2
6
0.24 Dx404
0.24 Dx404
3
1.8
1.8
360
360
360
390
0.0125
0.0125
80*
80*
100
100
IXFT80N10
IXFT80N10
IXFT80N10
Fmly
A m p s Ohms W a t t s
1.26 Dx414*
16.8
0.6
0.8
1.8 Dx414*
24.0
0.4
2.1
0.72 Dx409
9.6
1.0
4.8
2.1
0.72 Dx409
9.6
1.0
8.4
8.4
1.2
1.2
1.26 Dx414*
1.26 Dx414*
16.8
16.8
0.6
0.6
0.63 Dx409
8.4
1.2
1.26 Dx414*
16.8
0.72 Dx409
0.63 Dx409
9.6
8.4
1.0
1.2
1.44 Dx414*
1.26 Dx414*
19.2
16.8
2.4
2.1
0.63 Dx409
0.72 Dx409
8.4
9.6
1.2
1.0
1.26 Dx414*
1.44 Dx414*
2.4
1.7
0.63 Dx409
0.9 Dx409
8.4
12
1.2
0.8
1.26 Dx414*
1.8 Dx414*
3.2
3.1
0.48 Dx404
6.4
1.6
3.2
3.1
0.48 Dx404
6.4
1.6
0.45 Dx404
6
1.7
0.9 Dx409
12
6
6
0.27 Dx404
0.27 Dx404
3.6
3.6
2.8
2.8
0.54 Dx404
0.54 Dx404
1.8
1.8
6
6
0.27 Dx404
0.27 Dx404
3.6
3.6
2.8
2.8
1.8
1.95
6
5
0.27 Dx404
0.29 Dx404
3.6
3.9
230
1.15
9
0.17 Dx402
2.3
230
1.15
9
0.17 Dx402
2.3
0.0125
0.0125
230
230
1.15
1.15
9
9
0.17 Dx402
0.17 Dx402
80*
80*
0.0125
0.0125
230
230
1.15
1.15
9
9
100
100
100
80*
80*
80*
0.0125
0.0125
0.0125
230
230
230
1.15
1.15
IXFR150N15
150
105*
0.0125
360
IXFK150N15
IXFN150N15
IXFX150N15
150
150
150
150*
150*
150*
0.0125
0.0125
0.0125
IXFH74N20
IXFT74N20
IXFK80N20
200
200
200
74
74
80
IXFR120N20
200
IXFK120N20
IXFN120N20
Ohms W a t t s
Fmly
33.6
0.3
3.6 Dx430*
48
0.2
7.2 Dx430*
1.44 Dx414*
19.2
0.5
2.88 Dx430*
1.44 Dx414*
19.2
0.5
2.88 Dx430*
2.52 Dx430*
2.52 Dx430*
33.6
33.6
0.3
0.3
5.04 Dx430*
5.04 Dx430*
0.6
2.52 Dx430*
33.6
0.3
5.04 Dx430*
0.5
0.6
2.88 Dx430*
2.52 Dx430*
38.4
33.6
0.3
0.3
5.76 Dx430*
5.04 Dx430*
16.8
19.2
0.6
0.5
2.52 Dx430*
2.88 Dx430*
33.6
38.4
0.3
0.3
5.04 Dx430*
5.76 Dx430*
16.8
24.0
0.6
0.4
2.52 Dx430*
3.6 Dx430*
33.6
48
0.3
0.2
5.04 Dx430*
7.2 Dx430*
0.96 Dx409*
12.8
0.8
1.92 Dx414*
25.6
0.4
3.84 Dx430*
0.96 Dx409*
12.8
0.8
1.92 Dx414*
25.6
0.4
3.84 Dx430*
0.8
1.8 Dx414*
24.0
0.4
3.6 Dx430*
48
0.2
7.2 Dx430*
7.2
7.2
1.4
1.4
1.08 Dx409*
1.08 Dx409*
14.4
14.4
0.7
0.7
2.16 Dx430*
2.16 Dx430*
28.8
28.8
0.3
0.3
4.32 Dx430*
4.32 Dx430*
0.54 Dx404
0.54 Dx404
7.2
7.2
1.4
1.4
1.08 Dx409*
1.08 Dx409*
14.4
14.4
0.7
0.7
2.16 Dx430*
2.16 Dx430*
28.8
28.8
0.3
0.3
4.32 Dx430*
4.32 Dx430*
2.8 0.54 Dx404
2.6 0.585 Dx404
7.2
7.8
1.4
1.3
1.08 Dx409*
1.17 Dx409*
14.4
15.6
0.7
0.6
2.16 Dx430*
2.34 Dx430*
28.8
31.2
0.3
0.3
4.32 Dx430*
4.68 Dx430*
4.3 0.345 Dx404
4.6
2.2
0.69 Dx409
9.2
1.1
1.38 Dx414*
18.4
0.5
2.76 Dx430*
4.3 0.345 Dx404
4.6
2.2
0.69 Dx409
9.2
1.1
1.38 Dx414*
18.4
0.5
2.76 Dx430*
2.3
2.3
4.3 0.345 Dx404
4.3 0.345 Dx404
4.6
4.6
2.2
2.2
0.69 Dx409
0.69 Dx409
9.2
9.2
1.1
1.1
1.38 Dx414*
1.38 Dx414*
18.4
18.4
0.5
0.5
2.76 Dx430*
2.76 Dx430*
0.17 Dx402
0.17 Dx402
2.3
2.3
4.3 0.345 Dx404
4.3 0.345 Dx404
4.6
4.6
2.2
2.2
0.69 Dx409
0.69 Dx409
9.2
9.2
1.1
1.1
1.38 Dx414*
1.38 Dx414*
18.4
18.4
0.5
0.5
2.76 Dx430*
2.76 Dx430*
9
9
0.17 Dx402
0.17 Dx402
2.3
2.3
4.3 0.345 Dx404
4.3 0.345 Dx404
4.6
4.6
2.2
2.2
0.69 Dx409
0.69 Dx409
9.2
9.2
1.1
1.1
1.38 Dx414*
1.38 Dx414*
18.4
18.4
0.5
0.5
2.76 Dx430*
2.76 Dx430*
1.15
9
0.17 Dx402
2.3
4.3 0.345 Dx404
4.6
2.2
0.69 Dx409
9.2
1.1
1.38 Dx414*
18.4
0.5
2.76 Dx430*
1.8
6
0.27 Dx404
3.6
2.8
0.54 Dx404
7.2
1.4
1.08 Dx409*
14.4
0.7
2.16 Dx430*
28.8
0.3
4.32 Dx430*
360
360
360
1.8
1.8
6
6
0.27 Dx404
0.27 Dx404
3.6
3.6
2.8
2.8
0.54 Dx404
0.54 Dx404
7.2
7.2
1.4
1.4
1.08 Dx409*
1.08 Dx409*
14.4
14.4
0.7
0.7
2.16 Dx430*
2.16 Dx430*
28.8
28.8
0.3
0.3
4.32 Dx430*
4.32 Dx430*
1.8
6
0.27 Dx404
3.6
2.8
0.54 Dx404
7.2
1.4
1.08 Dx409*
14.4
0.7
2.16 Dx430*
28.8
0.3
4.32 Dx430*
0.03
0.03
0.03
280
280
280
1.4
1.4
7
7
0.21 Dx402
0.21 Dx402
2.8
2.8
3.6
3.6
0.42 Dx404
0.42 Dx404
5.6
5.6
1.8
1.8
0.84 Dx409*
0.84 Dx409*
11.2
11.2
0.9
0.9
1.68 Dx414*
1.68 Dx414*
22.4
22.4
0.4
0.4
3.36 Dx430*
3.36 Dx430*
1.4
7
0.21 Dx402
2.8
3.6
0.42 Dx404
5.6
1.8
0.84 Dx409*
11.2
0.9
1.68 Dx414*
22.4
0.4
3.36 Dx430*
105
0.017
360
1.8
6
0.27 Dx404
3.6
2.8
0.54 Dx404
7.2
1.4
1.08 Dx409*
14.4
0.7
2.16 Dx430*
28.8
0.3
4.32 Dx430*
200
200
120
120
0.017
0.017
360
360
1.8
1.8
6
6
0.27 Dx404
0.27 Dx404
3.6
3.6
2.8
2.8
0.54 Dx404
0.54 Dx404
7.2
7.2
1.4
1.4
1.08 Dx409*
1.08 Dx409*
14.4
14.4
0.7
0.7
2.16 Dx430*
2.16 Dx430*
28.8
28.8
0.3
0.3
4.32 Dx430*
4.32 Dx430*
IXFX120N20
IXFN180N20
200
200
120
180
0.017
0.01
360
380
1.8
1.9
6
5
0.27 Dx404
0.29 Dx404
3.6
3.8
2.8
2.6
0.54 Dx404
0.57 Dx404
7.2
7.6
1.4
1.3
1.08 Dx409*
1.14 Dx409*
14.4
15.2
0.7
0.7
2.16 Dx430*
2.28 Dx430*
28.8
30.4
0.3
0.3
4.32 Dx430*
4.56 Dx430*
IXFR100N25
250
87
0.027
300
1.5
7
0.23 Dx402
3
3.3
0.45 Dx404
6
1.7
0.9 Dx409*
12.0
0.8
1.8 Dx414*
24
0.4
3.6 Dx430*
IXFK100N25
IXFN100N25
IXFX100N25
250
250
250
100
100
100
0.027
0.027
0.027
300
300
300
1.5
1.5
7
7
0.23 Dx402
0.23 Dx402
3
3
3.3
3.3
0.45 Dx404
0.45 Dx404
6
6
1.7
1.7
0.9 Dx409*
0.9 Dx409*
12.0
12.0
0.8
0.8
1.8 Dx414*
1.8 Dx414*
24
24
0.4
0.4
3.6 Dx430*
3.6 Dx430*
1.5
7
0.23 Dx402
3
3.3
0.45 Dx404
6
1.7
0.9 Dx409*
12.0
0.8
1.8 Dx414*
24
0.4
3.6 Dx430*
IXFK73N30
300
73
0.045
360
1.8
6
0.27 Dx404
3.6
2.8
0.54 Dx404
7.2
1.4
1.08 Dx409*
14.4
0.7
2.16 Dx430*
28.8
0.3
4.32 Dx430*
IXFN73N30
IXFR90N30
300
300
73
75
0.045
0.033
360
360
1.8
1.8
6
6
0.27 Dx404
0.27 Dx404
3.6
3.6
2.8
2.8
0.54 Dx404
0.54 Dx404
7.2
7.2
1.4
1.4
1.08 Dx409*
1.08 Dx409*
14.4
14.4
0.7
0.7
2.16 Dx430*
2.16 Dx430*
28.8
28.8
0.3
0.3
4.32 Dx430*
4.32 Dx430*
IXFK90N30
IXFN90N30
300
300
90
90
0.033
0.033
360
360
1.8
1.8
6
6
0.27 Dx404
0.27 Dx404
3.6
3.6
2.8
2.8
0.54 Dx404
0.54 Dx404
7.2
7.2
1.4
1.4
1.08 Dx409*
1.08 Dx409*
14.4
14.4
0.7
0.7
2.16 Dx430*
2.16 Dx430*
28.8
28.8
0.3
0.3
4.32 Dx430*
4.32 Dx430*
IXFX90N30
IXFN130N30
300
300
90
130
0.033
0.018
360
380
1.8
1.9
6
5
0.27 Dx404
0.29 Dx404
3.6
3.8
2.8
2.6
0.54 Dx404
0.57 Dx404
7.2
7.6
1.4
1.3
1.08 Dx409*
1.14 Dx409*
14.4
15.2
0.7
0.7
2.16 Dx430*
2.28 Dx430*
28.8
30.4
0.3
0.3
4.32 Dx430*
4.56 Dx430*
IXFH30N50
500
30
0.16
227
1.135
9
0.17 Dx402
2.27
4.4 0.341 Dx404
4.54
2.2 0.681 Dx409
9.1
1.1 1.362 Dx414* 18.16
0.6 2.724 Dx430*
IXFT30N50
500
30
0.16
227
1.135
9
0.17 Dx402
2.27
4.4 0.341 Dx404
4.54
2.2 0.681 Dx409
9.1
1.1 1.362 Dx414* 18.16
0.6 2.724 Dx430*
IXFH32N50
IXFT32N50
500
500
32
32
0.15
0.15
227
227
1.135
1.135
9
9
0.17 Dx402
0.17 Dx402
2.27
2.27
4.4 0.341 Dx404
4.4 0.341 Dx404
4.54
4.54
2.2 0.681 Dx409
2.2 0.681 Dx409
9.1
9.1
1.1 1.362 Dx414* 18.16
1.1 1.362 Dx414* 18.16
0.6 2.724 Dx430*
0.6 2.724 Dx430*
IXFK33N50
IXFK35N50
500
500
33
35
0.16
0.15
227
227
1.135
1.135
9
9
0.17 Dx402
0.17 Dx402
2.27
2.27
4.4 0.341 Dx404
4.4 0.341 Dx404
4.54
4.54
2.2 0.681 Dx409
2.2 0.681 Dx409
9.1
9.1
1.1 1.362 Dx414* 18.16
1.1 1.362 Dx414* 18.16
0.6 2.724 Dx430*
0.6 2.724 Dx430*
IXFE48N50D2**
500
42
0.1
270
1.35
7
0.20 Dx402
2.7
3.7 0.405 Dx404
5.4
0.9
0.5
1.9
0.81 Dx409*
10.8
2.52 Dx430*
1.62 Dx414*
21.6
5.04 Dx430*
3.24 Dx430*
IXAN0009
Table: 3
tr + tf in % of tPWM = ~ 1 %
*Note:Gate Drivers Dx409* and Dx414* require heat sinks. All others without (*) don't require them.
Rg listed below are max. acceptable values for application; lower Rg values may be used.
Gate Driver Selection Table for HiPerFETs
Power Discretes/N-channel Power
HiPerFET Power MOSFETS
RDS(on) Q G(on)
fPWM = 100 KHz
fPWM = 200 KHz
fPWM = 400 KHz
fPWM = 800 KHz
V DSS
Max .
IXFR50N50
IXFK44N50~
500
500
43
44
0.1
0.12
330
270
1.65
1.35
6
7
0.25 Dx402
0.20 Dx402
3.3
2.7
3.0 0.495 Dx404
3.7 0.405 Dx404
6.6
5.4
1.5
1.9
0.99 Dx409*
0.81 Dx409*
13.2
10.8
0.8
0.9
1.98 Dx414*
1.62 Dx414*
26.4
21.6
0.4
0.5
3.96 Dx430*
3.24 Dx430*
IXFN44N50~
500
IXFN44N50U2** 500
44
44
0.12
0.12
270
270
1.35
1.35
7
7
0.20 Dx402
0.20 Dx402
2.7
2.7
3.7 0.405 Dx404
3.7 0.405 Dx404
5.4
5.4
1.9
1.9
0.81 Dx409*
0.81 Dx409*
10.8
10.8
0.9
0.9
1.62 Dx414*
1.62 Dx414*
21.6
21.6
0.5
0.5
3.24 Dx430*
3.24 Dx430*
IXFN44N50U3** 500
V
ID
fPWM = 50 KHz
Device
Type
Amps Ohms
nC
Ig
Rg
Pg
Gate
Ig
Rg
Pg
Peak
Max
Max
Driver
Peak
Max
Max
A m p s Ohms Watts
Fmly
A m p s Ohms W a t t s
Gate
Ig
Driver P e a k
Fmly
Rg
Pg
Gate
Ig
Rg
Pg
Max
Max
Driver
Peak
Max
Max
Amps O h m s W a t t s
Fmly
A m p s O h m s Watts
Gate
Ig
Driver P e a k
Fmly
Rg
Pg
Gate
Max
Max
Driver
Amps Ohms Watts
Fmly
44
0.12
270
1.35
7
0.20 Dx402
2.7
3.7 0.405 Dx404
5.4
1.9
0.81 Dx409*
10.8
0.9
1.62 Dx414*
21.6
0.5
3.24 Dx430*
500
500
48
48
0.1
0.1
270
270
1.35
1.35
7
7
0.20 Dx402
0.20 Dx402
2.7
2.7
3.7 0.405 Dx404
3.7 0.405 Dx404
5.4
5.4
1.9
1.9
0.81 Dx409*
0.81 Dx409*
10.8
10.8
0.9
0.9
1.62 Dx414*
1.62 Dx414*
21.6
21.6
0.5
0.5
3.24 Dx430*
3.24 Dx430*
IXFN48N50U2** 500
48
0.1
270
1.35
7
0.20 Dx402
2.7
3.7 0.405 Dx404
5.4
1.9
0.81 Dx409*
10.8
0.9
1.62 Dx414*
21.6
0.5
3.24 Dx430*
IXFN48N50U3** 500
IXFR55N50
500
48
48
0.1
0.08
330
330
1.65
1.65
6
6
0.25 Dx402
0.25 Dx402
3.3
3.3
3.0 0.495 Dx404
3.0 0.495 Dx404
6.6
6.6
1.5
1.5
0.99 Dx409*
0.99 Dx409*
13.2
13.2
0.8
0.8
1.98 Dx414*
1.98 Dx414*
26.4
26.4
0.4
0.4
3.96 Dx430*
3.96 Dx430*
IXFK48N50
IXFN48N50
IXFK50N50
500
50
0.1
330
1.65
6
0.25 Dx402
3.3
3.0 0.495 Dx404
6.6
1.5
0.99 Dx409*
13.2
0.8
1.98 Dx414*
26.4
0.4
3.96 Dx430*
IXFN50N50
IXFX50N50
500
500
50
50
0.1
0.1
330
330
1.65
1.65
6
6
0.25 Dx402
0.25 Dx402
3.3
3.3
3.0 0.495 Dx404
3.0 0.495 Dx404
6.6
6.6
1.5
1.5
0.99 Dx409*
0.99 Dx409*
13.2
13.2
0.8
0.8
1.98 Dx414*
1.98 Dx414*
26.4
26.4
0.4
0.4
3.96 Dx430*
3.96 Dx430*
IXFE50N50
IXFK55N50
500
500
52
55
0.1
0.08
330
330
1.65
1.65
6
6
0.25 Dx402
0.25 Dx402
3.3
3.3
3.0 0.495 Dx404
3.0 0.495 Dx404
6.6
6.6
1.5
1.5
0.99 Dx409*
0.99 Dx409*
13.2
13.2
0.8
0.8
1.98 Dx414*
1.98 Dx414*
26.4
26.4
0.4
0.4
3.96 Dx430*
3.96 Dx430*
IXFN55N50
IXFX55N50
500
500
55
55
0.08
0.08
330
330
1.65
6
0.25 Dx402
3.3
3.0 0.495 Dx404
6.6
1.5
0.99 Dx409*
13.2
0.8
1.98 Dx414*
26.4
0.4
3.96 Dx430*
1.65
6
0.25 Dx402
3.3
3.0 0.495 Dx404
6.6
1.5
0.99 Dx409*
13.2
0.8
1.98 Dx414*
26.4
0.4
3.96 Dx430*
IXFE80N50
500
71
0.05
380
1.9
5
0.29 Dx404
3.8
2.6
0.57 Dx404
7.6
1.3
1.14 Dx409*
15.2
0.7
2.28 Dx430*
30.4
0.3
4.56 Dx430*
IXFN75N50
IXFN80N50
500
500
75
80
0.055
0.05
380
380
1.9
1.9
5
5
0.29 Dx404
0.29 Dx404
3.8
3.8
2.6
2.6
0.57 Dx404
0.57 Dx404
7.6
7.6
1.3
1.3
1.14 Dx409*
1.14 Dx409*
15.2
15.2
0.7
0.7
2.28 Dx430*
2.28 Dx430*
30.4
30.4
0.3
0.3
4.56 Dx430*
4.56 Dx430*
IXFK48N55
550
48
0.11
330
1.65
6
0.25 Dx402
3.3
3.0 0.495 Dx404
6.6
1.5
0.99 Dx409*
13.2
0.8
1.98 Dx414*
26.4
0.4
3.96 Dx430*
IXFN48N55
550
48
0.11
330
1.65
6
0.25 Dx402
3.3
3.0 0.495 Dx404
6.6
1.5
0.99 Dx409*
13.2
0.8
1.98 Dx414*
26.4
0.4
3.96 Dx430*
IXFX48N55
IXFK32N60~
550
600
48
32
0.11
0.25
330
325
1.65
1.625
6
6
0.25 Dx402
0.24 Dx402
3.3
3.25
3.0 0.495 Dx404
3.1 0.488 Dx404
6.6
6.5
1.5 0.99 Dx409*
1.5 0.975 Dx409*
13.2
13.0
0.8
0.8
1.98 Dx414*
1.95 Dx414*
26.4
26
0.4
0.4
3.96 Dx430*
3.9 Dx430*
IXFN32N60~
600
32
0.25
325
1.625
6
0.24 Dx402
3.25
3.1 0.488 Dx404
6.5
1.5 0.975 Dx409*
13.0
0.8
1.95 Dx414*
26
0.4
3.9 Dx430*
IXFK36N60
IXFN36N60
600
600
36
36
0.18
0.18
325
325
1.625
1.625
6
6
0.24 Dx402
0.24 Dx402
3.25
3.25
3.1 0.488 Dx404
3.1 0.488 Dx404
6.5
6.5
1.5 0.975 Dx409*
1.5 0.975 Dx409*
13.0
13.0
0.8
0.8
1.95 Dx414*
1.95 Dx414*
26
26
0.4
0.4
3.9 Dx430*
3.9 Dx430*
IXFR44N60
600
38
0.13
330
1.65
6
0.25 Dx402
3.3
3.0 0.495 Dx404
6.6
1.5
0.99 Dx409*
13.2
0.8
1.98 Dx414*
26.4
0.4
3.96 Dx430*
IXFE44N60
IXFK44N60
600
600
40
44
0.13
0.13
330
330
1.65
1.65
6
6
0.25 Dx402
0.25 Dx402
3.3
3.3
3.0 0.495 Dx404
3.0 0.495 Dx404
6.6
6.6
1.5
1.5
0.99 Dx409*
0.99 Dx409*
13.2
13.2
0.8
0.8
1.98 Dx414*
1.98 Dx414*
26.4
26.4
0.4
0.4
3.96 Dx430*
3.96 Dx430*
IXFN44N60
IXFX44N60
600
600
44
44
0.13
0.13
330
330
1.65
1.65
6
6
0.25 Dx402
0.25 Dx402
3.3
3.3
3.0 0.495 Dx404
3.0 0.495 Dx404
6.6
6.6
1.5
1.5
0.99 Dx409*
0.99 Dx409*
13.2
13.2
0.8
0.8
1.98 Dx414*
1.98 Dx414*
26.4
26.4
0.4
0.4
3.96 Dx430*
3.96 Dx430*
IXFN60N60
IXFK27N80
IXFN27N80
600
800
800
60
27
27
0.08
0.32
0.32
380
350
350
1.9
1.75
5
6
0.29 Dx402
0.26 Dx404
3.8
3.5
2.6 0.57 Dx404
2.9 0.525 Dx404
7.6
7
1.3
1.4
1.14 Dx409*
1.05 Dx409*
15.2
14.0
0.7
0.7
2.28 Dx430*
2.1 Dx430*
30.4
28
0.3
0.4
4.56 Dx430*
4.2 Dx430*
1.75
6
0.26 Dx402
3.5
2.9 0.525 Dx402
7
1.4
1.05 Dx409*
14.0
0.7
2.1 Dx430*
28
0.4
4.2 Dx430*
IXFR34N80
800
28
0.24
270
1.35
7
0.20 Dx402
2.7
3.7 0.405 Dx404
5.4
1.9
0.81 Dx409*
10.8
0.9
1.62 Dx414*
21.6
0.5
3.24 Dx430*
IXFK34N80
800
34
0.24
270
1.35
7
0.20 Dx402
2.7
3.7 0.405 Dx404
5.4
1.9
0.81 Dx409*
10.8
0.9
1.62 Dx414*
21.6
0.5
3.24 Dx430*
IXFN34N80
IXFX34N80
800
800
34
34
0.24
0.24
270
270
1.35
1.35
7
7
0.20 Dx402
0.20 Dx402
2.7
2.7
3.7 0.405 Dx404
3.7 0.405 Dx404
5.4
5.4
1.9
1.9
0.81 Dx409*
0.81 Dx409*
10.8
10.8
0.9
0.9
1.62 Dx414*
1.62 Dx414*
21.6
21.6
0.5
0.5
3.24 Dx430*
3.24 Dx430*
IXFN44N80
800
44
0.145
380
1.9
5
0.29 Dx404
3.8
2.6
0.57 Dx404
7.6
1.3
1.14 Dx409*
15.2
0.7
2.28 Dx430*
30.4
0.3
4.56 Dx430*
IXFK25N90
IXFN25N90
900
900
25
25
0.33
0.33
240
240
1.2
1.2
8
8
0.18 Dx402
0.18 Dx402
2.4
2.4
4.2
4.2
0.36 Dx404
0.36 Dx404
4.8
4.8
2.1
2.1
0.72 Dx409
0.72 Dx409
9.6
9.6
1.0
1.0
1.44 Dx414*
1.44 Dx414*
19.2
19.2
0.5
0.5
2.88 Dx430*
2.88 Dx430*
IXFX25N90
900
25
0.33
240
1.2
8
0.18 Dx402
2.4
4.2
0.36 Dx404
4.8
2.1
0.72 Dx409
9.6
1.0
1.44 Dx414*
19.2
0.5
2.88 Dx430*
IXFK26N90
IXFN26N90
IXFX26N90
900
900
900
26
26
26
0.3
0.3
0.3
240
240
240
1.2
1.2
8
8
0.18 Dx402
0.18 Dx402
2.4
2.4
4.2
4.2
0.36 Dx404
0.36 Dx404
4.8
4.8
2.1
2.1
0.72 Dx409
0.72 Dx409
9.6
9.6
1.0
1.0
1.44 Dx414*
1.44 Dx414*
19.2
19.2
0.5
0.5
2.88 Dx430*
2.88 Dx430*
1.2
8
0.18 Dx402
2.4
4.2
0.36 Dx404
4.8
2.1
0.72 Dx409
9.6
1.0
1.44 Dx414*
19.2
0.5
2.88 Dx430*
IXFN39N90
900
39
0.22
390
1.95
5
0.29 Dx404
3.9
2.6 0.585 Dx404
7.8
1.3
1.17 Dx409*
15.6
0.6
2.34 Dx430*
31.2
0.3
4.68 Dx430*
IXFH14N100
1000
14
0.75
220
1.1
9
0.17 Dx402
2.2
4.5
0.33 Dx404
4.4
2.3
0.66 Dx409
8.8
1.1
1.32 Dx414*
17.6
0.6
2.64 Dx430*
IXFT14N100
1000
14
0.75
220
1.1
9
0.17 Dx402
2.2
4.5
0.33 Dx404
4.4
2.3
0.66 Dx409
8.8
1.1
1.32 Dx414*
17.6
0.6
2.64 Dx430*
IXFX14N100
IXFH15N100
1000
1000
14
15
0.75
0.7
220
220
1.1
1.1
9
9
0.17 Dx402
0.17 Dx402
2.2
2.2
4.5
4.5
0.33 Dx404
0.33 Dx404
4.4
4.4
2.3
2.3
0.66 Dx409
0.66 Dx409
8.8
8.8
1.1
1.1
1.32 Dx414*
1.32 Dx414*
17.6
17.6
0.6
0.6
2.64 Dx430*
2.64 Dx430*
IXFT15N100
1000
15
0.7
220
1.1
9
0.17 Dx402
2.2
4.5
0.33 Dx404
4.4
2.3
0.66 Dx409
8.8
1.1
1.32 Dx414*
17.6
0.6
2.64 Dx430*
IXFX15N100
IXFF24N100
1000
1000
15
22
0.7
0.39
220
250
1.1
1.25
9
8
0.17 Dx402
0.19 Dx402
2.2
2.5
4.5 0.33 Dx404
4.0 0.375 Dx404
4.4
5
2.3
2.0
0.66 Dx409
0.75 Dx409
8.8
10.0
1.1
1.0
1.32 Dx414*
1.5 Dx414*
17.6
20
0.6
0.5
2.64 Dx430*
3 Dx430*
IXFR24N100
IXFE24N100
1000
1000
22
23
0.39
0.39
250
250
1.25
1.25
8
8
0.19 Dx402
0.19 Dx402
2.5
2.5
4.0 0.375 Dx404
4.0 0.375 Dx404
5
5
2.0
2.0
0.75 Dx409
0.75 Dx409
10.0
10.0
1.0
1.0
1.5 Dx414*
1.5 Dx414*
20
20
0.5
0.5
3 Dx430*
3 Dx430*
IXFN23N100
1000
23
0.43
250
1.25
8
0.19 Dx402
2.5
4.0 0.375 Dx404
5
2.0
0.75 Dx409
10.0
1.0
1.5 Dx414*
20
0.5
3 Dx430*
IXFK24N100
IXFN24N100
IXFX24N100
1000
1000
1000
24
24
24
0.39
0.39
0.39
250
250
250
1.25
1.25
8
8
0.19 Dx402
0.19 Dx402
2.5
2.5
4.0 0.375 Dx404
4.0 0.375 Dx404
5
5
2.0
2.0
0.75 Dx409
0.75 Dx409
10.0
10.0
1.0
1.0
1.5 Dx414*
1.5 Dx414*
20
20
0.5
0.5
3 Dx430*
3 Dx430*
1.25
8
0.19 Dx402
2.5
4.0 0.375 Dx404
5
2.0
0.75 Dx409
10.0
1.0
1.5 Dx414*
20
0.5
3 Dx430*
IXAN0009
H.V.DC
L
O
A
D
+15VDC
V
INPUT
TIME
C1
3
8
7
IC2
6
4
C
D1
G
RG
5
ZD1
ZD2
Rp
CS
Q
RS
E
C1,C3 : 22 MFD, 25VDC Tantalum capacitors
C2 : 2200 MFD, 35VDC Electrolytic capacitors
T1 : 220 VAC to 15-0-15 VAC, 15VA control transformer
or 110VAC TO 15-0-15 VAC, 15 VA Control Transformer.
Q : IXLF19N250A
IC1 : 7815 Regulator
+15VDC
IC2 : IXDN409 or IXDN414
IC1
D1 : IN5817
+ C2
D2,D3 : IN4002
C3
ZD1,ZD2 : 18V, 400MW ZENERS
RG1 : 3.3 ohms to 27 ohms
depending on Turn-ON speed
Rp : 2K2, ¼ W, 5%
Cs, Rs : Snubber network to reduce IGBT switching
losses. Value depends upon fsw. Suggest:
Cs=0.1 MFD, Rs=10 to 33 ohms
R1 : 10K, ¼ w
D2
C.T.
T1
R1
1
2
D3
Fig. (1) Circuit schematic showing how to use IXDN409 or IXDN414 to drive an IGBT
Vcc : 3V to 25VDC
1
I/P
2
3
4
R5
U5
U4
I
X
D
D
4
1
4
Ld=10mH
8
+
C9
7
6
5
R6
R9
2
U2
U5
C3
1
1
7815
+ C1+
C2
D2
Rs=0.005 Ohm
+15V
R4
D1
MOSFET
MODULE
VMO580-02F
Q1
U2
T1
Rd=0.1 Ohm
R7
U5
C7
M
A
I
N
S
VDC
2
3
+
C2
+15V TO
ALL ICs
LM-317
2
+
U3
-
5
C4
Ls=20nH
(Representing
stray inductance)
4
3
R2
R3
Z1
P2
R1
P1
Fig. (2). Evaluation circuit to test IXDD408 and IXDD414 for soft turn off.
IXAN0009
Fig. (3) Photographic +ve and -ve and component layout with silk sceen diagram for Circuit of Fig.(2).
Bill of Materials for Fig.(2)
Resistors:
Capacitors:
Diodes:
ICs:
R1:
R2:
R3:
R4:
R5:
R6:
R7:
R8:
C1: 1000µF;35VWDC
C2: 22µF, 63 VWDC
C3: 1pF, silver dipped mica
C4: 100pF silver dipped mica
C5: 0.1µF, 35WDC Tantalum
C6: 0.1µF, 35VWDC Tantalum
C7: 1pF silver dipped mica
C8: 0.1µF, 35VWDC Tantalum
C9: 0.1µF, 35VWDC Tantalum
C10: 0.1µF, 35VWDC Tantalum
D1: 1N4002 or BA 159
D2: 1N4002 or BA 159
U1: IXDD408PI or
IXDD414PI
U2: CD4001
U3: LM339
U4: CD4011
U5: CD4049
U6: IXDD408YI or
IXDD414YI
Note: Either use U1
or U6, but not both.
240
560
10K
5K
1Meg
1K5
Rg-T.B.D.
1Meg
Trimmers:
P1: 5K, 3006P Bourns or Spectrol
P2: 1K, 3006P Bourns or Spectrol
Zener Diodes:
1. Z1: 1N821
Voltage Regulators:
1. 7815
2. LM317T
Transistors:
1. Q1: 2N7000
IXAN0009
Fig.(4) A circuit schematic showing how to drive upper and lower MOSFETs in a
phase leg topology using a Low and High side driver pair.
IXAN0009
SD2
C2
+
ON
Vcc
+
-
SD1
Q1
OFF
2xVcc -V SD2
+
t
-
ON
18V
Zener
+
C1
Q2
OFF
t
-
f sw ~
~ 400 KHz
Fig. (5) Basic Charge Pump Doubler
Vcc = +15V
DB
V DC ~
~ 500VDC
R5
+
R2
P1
V
R1
CB R4
C1
2
3
R3
15V
1
D2
4
I
I
X
X
D
D
D or D
4
4
0
1
8
4
8
7
C2
Rgext
Q4
6
5
D1
Q1
0
L
CF
L
O
A
D
t
I/P
DB : DSEP9-06CR
D1 : DSEC60-12A
D2 : 1N5817
C1 : 20MFD,25V
C2 : 20MFD,1000 Volts,CSI 10DC0020
CB : 10MFD,25VDC
P1 : 5K Trim pot
Rgext: 1.0 Ohm to 4.7 Ohm
CF : GE A28F5601
0.1MFD,1000 Volts
R1 : 1K
R2 : 10K
R3 : 2K
R4 : 10K
R5 : 1 Ohm
L : 5µH,DALE IH-5
Fig. (6) Basic bootstrap gate drive technique
IXAN0009
Fig.(7) Boosting output gate drive to +/-8 A and charge pump output to 350mA for 400kHz switching of
size 9 devices with the IXBD4410/4411 gate driver chip set.
Fig.(8) Boosting output gate drive to +/-14 A and charge pump output to 500mA for 400kHz switching
of size 9 devices with the IXBD4410/4411 gate driver chip set.
IXAN0009
IXAN0009
V
V
t
4
3
2
t
V EE = -5V
I/P
+Vcc=+15V
5
6
7
8
4
3
2
1
CF
+
Z2
Z1
I/P
4
3
2
1
I
X
D
D
4
0
8
5
6
7
8
+
Rb
CF
Q2
Rgoff
Rgon
I
I
X
X
D
D
D or D
4
4
0
1
8
4
Q1
Fig (9) Turn-off enhancement methods.
D Rgoff
D Rgon
L
O
A
D
H.V.DC
5
6
7
8
Z2
Z1
Q2
D
H.V.DC
Z2
Z1
L
O
A
D
1. Z1,Z2 : 18V,400mW
Zener diodes
2. D : 1N5821
3. Q1 : D44VH10
4. Q2 : D45VH10
5. Rb : 10 Ohms,1/4w,1%
LOAD
Rgon
CF
IGBT
+
Vcc
Fig. (10). Technique to boost current output and provide -ve bias to achieve faster turn off for high power MOSFET and IGBT Modules
0.0V
-5V
+15V
I/P
1
I
I
X
X
D
D
D or D
4
4
0
1
8
4
Vcc
H.V.DC
IXAN0009
D
C
B
4
3
2
1
4
3
2
I
X
D
D
4
0
4
R7
I
X
D
D
4
0
4
R5
+
5
6
7
8
5
6
7
8
R8
R6
T1
R1
D1
R G4
R4
D4
R G1
Vcc=16VDC
Q4
Q1
Z6
Z8
D.C.SUPPLY
COMMON
Z5
Z7
M3
Z4
Z2
LOAD
Z3
T3
M2
Z1
M4
M1
Q3
Q2
R G2
R3
R G3
D3
R2
D2
D.C. SUPPLY ~
~ 300 to 375 V.D.C.
+VE
SUGGESTED PARTS:
1. U1 : T.I. UC 3875
2. T1,T2 : Coilcraft Part No. SD250-3 or Vanguard Pulse Transformer Part No: GD 203
3. Q1,Q2,Q3,Q4 : 2N2905A
4. D1,D2,D3,D4 : DSEP8-02A IXYS HiPerFRED
5. T3 : OUTPUT Transformer
6. CF : 22MFD,35 VWDC Tantalum Capacitor
7. R1,R2,R3,R4 : 560 Ohms, ¼ w,1% Metal film
8. M1,M2,M3,M4 : IXFN55N50 IXYS HiPerFET or IXFN80N50 IXYS HiPerFET
9. Z1,Z2,...Z8 : 18V, 400mW Zener diodes.
10. RG1,RG2,RG3,RG4 : 3.3 Ohms, ¼ w, 1% Metal Film resistors.
11. R5,R6,R7,R8 : 10K, ¼ w, 5%
Fig. (11) Transformer coupled Gate Drive arrangement for "H" Bridge in a Phase Shift PWM Controller at fixed Switching frequency.
Phase
Shift
Resonant
Controller
IC
U1
A
1
CF
T2
GND1
+5.0 V
ISOLATED
DC TO DC
CONVERTER
GND1
T2
T3
15
3
14
Vcc1
GND1
4
T6 RESET
5
T4
T5
Dynamic
Brake
2
T7
RESET
FAULT
6
+
7
VLED1
OVERTEMP
- 8
VLED1
13
12
R5 RD
11
FAULT
U1
10
U6
9
+15V
VE
COM -5V
+
VLED2+
+
C1
C2
DESAT
IXDD414PI
U2
Vcc2
Vc
ENABLE
Vin-
16
HCPL-316J
I/P
1
Vcc
O/P
TMS320F2407A
Vin+
T1
SineWave
PWM
Signals
for
3-Phase
Inverter
GROUND
+5.0 V
+5.0V
VOUT
VEE
R6
VEE
-5V
GND1
IXYS's (CBI)
CONVERTER
BRAKE
INVERTER
MODULE
LF
22
R4
T1
1
D12
D13
2
D14
D15
D15
CF
3
16
7
15
D16
T5
D3
20
17
19
T2
24
SHUNT
R7
C3
R8
R13
R9
U4
8
R2
9
Z1
D6
13
T4
R3
T6
C4
-
N
T
C
D4
12
11
D5
18
D2
23
R12
T3
D1
14
GND1
U3
Dd
+
21
D11
-
R1
Rg
+5V
R10
+
-
R11
U5
FAULT
+
P1
Fig. (12) 3-Phase AC Motor drive schematic showing how IXYS CBI (Converter-Brake-Inverter)
Module can be driven by IXDD414 using opto-couplers.
All protection features are incorporated.
IXAN0009
IXAN0009
W
V
U
FW
FV
FU
D12
1
GND1
D14
2
D13
D16
3
D15
LF
23
24
HCPL316J
IXDD414
-5V
+5V
Rg
DYNAMIC
BRAKE
REGISTOR
DC t o DC
+15V
SHUNT
T7
R
F
+ CF
22
14
7
T7
T2
R
F
T1
R
F
+15V
+15V
HCP L316J
IXDD414
DC to DC
T1
T2
DESAT
-5V Rg
DESAT
-5V Rg
+5V
HCP L316J
IXDD414
DC to DC
+5V
T3
RD
10
11
T4
T2
RD
15
16
T1
6
T6
T4
R
F
Dd
T3
R
F
Dd
T7
+15V
R
F
HCPL316J
IXDD414
DC to DC
RD
12
T4
RD
17
18
T3
OVERTEMP
DESAT
-5V Rg
DESAT
-5V Rg
+5V
HCPL316J
IXDD414
DC to DC
TMS320F2407A DSP CHIP
T5
D2
D1
+15V
+5V
5
D4
D3
+15V
+5V
T6
R
F
Dd
T5
R
F
Dd
+15V
HCPL316J
IXDD414
DC to DC
DESAT
-5V Rg
DESAT
-5V Rg
+5V
HCPL316J
IXDD414
DC to DC
+5V
FIG(13) IXYS CONVERTER, BRAKE INVERTER (CBI) MODULE BEING DRIVEN BY IXDD414 WITH
OPTO-COUPLER AND DESAT, OVERTEMP AND SHORT CIRCUIT/OVERLOAD PROTECTIONS.
NOTES: 1. ALL F = FAULT SIGNALS ARE TIED TOGETHER (BEING OPEN COLLECTOR) AND FED INTO TMS320F2407A DSP CHIP.
2. ALL R = RESET SIGNALS ARE TIED TOGETHER AND FED TO HCPL-316J.
3. OVERTEMP AND OVERLOAD/SHORT CIRCUIT FAULT SIGNALS ARE GENERATED AS PER FIG(16)
OVERTEMP IS ALSO FED IN TMS320F2407A DSP CHIP .
MCB
3ø MAI NS
D11
21
T5
RD
13
T6
RD
19
20
4
Dd
D6
Dd
M
3 P hase A.C.
MOTOR
D5
Bill of Materials for Fig. (12) and
Fig. (13)
R1, R3, R5, R10, R11: 10K, 1/4W, 1% MFR
R2: 560 Ohms, 1/4W, 1% MFR
R4: 2.2 Meg, 1/4W, 5%
R6: 100 Ohms,1/4 W,1% MFR
R7: 20K, 1/4W, 1% MFR
R8, R9: 61.9K, 1/4W, 1% MFR
R12, R13: 1.24K, 1/4W, 1% MFR
Rg: T.B.D. based on ton and toff & size of IGBT
RD:100 Ohms,1/4 w, 5%
P1: 10K, multi turn trimpot, Bourns 3006P or
Spectrol
C3, C4: 33 pf, silver dipped mica
Dd: General Semiconductor make,
Type: RGP02-20E, 0.5 A, 2000 V, trr: 300 ns
Z1: Zener LM336, 2.5 Volt
U3, U5 : LM339 Comparator
U4 : LM-101 Op Amp
SHUNT : 75 mV @ full load current
LF: Gapped D C Choke for filtering rectified
power
CF: Electrolytic Filter Capacitor with very low
ESR & ESL and screw type terminals to
handle high ripple current. Voltage rating is
determined by DC Voltage plus AC ripple
Voltage
CBI Module: IXYS Corporation Type Nos:
MUBW 50-12A8 or any MUBW module from
CBI 1, CBI 2 or CBI 3 series, depending on
Motor H.P. rating.
U1:
Texas
Instrument’s
TMS320F2407A,
FLASH
programmable
Digital
Signal
Processor with embedded software for AC
Drive, using brake feature.
IXDD414 Driver chip:
7 are required to
implement the AC Drive, using Brake
feature.
HCPL316J (Opto-coupler): 7 are required to
implement the AC Drive With Brake feature.
Isolated DC-t o-DC Converter: 7 are required
with specified isolation.
IXAN0009
IXAN0009
V
V
t
t
+
L
+
I/P
H
I/P
CF
RH
CF
RH
4
3
2
1
4
3
2
1
5
6
7
8
5
6
7
8
R1 C1
D1
Vcc
D1
R1 C1
T1
D2
Z1
C2
D2
Z1
C2
Rp
Rp
Q4
Q1
Q3
H.V.D.C.
COMMON
LOAD
T3
Q2
H.V.D.C.
Rp
Rp
FIG(14) A Transformer coupled Gate Drive circuit employing D.C. restore
technique and showing how to generate -ve bias during turn-off.
I
I
X
X
D
D
D OR D
4
4
0
1
8
4
I
I
X
X
D
D
D OR D
4
4
0
1
8
4
Vcc
D2
Z1
C2
D2
Z1
C2
T2
H. V. AC
INPUT
C1
H. V. AC
INPUT
E1
T1
T1
G1
G1
E1,E2
C1
C2
T2
R5
G2
E2
V
H. V. AC
INPUT
R6
1
I/P
t
V
T2
G2
+15VDC
A
2
250KHz > fsw > 10KHz
I/P
3
B
4
C2 H. V. AC
INPUT
8
I
X
D
D
4
0
4
P
I
T1
G1
7
C1
Z3
6
C1
R1
+
H. V. AC
INPUT
E2
T1
R3
Z4
G2
G1
5
E1
t
T2
E1
T2
R1,R2,R3,R4: 2K2,1/4w,5%
R5,R6:10K,1/4w,5%
C1:47MFD,35WVDC Tantalum Capacitor
T1,T2: Coilcraft Gate Drive Xformer SD250-1 or Vanguard P/N:GD203
Z1,Z2,Z3,Z4: 18V,400mw Zener Diodes
H. V. AC
INPUT
G2
C2
Z1
R2
R4
Z2
T1
H. V. AC
INPUT
H. V. AC
INPUT
E2
E1
G1
Fig (15) A Simple scheme to drive Bi-directional switches, using gate drive transformers
to ensure galvanic isolation.
H. V. AC
INPUT
Vcc +5V
U1
R1
C1
A
DATA
INPUT
TTL or
LSTTL
U3
+
1
8
2
7
2
3
3
4
4
5
8
I
X
D
N
4
0
9
P
I
R3
6
H. V. AC
INPUT
E1
1
C2 -
C1
T1
T1
G1
7
R G1
G1
G1
6
C1
C2
Z1
R5
5
E1,E2
T2
T2
G2
Z2
G2
+15VDC
INPUT
Isolated
DC to DC
Converter
+
OUT
-
+
OUT
-
E1
E2
C3
H. V. AC
INPUT
C2 H. V. AC
INPUT
C4
-5VDC
C1
Vcc
U2
B
DATA
INPUT
C1
LSTTL or
TTL
U4
+
1
8
2
7
R2
6
-
C2
R4
2
3
3
4
5
4
INPUT
OUT
+
+
OUT
E2
T1
8
G1
G2
7
R G2
6
5
G2
R6
Z3
T2
E1
H. V. AC
INPUT
C2
Z4
+15VDC
Isolated
DC to DC
Converter
I
X
D
N
4
0
9
P
I
H. V. AC
INPUT
-
C3
C4
-5VDC
E2
U1,U2: HCPL2201 opto-coupler
C1: 100Pf ceramic
R1,R2: 1K,1/4W,5%
C2: 10MFD,35vdc Tantalum
C3,C4: 47MFD,35VDC Tantalum
R3,R4: 10K,1/4W,5%
RG1,RG2: 1 ohm to 10 ohm depending on
IGBT & risetime desired
Z1,Z2,Z3,Z4: 18V,400mw zener diodes
R5,R6: 2K2,2W,5%
H. V. AC
INPUT
T1
H. V. AC
INPUT
E1
G1
Fig.(16) A Scheme of driving Bi-directional switches with built-in galvanic isolation, using opto-couplers.
IXAN0009
V
V
Rg on
Rg on
+
+15V
Cg s
Rg off
+15V
Rg off
Cg s
t
t
TURN-ON PULSE
TURN-OFF PULSE
Rg on
V
Rg on
V
Rg off
+15V
i on
+
i off
Rg off
Cg s
+15V
Rsc O/L
Sensing
t
Cg s
Rsc O/L
t
TURN-OFF PULSE
TURN-ON PULSE
Fig.(17) Basic circuit showing use of pulse transformer to give isolation for upper
MOSFET/IGBT in a phase leg configuration
R
LF
S7
S4
S1
C
F
C
F
S
LF
S8
S5
S2
C
F
T
LF
S6
S9
S3
3 phase
LOAD
Fig.(18) Basic 3 Phase to 3 Phase matrix converter employing nine Bi-directional switches
IXAN0009