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18.2.2013 15.15
Aalto University publication series
DOCTORAL DISSERTATIONS 40/2013
Impedance matching and tuning of
non-resonant mobile terminal
antennas
Risto Valkonen
A doctoral dissertation completed for the degree of Doctor of
Science (Technology) to be defended, with the permission of the
Aalto University School of Electrical Engineering, at a public
examination held at the lecture hall S1 of the school on 15 March
2013 at 12.
Aalto University
School of Electrical Engineering
Department of Radio Science and Engineering
Supervising professors
Prof. Pertti Vainikainen, until June 30, 2012
Prof. Antti Räisänen, as of July 1, 2012
Thesis advisor
Dr. Sc (Tech.) Clemens Icheln
Preliminary examiners
Dr.-Ing. Marta Martínez-Vázquez, IMST GmbH, Germany
Associate Prof. Buon Kiong Lau, Lund University, Sweden
Opponent
Prof. Anja Skrivervik, École Polytechnique Fédérale de Lausanne,
Switzerland
Aalto University publication series
DOCTORAL DISSERTATIONS 40/2013
© Risto Valkonen
ISBN 978-952-60-5054-6 (printed)
ISBN 978-952-60-5053-9 (pdf)
ISSN-L 1799-4934
ISSN 1799-4934 (printed)
ISSN 1799-4942 (pdf)
http://urn.fi/URN:ISBN:978-952-60-5053-9
Unigrafia Oy
Helsinki 2013
Finland
Abstract
Aalto University, P.O. Box 11000, FI-00076 Aalto www.aalto.fi
Author
Risto Valkonen
Name of the doctoral dissertation
Impedance matching and tuning of non-resonant mobile terminal antennas
Publisher School of Electrical Engineering
Unit Department of Radio Science and Engineering
Series Aalto University publication series DOCTORAL DISSERTATIONS 40/2013
Field of research Radio Engineering
Manuscript submitted 25 October 2012
Date of the defence 15 March 2013
Permission to publish granted (date) 8 January 2013
Language English
Monograph
Article dissertation (summary + original articles)
Abstract
The volume reserved for antennas in a mobile handset is very limited. Furthermore, the laws
of physics state that the efficiency and/or the bandwidth of a small antenna are also limited.
Despite this fact, the antennas of the future mobile terminals should operate efficiently at
increasingly many frequency bands. This doctoral thesis contributes to solving some of the
main challenges arising from the theoretical and practical limitations of handset antennas.
Non-resonant capacitive coupling element (CCE) based antennas are an attractive option
for future mobile handsets, having many useful characteristics such as compact size and broad
frequency tunability. A circuit model improving the physical understanding of the operation of
a capacitive coupling element antenna is developed in the thesis work. This work also advances
the usability of CCE antennas by developing impedance matching and tuning solutions as
enablers of future mobile communication systems, including software defined radio. The
properties of non-self-resonant antennas are utilized in developing novel multi-band antennas
for mobile terminals. A method for implementing simultaneous multi-band operation of singleelement CCE antennas is developed, based on a novel diplexing impedance matching circuit
and a distributed CCE feeding structure. The realized prototypes operate at 4th generation
mobile communication system (LTE-A) frequency bands 698—960 MHz and 1710—2690 MHz,
reaching state-of-the-art performance. This versatile CCE structure is shown to operate
equally well with single- or dual-feed RF front-end interfaces. Broadband tunable antenna
prototypes are realized as a proof of concept regarding CCE tunability. Also design rules for
avoiding excess losses and nonlinearity in tuning circuits are given, based on numerical and
experimental analyses.
An antenna in a mobile terminal is also subject to electromagnetic interaction with the user
of the handset. This interaction deteriorates the operation of the antenna through absorption
of power and detuning of the antenna impedance. In this work, methods for avoiding the
impedance detuning are developed. CCE geometry design and antenna selection diversity are
shown to decrease the impedance mismatch and improve radiation efficiency by up to 4 dB in
total in the case of some typical hand grips. Also matching circuits that result in improved
robustness against the mistuning are studied and developed.
Keywords mobile antennas, capacitive coupling element, impedance matching, frequency
tuning, equivalent circuits, electromagnetic user interaction
ISBN (printed) 978-952-60-5054-6
ISBN (pdf) 978-952-60-5053-9
ISSN-L 1799-4934
ISSN (printed) 1799-4934
ISSN (pdf) 1799-4942
Location of publisher Espoo
Pages 149
Location of printing Helsinki
Year 2013
urn http://urn.fi/URN:ISBN:978-952-60-5053-9
Tiivistelmä
Aalto-yliopisto, PL 11000, 00076 Aalto www.aalto.fi
Tekijä
Risto Valkonen
Väitöskirjan nimi
Mobiilipäätelaitteiden resonoimattomien antennien impedanssisovitus ja –säätö
Julkaisija Sähkötekniikan korkeakoulu
Yksikkö Radiotieteen ja -tekniikan laitos
Sarja Aalto University publication series DOCTORAL DISSERTATIONS 40/2013
Tutkimusala Radiotekniikka
Käsikirjoituksen pvm 25.10.2012
Julkaisuluvan myöntämispäivä 08.01.2013
Monografia
Väitöspäivä 15.03.2013
Kieli Englanti
Yhdistelmäväitöskirja (yhteenveto-osa + erillisartikkelit)
Tiivistelmä
Antenneille varattu tila matkapuhelimissa on erittäin rajattu. Lisäksi fysiikan lakien mukaan
pienen antennin hyötysuhde ja/tai kaistanleveys ovat myös rajoitettuja. Tästä huolimatta
tulevaisuuden mobiilipäätelaitteissa olevien antennien olisi toimittava tehokkaasti yhä
useammalla taajuuskaistalla. Tämä väitöskirja edistää matkapuhelinantennien teoreettisista
ja käytännöllisistä rajoituksista johtuvien haasteiden ratkaisemista.
Resonoimattomaan kapasitiiviseen kytkentäelementtiin (CCE) perustuvat antennit ovat
houkutteleva vaihtoehto tulevaisuuden matkapuhelimiin. Niillä on useita hyödyllisiä
ominaisuuksia, kuten pieni koko ja laaja taajuussäädettävyys. Väitöskirjatyössä on kehitetty
piirimalli, joka parantaa CCE-antennin fysikaalisen toiminnan ymmärtämistä. Tässä työssä
myös edistetään CCE-antennien käytettävyyttä kehittämällä impedanssin sovittamiseen ja
säätöön perustuvia ratkaisuja, jotka mahdollistavat osaltaan tulevaisuuden mobiiliviestintäjärjestelmien, kuten ohjelmistoradion, toteutusta. Resonoimattomien antennien
ominaisuuksia käytetään hyödyksi uusien monitaajuusantennien kehittämisessä
matkapuhelimiin. Usealla taajuuskaistalla samanaikaisesti toimivan CCE-antennin
toteuttamiseksi kehitetään uusi menetelmä, joka perustuu taajuuslomituksen tekevään
impedanssisovituspiiriin ja yhden kytkentäelementin hajautettuun syöttörakenteeseen.
Toteutetut prototyypit toimivat neljännen sukupolven matkapuhelinverkossa (LTE-A)
taajuuksilla 698—960 MHz ja 1710—2690 MHz saavuttaen kehityksen huippua edustavan
suorituskyvyn. Tämä monipuolinen CCE-rakenne toimii yhtä hyvin niin yhdellä kuin kahdella
antennisyötöllä varustetun radiotaajuisen esiasteen kanssa. CCE-antennien
taajuussäädettävyys osoitetaan toimivaksi ratkaisuksi toteuttamalla laajakaistaisesti
säädettäviä antenniprototyyppejä. Lisäksi esitetään numeerisiin ja kokeellisiin analyyseihin
perustuen yksinkertaisia suunnitteluohjeita taajuusvirityspiirin liiallisten tehohäviöiden ja
epälineaarisuuden välttämiseksi.
Matkapuheliantenni on myös altis sähkömagneettiselle vuorovaikutukselle laitteen käyttäjän
kanssa. Tämä vuorovaikutus heikentää antennin toimintaa aiheuttamalla impedanssin
sivuunvirittymistä ja tehon absorboitumista. Tähän ongelmaan on tartuttu tässä työssä
kehittämällä menetelmiä välttää impedanssin sivuunvirittymistä. CCE:n geometristen
yksityiskohtien suunnittelun ja antennielementin valintadiversiteetin osoitetaan vähentävän
käyttäjän käden aiheuttamaa impedanssiepäsovitusta ja parantavan säteilyhyötysuhdetta
yhteensä jopa 4 dB joidenkin tyypillisten käsiotteiden tapauksessa. Sovituspiirejä, jotka
kestävät paremmin impedanssin sivuunvirittymistä on niin ikään tutkittu ja kehitetty.
Avainsanat mobiiliantennit, kapasitiivinen kytkentäelementti, impedanssisovitus,
taajuussäätö, piirimalli, sähkömagneettinen käyttäjävuorovaikutus
ISBN (painettu) 978-952-60-5054-6
ISBN (pdf) 978-952-60-5053-9
ISSN-L 1799-4934
ISSN (painettu) 1799-4934
ISSN (pdf) 1799-4942
Julkaisupaikka Espoo
Sivumäärä 149
Painopaikka Helsinki
Vuosi 2013
urn http://urn.fi/URN:ISBN:978-952-60-5053-9
Preface
My warmest thanks go to my first supervisor, Prof. Pertti Vainikainen,
for allowing me to do interesting research under his guidance, and for
the numerous conversations during which I learned a lot about science
and technology. Prof. Antti Räisänen deserves warm thanks as well, for
helping me through the final phase of the dissertation process. I want to
thank my instructor and boss, Dr. Clemens Icheln, for the constructive
discussions and helpful comments on my work. Many thanks also to Prof.
Cyril Luxey (University of Nice) who instructed my work during my stay
in Nice, France, in 2009, and has co-operated with us for many years.
Close co-workers are invaluable. Thanks to Mr. Janne Ilvonen and Dr.
Jari Holopainen for so many shared ideas, productive discussions, entertaining travel companionship, etc. during the years. In addition to the coauthors of the papers, also other colleagues are important e.g. in creating
a good working atmosphere. Thanks to the current and former colleagues:
Mikko, Juho, Anu, Kaltsi, Katsu, Veli-Matti, Outi, Kimmo, Azremi, Aki,
Aleksi, Pekka, Antti, Olli, Tero, Vasilii, Dmitry, Viktor, Eikka, and everybody at the Department of Radio Science and Engineering. I want
to thank also the project partners, especially Dr. Jani Ollikainen from
Nokia Research Center and Dr. Jussi Rahola from Optenni for valuable
comments and good co-operation.
This thesis work was supported by the GETA Graduate School, as well
as SMARAD (Academy of Finland CoE in Smart Radios and Wireless Research), Jenny ja Antti Wihurin rahasto, and Tekniikan edistämissäätiö
(TES). I am grateful to all of the supporters. Ja lopuksi, kiitokset perheelle tuesta ja huolenpidosta.
Espoo, February 15, 2013,
Risto Valkonen
7
Preface
8
Contents
Preface
7
Contents
9
List of Publications
11
Author’s Contribution
13
List of abbreviations
14
List of symbols
16
1. Introduction
19
1.1 Background . . . . . . . . . . . . . . . . . . . . . . . . . . . .
19
1.2 Objective of this work . . . . . . . . . . . . . . . . . . . . . . .
20
1.3 Contents and organization of the thesis . . . . . . . . . . . .
20
1.4 Main scientific merits . . . . . . . . . . . . . . . . . . . . . . .
22
2. Characterization of mobile terminal antennas at an early
design stage
23
2.1 Physical limitations of small antennas . . . . . . . . . . . . .
24
2.1.1 Historical background . . . . . . . . . . . . . . . . . .
24
2.1.2 Special limitations related to internal mobile terminal antennas . . . . . . . . . . . . . . . . . . . . . . . .
26
2.2 Understanding the non-resonant antenna . . . . . . . . . . .
27
2.2.1 Circuit modeling of capacitive coupling element antennas
. . . . . . . . . . . . . . . . . . . . . . . . . . .
29
2.3 Bandwidth evaluation of mobile terminal antennas . . . . .
31
2.3.1 Quality factor and bandwidth . . . . . . . . . . . . . .
32
2.3.2 Tool for approximating the bandwidth performance .
33
9
Contents
3. Multi-band non-resonant mobile terminal antennas
37
3.1 Bandwidth enhancement based on antenna geometry . . . .
38
3.1.1 Self-resonant antenna elements . . . . . . . . . . . . .
38
3.1.2 Modified radiating ground plane . . . . . . . . . . . .
38
3.1.3 Location of CCE and feeding configuration . . . . . .
40
3.2 Multi-band matching of non-resonant antennas . . . . . . .
41
3.2.1 Background of broadband matching . . . . . . . . . .
41
3.2.2 Dual-branch matching circuits for a CCE antenna . .
42
4. Frequency tunable CCE antennas
47
4.1 Frequency tuning of handset antennas . . . . . . . . . . . . .
47
4.1.1 Enabling technologies . . . . . . . . . . . . . . . . . .
47
4.1.2 Methods for tuning mobile terminal antennas . . . .
49
4.2 Broadband tuning of CCE antennas . . . . . . . . . . . . . .
51
4.2.1 Tunability . . . . . . . . . . . . . . . . . . . . . . . . .
51
4.2.2 CCE antennas with wide tuning range . . . . . . . . .
52
4.3 Challenges set by different impedance environments . . . .
54
4.3.1 Dissipative losses . . . . . . . . . . . . . . . . . . . . .
55
4.3.2 Nonlinear phenomena . . . . . . . . . . . . . . . . . .
56
5. Antenna as a part of a radio system
57
5.1 Mitigation of electromagnetic user interactions . . . . . . . .
58
5.1.1 User effects on mobile handset antennas . . . . . . .
58
5.1.2 Compensation of impedance mismatch . . . . . . . . .
60
5.1.3 Proactive methods against impedance mismatch . . .
61
5.2 Interface between antenna and transceiver . . . . . . . . . .
63
5.2.1 Single-feed and multi-feed architectures . . . . . . . .
63
6. Summary of publications
67
7. Conclusions
71
Bibliography
73
Errata
91
Publications
93
10
List of Publications
This thesis consists of an overview and of the following publications which
are referred to in the text by their Roman numerals.
I J. Holopainen, R. Valkonen, O. Kivekäs, J. Ilvonen, and P. Vainikainen,
”Broadband equivalent circuit model for capacitive coupling element–
based mobile terminal antenna,” IEEE Antennas and Wireless Propagation Letters, Vol. 9, pp. 716–719, 2010.
II R. Valkonen, J. Ilvonen, C. Icheln, and P. Vainikainen, ”Inherently nonresonant mobile terminal antenna with simultaneous multi-band operation,” Electronics Letters, Vol. 49, No. 1, pp.11–13, 2013.
III R. Valkonen, M. Kaltiokallio, and C. Icheln, ”Capacitive coupling element antennas for multi-standard mobile handsets,” IEEE Transactions
on Antennas and Propagation, Accepted for publication in a future issue,
9 p., 2013.
IV R. Valkonen, J. Holopainen, C. Icheln, and P. Vainikainen, ”Minimization of power loss and distortion in a tuning circuit for a mobile terminal
antenna,” In International Symposium on Antennas and Propagation
(ISAP), Taipei, Taiwan, pp. 449-452, October 2008.
V R. Valkonen, C. Luxey, J. Holopainen, C Icheln, and P. Vainikainen,
”Frequency-reconfigurable mobile terminal antenna with MEMS switches,”
In 4th European Conference on Antennas and Propagation (EuCAP),
Barcelona, Spain, 5 p., April 2010.
11
List of Publications
VI R. Valkonen, C. Icheln, and P. Vainikainen, ”Power dissipation in mobile antenna tuning circuits under varying impedance conditions,” IEEE
Antennas and Wireless Propagation Letters, Volume 11, pp. 37–40, 2012.
VII R. Valkonen, J. Ilvonen, K. Rasilainen, J. Holopainen, C. Icheln, P.
Vainikainen, ”Avoiding the interaction between hand and capacitive coupling element based mobile terminal antenna,” In 5th European Conference on Antennas and Propagation (EuCAP), Rome, Italy, pp. 2781–
2785, April 2011.
VIII R. Valkonen, J. Ilvonen, P. Vainikainen, ”Naturally non-selective
handset antennas with good robustness against impedance mistuning,”
In 6th European Conference on Antennas and Propagation (EuCAP),
Prague, Czech Republic, 5 p., March 2012.
12
Author’s Contribution
Publication I: ”Broadband equivalent circuit model for capacitive
coupling element–based mobile terminal antenna”
Dr. Holopainen had the main responsibility of developing the content for
the paper. The original idea was developed by Dr. Holopainen and the
author, and the author also participated in validation of the circuit model
and in writing the paper. Mr. Ilvonen assisted in writing the paper. Dr.
Kivekäs and Prof. Vainikainen supervised the work.
Publication II: ”Inherently non-resonant mobile terminal antenna
with simultaneous multi-band operation”
The paper is based on the author’s original idea, and the author was responsible of the content of the paper. Mr. Ilvonen participated in developing the antenna prototype and assisted in writing the paper. Dr. Icheln
and Prof. Vainikainen supervised the work.
Publication III: ”Capacitive coupling element antennas for
multi-standard mobile handsets”
The author was responsible of developing the idea and content of the paper. Mr. Kaltiokallio participated in the design and implementation of the
tunable antenna and assisted in writing the paper. Dr. Icheln supervised
the work.
13
Author’s Contribution
Publication IV: ”Minimization of power loss and distortion in a
tuning circuit for a mobile terminal antenna”
The author had a leading role in developing the idea and in writing the
paper. Dr. Holopainen developed the measurement system used for the
distortion measurements and had an instructive role in developing the
content for the paper. Dr. Icheln and Prof. Vainikainen supervised the
work.
Publication V: ”Frequency-reconfigurable mobile terminal antenna
with MEMS switches”
The author had the main responsibility of developing the idea and content
for the paper. Prof. Luxey worked as an instructor. Dr. Holopainen assisted in writing the paper. Dr. Icheln and Prof. Vainikainen supervised
the work.
Publication VI: ”Power dissipation in mobile antenna tuning circuits
under varying impedance conditions”
The idea and content of the paper are based on long term research work
of the author. Dr. Icheln and Prof. Vainikainen supervised the work.
Publication VII: ”Avoiding the interaction between hand and
capacitive coupling element based mobile terminal antenna”
The author had the main responsibility in developing the idea and content
for the paper. Mr. Ilvonen and Mr. Rasilainen assisted in measurements
and in writing the paper. Dr. Holopainen assisted in writing the paper.
Dr. Icheln and Prof. Vainikainen supervised the work.
Publication VIII: ”Naturally non-selective handset antennas with
good robustness against impedance mistuning”
The idea and content of the paper are mainly based on the author’s work.
Mr. Ilvonen participated in implementing the antenna prototypes and in
writing the paper. Prof. Vainikainen supervised the work.
14
List of abbreviations
2G
2nd generation mobile communication systems
3G
3rd generation mobile communication systems
4G
4th generation mobile communication systems
3GPP
3rd Generation Partnership Project
ANSI
American National Standards Institute
ATU
Antenna tuning unit
BAW
Bulk acoustic wave
BST
Barium strontium titanate
CA
Carrier aggregation
CCE
Capacitive coupling element
CMOS
Complementary metal-oxide-semiconductor
DF
Dual-feed (antenna)
DTV
Digital television
DVB-H
Digital Video Broadcasting – Handheld
EM
Electromagnetic
EMC
Electromagnetic compatibility
E-UTRA
Evolved UMTS terrestrial radio access
FDD
Frequency division duplex
FET
Field-effect transistor
GaAs
Gallium arsenide
GSM
Global system for mobile communications
HEMT
High electron mobility transistor
IC
Integrated circuit
ICNIRP
International Committee on Non-Ionizing Radiation
Protection
IEEE
Institute of Electrical and Electronics Engineers
ISM
Industrial, scientific and medical (radio band)
LTCC
Low temperature co-fired ceramics
LTE
Long term evolution communication systems
15
List of abbreviations
16
LTE-A
LTE advanced
MEMS
Microelectromechanical systems
MIMO
Multiple input, multiple output
PA
Power amplifier
PCB
Printed circuit board
PIFA
Planar inverted-F antenna
RF
Radio frequencies
RFIC
Radio frequency integrated circuit
RLC
Resistance, inductance and capacitance
RX
Receiver
SAR
Specific absorption rate
SAW
Surface acoustic wave
SDR
Software defined radio
SF
Single-feed (antenna)
SPDT
Single-pole, double-throw
SPMT
Single-pole, multi-throw
TX
Transmitter
UHF
Ultra-high frequencies (300–3000 MHz)
UMTS
Universal Mobile Telecommunications System
UTRA
Universal Terrestrial Radio Access
VSWR
Voltage standing wave ratio
WLAN
Wireless local area network
List of symbols
a
Radius describing maximum antenna dimension
BWr
Relative bandwidth
BWlow
Bandwidth at lower frequency band
BWhigh
Bandwidth at higher frequency band
BWsym
Symmetrical bandwidth
C
Capacitance
f
Frequency
f0
Matching frequency
IIP3
Third order input intercept point
IMD3
Third order intermodulation distortion
j
Imaginary unit
k
Wave number
L
Inductance
Lretn
Return loss
Pr
Radiated power
Q
(Radiation) quality factor
QZ
Quality factor calculated from input impedance
Ra
Antenna resistance
S
Maximum allowed VSWR
T
Coupling coefficient
W
Energy
We
Electric field energy
Wm
Magnetic field energy
X
Reactance
Xa
Antenna reactance
Z
Impedance
17
List of symbols
18
Za
Input impedance of an antenna
Z0
Characteristic impedance
η
Efficiency
ηrad
Radiation efficiency
ηtot
Total efficiency
λ
Wavelength
1. Introduction
1.1
Background
The importance of the antennas in mobile (cellular) radio handsets increased around the beginning of the 21st century. The number of supported radio systems and other features implemented in the handset devices increased while the consumers demanded smaller and smaller mobile phones. The traditional external handset antennas were replaced
by internal ones. The challenge of making compact and efficient multiband internal antennas resulted in a research boost which produced numerous theses, patents, and other scientific publications, as well as new
solutions in the consumer products. Compact planar and conformal antennas capable of multi-band and multi-system operation, such as the
planar inverted-F antenna (PIFA) derived from the microstrip patch antenna have been very popular and successful in handling the challenges
so far [1–4].
One remaining challenge in antenna design for mobile terminals is that
the number of radio systems and frequency bands is still constantly increasing [5]. Each of the radio systems needs an antenna while the space
allocated for antennas in a mobile terminal is limited. According to some
visions, future mobile handsets might include up to 20 different antennas and radio wave sensors for different communication, positioning and
broadcast systems across a wide variety of frequencies [6]. In order to
succeed in implementing all the radio systems in the mobile user equipment, new antenna structures are required to replace or to complement
the traditional antennas in mobile handsets.
The work for this thesis started from the author’s Master’s Thesis in
2007 [7], when the shrinkage of the mobile terminals was about to come
19
Introduction
to an end. The average length and width of the mobile handset itself have
actually increased during the last few years after the introduction of the
touch screen technology and a demand for larger displays capable of viewing high quality video, etc. The antenna size in the terminal has, however,
not increased. One reason for this is that the battery size has grown due
to higher power consumption of the device, and also the average thickness
of the handsets has decreased.
Mobile communications is a rapidly changing industry, and its products
have short life cycles. The future of mobile handsets might be very different from the current situation, and the changes might come sooner than
we can imagine. Although the platform for the antennas keeps changing,
the challenges related to the antennas will remain, thanks to the laws of
physics. The challenges motivate also scientific academic research in the
field for years to come. The perfect antenna always remains undiscovered.
1.2
Objective of this work
The ideal antenna for mobile terminals is a diminishingly small structure
which enables the radio to operate at all possible frequencies. This ideal
unfortunately contradicts with the laws of physics, but if we compromise
a little and accept, for example, that the radio works instantaneously at
a limited number of frequencies, which can then be adjusted, we get one
step closer to this ideal. The main objective of this thesis is to develop
practical communication antennas for mobile handsets towards this ideal
by studying and developing methods which allow the use of very compact
and efficient, frequency agile antennas.
Multi-antenna systems, e.g. MIMO (Multiple input, multiple output)
are left out of the scope of this thesis. However, the results obtained in
this thesis with single element antennas can be utilized in designing individual antenna elements with their matching circuits for mobile handsets
supporting MIMO operation required by future communication systems.
1.3
Contents and organization of the thesis
A certain type of non-self-resonant antenna elements, called the capacitive coupling elements (CCE) [8–10], are used as the basis for the research, as they have some advantageous characteristics compared to tra-
20
Introduction
ditional self-resonant antennas. The properties of a CCE include for example a simple geometrical structure, compact size and non-selective frequency response. An antenna based on a CCE consists of the coupling
element, a ground plane and an impedance matching circuit. This combination is called a CCE antenna.
In this thesis, CCE antennas are studied with impedance matching and
tuning in focus. Electromagnetic (EM) and circuit simulations, as well as
experimental studies including prototyping and antenna measurements,
are the methods used to obtain the results. The main scientific contribution of the thesis is found in the publications [I - VIII], and this summary
part collects the results, providing also references to other relevant research made in the related fields.
In Chapter 2, the fundamental challenges related to small antennas,
and handset antennas in general, are recapitulated. A broadband equivalent circuit model of a CCE antenna is developed [I] in order to improve
general understanding on the operation of the CCE antennas, thereby
helping to design better ones. A tool for bandwidth characterization of
especially CCE antennas is also briefly described, as it is utilized in developing some of the main results of the thesis [II, III], and illustrates the
motivation to use non-resonant antenna elements.
The versatility of CCE antennas is illustrated in Chapter 3 by designing antennas capable of operating simultaneously at multiple frequency
bands [III, II, VIII], and in Chapter 4 with antennas having broad tuning
ranges [III, IV, V]. These antennas are realized with the help of commercially available and inexpensive impedance matching and tuning components. When using lossy and/or nonlinear circuit elements, it is important to detect the practical limitations related to the performance of the
circuits. Power dissipation in tuning circuits of mobile terminal antennas
is studied [VI], taking into account the varying impedance environment
in the circuits. Nonlinearity issues of transistor-based RF switches used
in tuning circuits are addressed in [IV].
The effect of the user on the performance of a mobile terminal antenna is
usually unwanted, but also unavoidable, and not negligible in the design.
Methods for decreasing or partly avoiding the user-originated detuning
of the impedance of CCE antennas are presented in [VII, VIII], validated
with experimental results. Based on these results, Chapter 5 discusses
the interaction between the user and the mobile handset antenna. Also
the radio transceiver interface of the antenna is discussed with examples
21
Introduction
in order to acknowledge the fact that an antenna is always just one part
of the radio communication system [III].
1.4
Main scientific merits
The novel scientific results obtained in this thesis are related to the methods developed to improve CCE antenna operation, as well as to the implemented antenna prototypes which exceed the prior art of CCE antennas
for mobile communications, and also exhibit top performance in the field
of handset antennas in general. Regarding the contribution of the thesis
to the scientific community, the highlights of the thesis are:
1. Novel dual-branch impedance matching circuit design with a distributed
feeding structure facilitating a single-CCE antenna operation simultaneously at two broad, noncontiguous frequency bands. The realized
prototypes are operational across the LTE-A communication system frequency bands 698-960 MHz and 1710-2690 MHz.
2. An equivalent circuit model providing an accurate behavioral model for
the antenna impedance and improving the general understanding of the
physical operation of the CCE antenna.
3. Proving broadband tunability of CCE antennas in practice with antenna prototypes, based on both frequency band switching and continuous impedance tuning.
4. First critical analysis on the power losses of the handset antenna tuning circuit in varied and varying impedance conditions.
5. Evidence provided on the significance of the CCE and matching circuit
design on the antenna performance in the proximity of a user’s hand.
22
2. Characterization of mobile terminal
antennas at an early design stage
Traditional mobile terminal antennas are usually self-resonant at their
frequencies of operation. The resonances are created within the antenna
geometry and result in good impedance match between the antenna and
its feed, without additional matching circuits. The input impedance properties of a simulated self-resonant antenna design can be seen directly
from the electromagnetic simulation results (S-parameters), and the design can be improved by changing the antenna geometry resulting in a
better impedance match between the antenna and its feed [1]. With modern EM simulators, it is easy to make small modifications to the geometry
of an antenna design and to get quick feedback on the effects.
Non-resonant antennas, on the other hand, have to be matched with external circuits in order to realize satisfactory impedance matching, and
thus the simulated impedance does not directly reveal the potential performance of the antenna. Since the non-resonant antenna does not work
well without impedance matching, the simulated antenna would need to
be matched at each frequency band in order to evaluate its operation. Circuit simulations are needed for this purpose. Although the inclusion of circuit simulations makes the antenna design more complex, it is worthwhile
to consider non-resonant antennas in mobile terminals. Self-resonant antennas may suffer from limitations, such as limited tunability, which can
be overcome by using non-resonant antennas (Section 2.3).
The rather unorthodox approach of using non-resonant antennas is justified by explaining the operation of a capacitive coupling element antenna through equivalent circuit modeling based on the physical behavior
of the antenna in Section 2.2. Before that, the physical limitations of
small antennas are recapitulated in Section 2.1. These limitations relate
to all antennas, resonant or not, illustrating the problem of implementing
a well performing, broadband small antenna. The concept of bandwidth
23
Characterization of mobile terminal antennas at an early design stage
potential which was coined by Villanen et al. [10] is utilized in this thesis
work as a tool to support the design of inherently non-resonant mobile
terminal antennas, and to illustrate some main differences between nonresonant and resonant antennas in Section 2.3.
2.1
2.1.1
Physical limitations of small antennas
Historical background
The laws of physics dictate the performance of an electrically small antenna in terms of bandwidth, size and gain (or efficiency). The smaller
the physical dimensions of an antenna are compared to the wavelength λ
of the radiated signal, the harder it is for the antenna to radiate efficiently
over a wide frequency band [11–13].
An antenna is considered electrically small if the greatest dimension of
the antenna is small compared to λ. The wavelength is often expressed
embedded in the wave number k = 2π/λ, and the generally used quantity describing the antenna dimensions is a, the minimum radius of an
imaginary sphere surrounding the antenna [11]. A generally accepted
definition for a small antenna is that it satisfies ka < 1 [14]. Some require
ka ≤ 0.5 [15] or that ka is ”much smaller” than unity [16]. The product
ka is a convenient way to express the size of the antenna relative to the
wavelength. It allows direct comparison of antennas with similar electrical size, disregarding the frequency.
The fundamental limitations with respect to the physical size, maximum bandwidth or gain, and minimum quality factor Q of small antennas
have been a topic of numerous studies since the ground-breaking work of
Wheeler [12, 16] and Chu [11] in the 1940’s and 50’s. The main finding
was that there is a lower bound for the radiation quality factor of an electrically small antenna, limiting the bandwidth and/or efficiency of the antenna. In the following decades, many authors gave their contributions,
further specifying the limits of the radiation Q and the gain-bandwidth
product [13, 14, 17–21].
Revisions for the minimum Q, or the ”Chu limit” after [11], have been a
popular scientific topic also in the 2000’s, with a new point of view. The
earlier research was concentrating on a general idealized small antenna
which does not store any energy within the smallest possible spherical
24
Characterization of mobile terminal antennas at an early design stage
volume surrounding the antenna structure (excluding the antenna itself).
In practice, an antenna usually stores energy within the sphere [22], and
thus the earlier Q limits for practical antennas were usually too optimistic
[23]. Thereby new revisions of the Q limits have been recently published
[15, 22, 24], also for antennas of arbitrary shape [25, 26]. These limits
are never lower, but instead usually higher than the original Chu limit.
The limits for antennas occupying different shapes have also been called
Gustafsson limits after their author [27].
The exact realized radiation Q of an antenna tuned to have zero reactance at frequency f0 is defined by [11], [28]
Q(f0 ) =
2πf0 |W |
,
Pr
(2.1)
where W is the average energy stored within the antenna structure and
Pr is the total radiated power. If the antenna is not tuned to resonance,
the Q can still be defined by replacing W in (2.1) by 2 max(We , Wm ), where
We and Wm are the average stored electric and magnetic energies, respectively [14].
The radiation Q is an important quantity since it is inversely proportional to the bandwidth of an antenna, at least when Q 1, [29]. Comparison of the exact Q to the minimum Q of an antenna having similar
dimensions has been used in characterization of the performance of small
antennas. The closer the Q value is to the fundamental limit, the closer
the antenna is to optimal performance. On the other hand, determining Q
exactly is not very straightforward, since it involves the determination of
stored electric and magnetic field energies within the antenna structure,
which in practice have to be calculated indirectly [28].
An easier way to approximate the quality factor of a small antenna at
frequency f0 is to calculate it directly from the input impedance Za (f ) =
Ra (f ) + jXa (f ) of the antenna [28]:
2πf0
QZ (f0 ) ≈
2Ra (f0 )
Ra (f0 )2 + Xa (f0 ) +
|Xa (f0 )|
2πf0
2
,
(2.2)
where the subscript Z is used to express that the Q is calculated from the
input impedance data. The problem with QZ is that generally it is not accurate if the antenna is not electrically small or if it is multi-resonant [30].
Actually, it is easy to show that QZ of an antenna can be made zero [31],
which is not possible for the exact Q defined by (2.1). This is because the
equation for QZ (2.2) involves a linear approximation assuming that the
second and higher order frequency derivatives of the antenna impedance
are zero, which is not generally true for antennas.
25
Characterization of mobile terminal antennas at an early design stage
Despite this, QZ is usually valid for single-resonant small antennas, and
can be used sometimes even if the antenna is not electrically small or
strictly single-resonant [30, 32]. In these cases the validity of QZ has to
be checked case-specifically.
2.1.2
Special limitations related to internal mobile terminal
antennas
The basic platform for a mobile handset antenna consists of the metallic
chassis of the handset: ground planes, EMC shields etc. The width and
the length of the chassis typically correspond to the respective dimensions of the handset excluding the (usually plastic) cover. At the lower
frequencies of mobile communications, especially below 1 GHz, the chassis actually operates as the main radiating element in the handset [8].
This applies to both resonant and non-resonant handset antennas utilizing the chassis as a ground plane [8]. At these frequencies, the chassis
improves the operation of the antenna significantly in terms of bandwidth
by increasing the electrical size (ka) of the radiating structure [8].
The chassis should thus be considered as part of the antenna, therefore making it very difficult for typical mobile terminal antennas to reach
the Chu limit, or even the Gustafsson limit. In practice, the geometry of
the chassis cannot be freely optimized in order to obtain the minimum
Q. Accordingly, it is often misleading to compare the Q of a mobile terminal antenna to the lower bound given by the fundamental limitations
because a mobile terminal antenna can unlikely exploit the chassis radiation optimally, and thus the antenna performance seems inevitably poor
as compared to many different antennas of similar electrical size.
In addition to the laws of physics, there are several other practical limitations which constrict the design of a mobile terminal antenna. Firstly,
the antenna designer has to acknowledge that the antenna is a part of a
radio system. The interoperability with other parts of the system is important, as well as electromagnetic compatibility (EMC) in general. For
example, the antenna should have good impedance matching properties
not only for good efficiency, but also because the transmitter electronics
may suffer if the impedance mismatch causes a strong standing wave.
The standards of each communication system set requirements also with
respect to noise, linearity, maximum transmitter power, etc. The 3G Partnership Project (3GPP), which is a cooperation body of companies and
institutes in the field of mobile communications, prepares standards for
26
Characterization of mobile terminal antennas at an early design stage
the mobile communication systems [33].
Since the mobile terminal is also a consumer product, radiation safety
regulations limit the radiation exposure of the end users of the mobile
terminal. The International Committee on Non-Ionizing Radiation Protection (ICNIRP) has published guidelines for the exposure limits [34],
which are followed widely in Europe [35]. The United States and some
other countries follow the slightly different IEEE standard [36].
Several other design aspects may affect the antenna design as well. As
said, the available volume for the antennas in a mobile handset is always
limited. The shape and the location of the antenna within the handset
may be predetermined due to reasons related to the general handset design. For example a microphone or an external data interface connector
may reserve some space even in the middle of the antenna volume [37].
The available selection of materials might be limited due to cost efficiency,
mechanical durability or appearance. The presence of other electronics
and lossy materials in the device can limit the antenna efficiency. A mobile terminal antenna intended for a real product should thus not be designed as an independent module. In academic research of mobile antennas the requirements and limitations are not as strict as in industrial
product development, but the main constraints should not be ignored.
2.2
Understanding the non-resonant antenna
Even a ”non-resonant” antenna resonates at some frequency, like any
metallic structure. In this thesis, the non-resonant antenna means a radiating structure operating at frequencies that are much lower than the resonance frequency of the lowest-order wavemode of the antenna element.
Of course, the antenna can be operated at this first resonance frequency
or at any higher frequencies as well. Anyhow, the resonances might be
so weak that the antenna would not perform well as such even at these
frequencies. Additional impedance matching circuits are thus needed to
make the non-resonant antennas operate as desired.
A generic example of a CCE antenna consisting of a coupling element
and a compact mobile handset chassis, without a matching circuit, is
shown in Figure 2.1a. Its simulated input impedance at the frequency
range 0.3 – 3 GHz is presented in Figure 2.1b with real and imaginary
parts separated. At low frequencies, the impedance is highly capacitive.
The zero reactance occurs near the first resonance of the coupling element
27
Characterization of mobile terminal antennas at an early design stage
Reactance (Ω)
0
11
Feed point
6.6
nd
-100
-200
-300
100
40
-400
20
18
16
chassis 14
2
12
order
10
wave8
mode
6
4
IE3D simulation
equivalent circuit 2
0
1
1.5
2
2.5
3
Frequency (GHz)
(b)
st
chassis 1 order wavemode
0.5
(a)
2
3
2.5
0.5
2.5
Resistance (Ω)
100
3
1.5
2
1
0.92
0.5
1.5
0.92
1
Za
IE3D sim.
Lretn = 6 dB
(c)
(d)
Figure 2.1. (a) A generic CCE antenna with dimensions in mm. (b) The input resistance and reactance comparison of the EM simulation and the circuit model
of the CCE antenna from [I]. (c) Simulated input impedance of the antenna
on Smith chart with circular frequency markers every 500 MHz between 0.5–
3 GHz relative to 50 Ω. (d) Simulated input impedance of the CCE antenna
matched at f0 = 0.92 GHz with an optimal L-section matching circuit consisting of two inductors. Matching frequency is marked with ’x’.
at about 1.78 GHz, and at higher frequencies the antenna impedance is
on the inductive side. The radiation resistance of the radiating structure
is not very close to zero except for very low frequencies where the combination of the CCE and the ground plane is electrically very short. The
frequency derivatives of the input resistance and reactance are not very
high, except for the derivative of the reactance at low frequencies. The
maxima of the resistance occur at resonance frequencies of the chassis.
Regarding antenna selectivity, (2.2) reveals that a low input resistance
Ra and high derivatives of the real and imaginary parts of the impedance
at a given frequency indicate high Q, and thus high selectivity of an antenna. This is generally true, even though (2.2) does not always accurately predict the achievable antenna bandwidth, as will be discussed in
the next section. Thereby we can assume that the CCE antenna is not
very selective. The lack of inherent selectivity means that the antenna
can be matched at most of the frequencies, although the instantaneous
bandwidth may become relatively narrow.
28
Characterization of mobile terminal antennas at an early design stage
The same antenna impedance trace is presented in complex form on
the Smith chart in Figure 2.1c. The dashed circle on the plot presents a
threshold inside which the antenna would be sufficiently well matched.
In this case, the antenna is not well matched at any frequency without a
matching circuit. An example matching case for the GSM900 frequency
band (880-960 MHz) is shown with the matched impedance trace in Figure 2.1d. The center frequency of the band, 920 MHz, is marked with
an ’x’. The input impedance of the unmatched CCE (Figure 2.1c) is used
as the antenna load Za , which is then matched with an L-section circuit
consisting of two inductors. The resulting impedance trace makes a loop
surrounding the center of the Smith chart. Only now, after matching, the
CCE antenna becomes frequency selective, having low input resistance at
most frequencies outside the desired matched frequency band.
2.2.1
Circuit modeling of capacitive coupling element antennas
Equivalent circuit modeling offers one way to understand and analyze the
operation of antennas. The circuit model can be a behavioral model, in
which case the model mainly helps in expressing some specific features of
the antenna in a simplified way, such as the input impedance with closedform equations. The model can also be based on the physical operation
of the antenna, which gives more information about the operation of the
antenna, but might be less accurate with respect to the input impedance.
The approach based on physical operation is more useful for improving the
understanding of the antenna operation, and finally in helping to improve
the design.
The antenna supports certain resonant wavemodes, which results in
frequency selectivity of the antenna. By analogy to circuit theory, the
wavemodes correspond to resonators, or filters [11], which can be used
in equivalent circuit modeling of the antenna. Although the wavemodes
themselves are orthogonal and thus independent, the effect of the wavemodes is combined in the antenna. Ideally, all wavemodes of the antenna
together with a full coupling matrix should be modeled in order to guarantee the best possible accuracy. In practice, at a limited frequency range
the modeling of a few lowest wavemodes is sufficient. The theory of characteristic wavemodes [38] can be used to detect the individual wavemodes
and to compute their parameters.
The metallic chassis of the handset supports a variety of wavemodes,
and so does the antenna element [39–41]. The feeding configuration and
29
Characterization of mobile terminal antennas at an early design stage
CCE feed
matching
circuitry Cp
Rmm(f)=R0(f/fr,mm)
Rmm(f)
Lmm
2
Cmm
CCE monopole
mode
kc1:1
Cc1
Cc2
kc2:1
Lc1
Lc2
Rc1
Rc2
chassis half-wave
mode
chassis full-wave
mode
Figure 2.2. Equivalent circuit topology for the simple CCE antenna from Figure 2.1, [I].
the location of the antenna element on the chassis determine which of the
wavemodes are excited and which are not [42–44]. Furthermore, some of
the characteristic modes are radiating if excited, and some are not [42].
The inclusion of the chassis wavemodes in the circuit model is important,
which was shown in [8] with an analysis of a PIFA. The relative contribution of a typical size mobile terminal chassis (40 × 110 mm2 ) on the radiation was dominant (90%) at 900 MHz, and about 50 % at 1800 MHz [8].
In the case of non-resonant mobile terminal antennas, the effect of the
chassis size is even more important. The lowest-order wavemode resonance frequency comes from the chassis resonance, and the resonance(s)
of the antenna element may be weak. In this thesis work [I], an equivalent circuit for the simple generic CCE antenna shown in Figure 2.1a was
developed. The model is based on the coupled resonators approach, resulting in not only behavioral correspondence with the input impedance of an
actual CCE antenna but giving also physical insight on the contribution
of each part of the structure on the overall antenna performance.
The broadband equivalent model, intended to cover frequencies 0.3 – 3
GHz, is shown as a schematic in Figure 2.2. It consists essentially of three
resonators: two lowest-order major-axis wavemodes of the chassis and
one mode of the CCE. Modeling the chassis resonators in a cascade configuration with the CCE resonator was a practical choice. Consequently,
the model can be expressed in a compact form, and the coupling between
the wavemodes can be expressed in a straightforward manner with ideal
transformers, like in [8]. Alternatively, the coupling could be expressed
for example with the help of impedance inverters [45].
The models for the chassis resonances in [I] were derived from the theory of characteristic wavemodes, using the information of resonance frequencies and quality factors of each wavemode, after [46]. The relevant
wavemodes of the chassis in the case of the CCE antenna were the two
lowest major-axis modes, which are excited efficiently by the CCE feed located at the end of the chassis. The other chassis wavemodes do not affect
the antenna operation significantly below 3 GHz.
30
Characterization of mobile terminal antennas at an early design stage
The CCE itself resembles a short monopole at low frequencies, and its
radiation resistance is approximately proportional to the square of the
frequency. The CCE is thus modeled with a resonator describing the
monopole mode with resonance at 1.78 GHz [I]. The model represents
well the impedance behavior of the CCE antenna, shown by a comparison
with the EM simulated impedance in Figure 2.1b.
The presented model was used in [I] to analyze the effect of the chassis
wavemodes on the radiation, input impedance and the bandwidth potential of the CCE antenna. It is evident from the results that the CCE needs
the presence of the chassis to be operational at the frequencies of mobile
communications. Secondly, the significance of the chassis on the operation
of the antenna is as high or even higher than in the case of self-resonant
antennas. On the other hand, the CCE monopole mode dominates the input reactance behavior of the antenna. In terms of bandwidth, the chassis
facilitates the use of the CCE antenna at frequencies much below the first
CCE resonance frequency.
In [47], the same equivalent model was used to study qualitatively the
mechanisms with which the proximity of the user of the mobile terminal affects the antenna performance in terms of impedance detuning and
loss of efficiency. The model proved to be useful in explaining various
phenomena occurring when dielectric material is placed near the CCE
antenna. For example, dielectric loading next to the CCE affected both
the CCE monopole mode and the chassis modes, causing decrease of the
resonance frequency and deterioration of the impedance matching level of
the matched antenna, in a way predicted by the model [47].
Although the presented equivalent model explains and predicts well
some physical phenomena related to the non-resonant CCE antenna, it
does not replace the circuit simulations and it is not intended to be an
alternative of computing the characteristic wavemodes. Instead, it is a
complementary tool extending the general understanding of the CCE antenna operation.
2.3
Bandwidth evaluation of mobile terminal antennas
For a compact monoblock type mobile terminal chassis with an internal
antenna (total dimensions 100 × 40 × 10 mm3 ), the ka product is unity at
f ≈ 870 MHz. Contemporary smart phone devices are slightly longer and
wider than the monoblock, so the limit of being electrically small occurs
31
Characterization of mobile terminal antennas at an early design stage
for them even at a lower frequency. No matter how small the antenna element is, the antenna cannot be considered electrically small at the mobile
communications frequencies if, and when, the chassis contributes significantly to radiation. Despite this fact, mobile terminal antennas are often
categorized as ”small antennas” [4], and consequently, also characterized
as small antennas. An example of this is the use of QZ calculated with
(2.2) to evaluate the performance of mobile terminal antennas which are
not electrically small, e.g. [48, 49].
2.3.1
Quality factor and bandwidth
The half-power bandwidth of a simple series or parallel resonant RLC
circuit is simply the inverse of the quality factor. The input impedance
and Q of an antenna have a similar dependence if the input impedance of
the antenna can be modeled with such a circuit. The relative bandwidth
given by the Q, using a predefined matching criterion VSWR < S and a
coupling coefficient T is [50]
1
BWr =
Q
(T S − 1)(S − T )
.
S
(2.3)
The maximum bandwidth, which can be reached by adding infinitely many
lossless matching resonators to the original RLC circuit, is also inversely
proportional to Q [51]. Possible problems arise if the antenna does not
fit the presumed simple model. In special cases, these problems can be
avoided by assuming another model. For example, the relation between
the quality factor of a dual resonant antenna and its bandwidth has been
derived in [52] and [53].
In the case of non-resonant antennas such as the CCE antenna, the
bandwidth estimation by using QZ (2.2) and (2.3) can be overoptimistic.
This can be understood for example by looking at the equivalent circuit of
the CCE antenna and its impedance trace given in Section 2.2. The second
frequency derivative of the antenna impedance is clearly far from zero
around the first chassis resonance, and thus QZ is inevitably inaccurate.
Consequently, (2.3) cannot give accurate results either. Instead of QZ , the
following method which cannot overestimate the achievable bandwidth
was used in evaluation of the CCE antennas in this thesis.
32
Characterization of mobile terminal antennas at an early design stage
2.3.2
Tool for approximating the bandwidth performance
An alternative method for determining the bandwidth, a concept called
”bandwidth potential”, was first used in [10], and later e.g. in [54]. It is
based on computationally matching the input impedance of an antenna
with so called L-section matching circuits at different frequencies. An Lsection is a two-component matching circuit consisting of two reactances
in L-shape configuration: A series or shunt inductor or capacitor with a
shunt or series inductor or capacitor [55]. The L section is used because
it is the simplest general matching topology which can match an arbitrary load impedance having a positive real part to a generator impedance
(which also needs to have a positive real part). To obtain the bandwidth
potential, the L-section matching is applied at each frequency of interest,
and the achievable relative bandwidths as a function of frequency then
constitute the bandwidth potential. In this thesis work the concept is
used as a supporting tool in optimization of CCE designs [III].
Here we define the relative bandwidth BWr (f0 ) to be the maximum symmetrical frequency band around frequency f0 which satisfies the used
impedance matching criterion defined by return loss (Lretn ) or voltage
standing wave ratio (VSWR). This way, the bandwidth potential is defined
and unambiguous for every frequency [56]. For mobile terminal antennas
a typical matching criterion is Lretn ≥ 6 dB, or equivalently, VSWR ≤ 3.
In addition, we require the L-section to be the optimal L-section which
maximizes the symmetrical bandwidth around f0 , even if the antenna is
not perfectly matched at f0 .
An illustration of the optimal L-section matched symmetrical bandwidth
BWsym is seen in Figure 2.3a, where the colors on the Smith chart indicate
the symmetrical bandwidth of the CCE antenna of Figure 2.1 around f0
= 920 MHz achieved by adjusting the antenna impedance at f0 to a given
impedance value relative to 50 Ω. The optimal L-section (Figure 2.1d)
happens to consist of two inductors, and the impedance of the matched
antenna is slightly overcoupled, which is typical for small antennas [50].
The BWsym as a function of matching frequency constitutes the bandwidth
potential (Figure 2.3b). E.g. at 920 MHz, max(BWsym ) ≈ 11 %.
Next we exemplify the utilization of the bandwidth potential concept
in early stage characterization of CCE antennas. Figure 2.4 shows two
similar CCE antennas with slightly different feeding configurations. The
CCE antenna in Figure 2.4a has a narrow feeding strip located along the
33
Characterization of mobile terminal antennas at an early design stage
BWsym (%)
0.35
10
6
4
Relative bandwidth
8
0.3
0.25
0.2
0.15
0.1
0.05
2
0
0
0.5
1
(a)
1.5
2
Frequency (GHz)
2.5
3
(b)
Figure 2.3. (a): Available relative symmetrical 6 dB return loss bandwidth of the simple
CCE antenna of Figure 2.1 around f0 = 0.92 GHz as a function of the matched
antenna impedance at f0 , using L-section matching [56]. (b): Bandwidth
potential of the CCE antenna constructed from the maximum symmetrical
bandwidths as a function of frequency, with 0.92 GHz value marked with ’x’.
5
Centered feeding strip
Offset feeding strip
6
5
50
10
100
(a)
(b)
Figure 2.4. A simple CCE antenna on a typical smartphone chassis with two different
feeding configurations [III]. (a): Centered narrow feeding strip. (b): Offset
feeding strip with an opening angle. The dimensions are in mm. The dashed
line shows the location of the long axis of the chassis.
long axis of the chassis, while the antenna in Figure 2.4b has a trapezoidal
shape feeding strip located 6 mm offset from the same axis.
Depending on the feeding strip location and shape, the antennas have
quite different behavior at some frequencies. Figure 2.5 shows on the
Smith chart the input impedance of these two unmatched antennas at
frequency range 0.5–3 GHz. Both antennas have moderate impedance
matching levels across the entire frequency band. Both antennas also
excite the first order chassis resonance, which is seen as bumps in the
impedance traces at 1.1 GHz. Another small bump in the impedance curve
of the center-fed CCE is seen at 2.5 GHz where the second order chassis
resonance occurs. In the case of the offset-fed CCE this bump is seen as
a loop in the impedance trace, since the offset-fed CCE happens to have
a self-resonance at 2.7 GHz, increasing the coupling to the chassis. It
has to be noted here that a resonance of the chassis or a resonance of the
CCE alone does not indicate zero reactance of the antenna, which is a
combination of these two parts [I].
Figure 2.5 shows also the bandwidth potential of these two antennas.
34
Achievable relative bandwidth
with L section matching
Characterization of mobile terminal antennas at an early design stage
Center feeding strip
Offset feeding strip
0.8
0.6
0.4
0.2
0
0.5
1
2.5
2
1.5
Center frequency (GHz)
3
5 5
5
50
10
5
100
50
5
10
50
100
Achievable relative bandwidth
with L-section matching
Figure 2.5. Input impedance trace at frequencies 0.5–3 GHz and bandwidth potential
(Lretn ≥ 6 dB) of the CCE antennas in Figure 2.4 with the two different
feeding configurations. The impedance travels clockwise as a function of frequency, frequency markers are shown at 0.5 GHz intervals.
0.8
0.6
CCE 5x5x5
CCE 5x50x5
0.4
0.2
0
0.5
1
1.5
2
2.5
Center frequency (GHz)
3
Figure 2.6. A small cubical corner-fed CCE and a ”full-width” corner-fed CCE on a typical
smartphone chassis with bandwidth potentials. Dimensions in mm.
The center-fed CCE has slightly better bandwidth potential at low frequencies, below the first chassis resonance. The offset-fed CCE, on the
other hand, has notably better bandwidth potential at frequencies above
2.5 GHz, obviously thanks to the low-Q resonance of the CCE at 2.7 GHz.
These differences would have been very difficult to detect by looking at
the impedance trace only.
The L-section matched case can be used to predict differences in the
bandwidth performance of the slightly different antennas matched with
more complex circuits. Therefore we can conclude that the center feeding
strip should be used if the antenna is designed to operate only at frequencies below 1 GHz. If frequencies above 2.3 GHz are desired, the offset feed
should be preferred. In Chapter 3 we show how the information given by
this example is utilized in having the two feeding strip locations in the
same antenna [II, III].
In addition to the feeding configuration and the chassis size, also the
size of the CCE has a major influence on the antenna performance. For
example, Figure 2.6 presents a comparison of the bandwidth potentials
35
Characterization of mobile terminal antennas at an early design stage
0.4
Relative bandwidth
0
|S11| (dB)
-5
-10
-15
-20
-25
-30
0.5
1
2
1.5
Frequency (GHz)
2.5
3
0.3
0.2
0.1
0
0.5
1
1.5
2
2.5
Frequency (GHz)
3
Figure 2.7. A generic dual-band PIFA antenna with its simulated impedance matching
result and bandwidth potential at 0.5 – 3 GHz.
of a small cubical CCE and a wide CCE, both fed with a feeding strip
at the corner of the chassis. The bandwidth potential is presented using
the same bandwidth scale as in Figure 2.5. The bandwidth potentials of
the smaller and larger elements at 900 MHz are 10 % and 37 %, respectively. With dual resonant matching, the small element can reach enough
bandwidth for GSM850/900 systems. At high frequencies above 2 GHz
the small element can operate with very wide bandwidths. Similar small
elements have been utilized succesfully e.g. in [57]. However, the small
element is not sufficient if operation at e.g. 700 MHz is desired in addition
to GSM850/900. In [58], the problem of minimum CCE size with respect
to desired bandwidth at the frequencies 700-960 MHz was addressed with
a parametric study. The main conclusion was that sufficient bandwidth
with small CCE and compact chassis could only be achieved by compromising antenna performance.
Yet another example is shown to clarify the limitations of frequency
tunability caused by high-Q resonances of some antenna structures. Figure 2.7 shows a generic dual-band PIFA structure (adapted from [59])
designed to operate at GSM850 and GSM1900 frequencies. The PIFA resonances are quite narrow, and the bandwidth is modest. The bandwidth
potential of the antenna (Figure 2.7) indicates that some additional bandwidth is available around the natural resonance frequencies with the help
of the L-section matching, but for example around 1.4 GHz practically no
bandwidth is available. This PIFA would thus not be very suitable for
a tunable antenna having a broad continuous tuning range between the
antenna resonances, unless the tuning can be realized by dynamically
modifying the antenna geometry during operation. This option is briefly
discussed in Section 4.1.
36
3. Multi-band non-resonant mobile
terminal antennas
Today’s multi-system mobile radios require that the handset antennas
operate at multiple frequency bands extending over a wide range of frequencies, as specified by 3GPP [60,61]. The inherently narrow bandwidth
of a single-resonant mobile terminal antenna cannot fulfil these requirements. Therefore it is necessary to reach either multi-resonant operation
or frequency reconfigurability.
Most of the bandwidth enhancement methods for mobile handset antennas have first been used with self-resonant antennas. To situate the
multi-band non-resonant antennas in the right historical context, it is necessary to briefly discuss also the multi-band self-resonant antennas in this
chapter. Moreover, many of the bandwidth enhancement methods have
been developed originally for microstrip patch antennas, slot antennas,
monopoles, or other ancestors of the modern mobile handset antennas.
The main focus here is, however, on the inherently non-resonant mobile
handset antennas, especially the CCE antennas.
Bandwidth enhancement is also closely related to the miniaturization
of mobile phone antennas. Shorting posts, meander lines etc. have been
widely used to shrink the antenna to an acceptable size. Thereby, it should
be remembered that the methods used to realize multi-band operation are
often targeted primarily to reach an acceptable level of performance with
a minimum size structure, and not to maximize bandwidth per se.
Section 3.1 gives an overview of geometry based bandwidth enhancement methods for mobile handset antennas. These methods are typically
used with self-resonant antennas, but sometimes also with non-resonant
antennas. Impedance matching has been widely used as an antenna
bandwidth enhancement method for several decades [19, 51]. In Section
3.2, impedance matching is discussed in the context of matching of nonresonant antennas simultaneously at noncontiguous frequency bands.
37
Multi-band non-resonant mobile terminal antennas
3.1
Bandwidth enhancement based on antenna geometry
Given the total dimensions of the antenna element and chassis combination, there are several alternative methods to improve the bandwidth of
a generic mobile terminal antenna with geometry changes. Optimizing
the geometry of the antenna element is one option. Another option is to
optimize the geometry of the mobile terminal chassis. Also parasitic resonators can be added to improve the bandwidth.
3.1.1
Self-resonant antenna elements
The multi-resonant behavior of mobile terminal antennas is conventionally achieved by optimizing the antenna geometry so that multiple resonances are excited in the radiating structure, realizing impedance matching and good radiation properties at all desired frequency bands. This
can be done with e.g. slots or meandering in the radiating element, or
with parasitic resonating elements coupled to the main (driving) element.
The resulting multi-band self-resonant antenna may include very complex geometrical and mechanical details, which can make the antenna
fragile, or sensitive to proximity effects. On the other hand, the geometrical complexity can ensure additional frequency bands without increasing
the antenna size.
The publishing activities in this field have been extensive since the
1990’s, and new designs are still published actively. Design methods
and example structures are discussed quite extensively in the literature,
e.g. [1–5, 62, 63], so they are not covered in this thesis.
3.1.2
Modified radiating ground plane
The bandwidth of a mobile terminal antenna can be significantly improved by using a segmented ground plane instead of a uniform ground
plane [8,64–70]. Since the chassis of the mobile handset is in any case the
main radiating element at the lower end of the mobile communication frequencies [8], it is obvious that optimizing the chassis shape enables even
more efficient use of the chassis as a radiating element than modifying
just the ”antenna element”.
In fact, a separate antenna element is not necessary at all if the chassis resonances are excited with a suitable feeding configuration over a
slot in the chassis [66–68]. A slotted ground plane can also be used
38
Multi-band non-resonant mobile terminal antennas
to enhance the inherent frequency band of a conventional antenna element [8, 64, 65, 71–77]. The slot in the ground plane decreases the lowest chassis resonance frequency and the quality factor at low frequencies,
thus improving the achievable bandwidth at those frequencies [66, 78].
On the other hand, slots placed under an antenna element near the end
of the chassis can operate as parasitic resonators at the higher frequencies, improving the bandwidth at those frequencies as well [73, 75, 76].
In [68], a slotted chassis is used as the antenna for lower frequencies
(850/900 MHz) while a CCE antenna on one end of the chassis is used
at higher frequencies (1800/1900/2100 MHz). The slot divides the chassis
into two parts, and the CCE uses only one of the two parts as its radiating
ground plane. The location of the slot in the optimized design in [68] was
chosen so that sufficient bandwidth at the low frequencies was obtained
while the part of the ground plane used by the CCE had optimal length,
i.e. half-wave resonance at the higher frequencies, enabling optimal utilization of the chassis wavemodes.
Realizing slots etc. in the middle of the chassis of a realistic mobile
handset is not straightforward in practice. Many conductive parts of
the mobile terminal, which cannot be segmented, operate as parts of the
ground plane. For example, the background of the display of mobile terminals is usually metallic. If this metallic surface is placed over a slot in
the ground plane, the radiation of the slotted ground plane is disturbed,
which results in significantly decreased bandwidth [66, 79]. Countermeasures against this decreased bandwidth can be applied and some of the
lost bandwidth can possibly be restored [79]. The design of the whole mobile terminal structure can also be possibly made so that no conductive
parts coincide with the ground plane slots [66, 80]. With today’s large
handset displays this is very difficult, though.
With the clamshell (folder) type mobile terminal, it is possible to realize
an RF-wise undisturbed slot in the middle of the chassis [81]. However,
this approach does not work if the clamshell is closed. In [81], the chassis
radiation of the clamshell type terminal is exploited in digital television
(DTV) reception, for which the approach is feasible. In [69, 82], two longitudinal slots are made on the chassis in order make the chassis a symmetrical folded dipole. The resulting antenna has quite wideband operation,
but again the realization of the slots on the chassis might be problematic
in practice. In addition to the difficulty of segmentation, also radiation
safety issues have limited the use of segmented ground planes in mobile
39
Multi-band non-resonant mobile terminal antennas
terminals [83]. High Specific Absorption Rate (SAR) values, exceeding the
radiation safety limits, have been reported [68, 75, 77, 83].
Removing the ground plane under the antenna element is currently a
popular and efficient method to improve the achievable bandwidth of mobile terminal antennas, e.g. [37, 57, 84–90], [II, III, VIII]. By using the
so called off-ground antenna elements, the bandwidths can be made significantly larger than with on-ground antenna elements [5]. A possible
drawback is the tendency for increased SAR [91], [VIII], like in the case
of the segmented ground plane. However, in this thesis also off-ground
coupling elements are used since the benefit from the bandwidth point
of view exceeds the possible disadvantages, as the SAR limits are always
finally fulfilled in the designs presented in this thesis.
Further methods of improving the bandwidth of mobile terminal antennas with ground plane modifications include wavetraps [92, 93] and other
metallic strips connected to the ground plane [94, 95]. Quarter wavelength wavetraps located at the long edges of the ground plane can be
used to make the ground plane seem shorter in electrical length than it
is around the resonance frequency of the wavetraps [92]. Since the wavetraps make the chassis effectively shorter, they cannot improve the antenna operation at frequencies lower than the lowest chassis resonance.
By contrast, metallic strips connected to the end of the ground plane opposite the antenna element can be used to make the chassis seem longer,
which can improve the bandwidth at frequencies below the lowest chassis
resonance [94].
3.1.3
Location of CCE and feeding configuration
The characteristic wavemode theory can be used to find the optimal location for the antenna element from the bandwidth point of view [43], or in
the case of multi-antenna systems, to optimize the interoperability of the
antenna elements [96]. Also the location of the antenna feed can be optimized for the best possible performance by using the information of the
characteristic modes [72]. The shape of the feeding probe/strip/plate can
be modified to optimize the bandwidth as well [97–100]. The capacitive
feeding structure (e.g. [87,89,97,101]) has proven to be especially efficient
in the bandwidth enhancement.
In terms of bandwidth potential, the optimal location of a small probetype capacitive coupling element on a chassis is at a corner of the chassis
[10, 43]. However, the small probe does not provide as good bandwidth
40
Multi-band non-resonant mobile terminal antennas
potential as a larger CCE (Figure 2.6). To obtain adequate instantaneous
bandwidth for today’s multi-standard handsets, the CCE should be made
as wide as the chassis. In this case, the optimal location is at the narrow
edge of the chassis [44]. The optimal feeding configuration of the CCE
depends on the desired frequency bands, and can be optimized with EM
and circuit simulations.
The significance of the shape and location of the feeding strip was illustrated in the example presented in Figures 2.4 and 2.5 in Section 2.3 with
the simple off-ground type CCE and two different feeding configurations.
The two different feeding strip locations cause the inherent impedance
and also the bandwidth potential of the antennas to be different at some
frequencies. Not only the location of the offset feeding strip but also the
trapezoidal shape helps the CCE antenna to improve the bandwidth potential at the higher frequencies. As the strip is wider towards the CCE
end, the effective series inductance of the strip is low compared to the
narrow rectangular strip. Also, since the strip is narrow at the chassis
end, the effective parallel capacitance of the strip is quite low. This kind
of arrangement helps to realize impedance matching of the antenna at
higher frequencies because high series inductance and parallel capacitance would result in lowpass frequency response. These observations on
the CCE feeding configurations become more meaningful when a multiband matching circuit design for the CCE of Figure 2.4 is presented in
Section 3.2.
3.2
Multi-band matching of non-resonant antennas
Inherently non-resonant radiating structures such as CCE antennas, short
dipoles and short monopoles or small loops always need additional matching to operate well as antennas, but impedance matching can be applied
to self-resonant antennas as well: Matching resonators can be used to extend the inherent frequency band of the already resonant antenna [50,51].
The matching resonators can be thereby used as an alternative to modifying the antenna geometry [48].
3.2.1
Background of broadband matching
Broadband impedance matching originates from the circuit theory [51,
102–104], and antennas are only one of its many application areas. The
41
Multi-band non-resonant mobile terminal antennas
theory of broadband matching has been widely used e.g. in design of
microwave amplifiers [105–108], where a lot of its advances come from.
Also antennas have benefited from the broadband matching theory. One
popular method, or group of methods are the Real Frequency Techniques
[103, 106, 109], covered quite extensively from the antenna point of view
in [109].
Typically for the broadband matching theory, the load impedance is
modeled either as a constant resistance [108], in which case the broadband matching is equivalent to filter design, or as a simple resonator type
circuit [103, 110]. This suits well short monopole and short dipole type of
antennas, e.g. [111]. With resonant PIFA antennas, multi-band matching
circuits enhancing the inherently narrow bandwidths have been successfully applied at noncontiguous frequency bands [48, 112, 113].
In the case of CCE antennas, the broadband equivalent circuit is much
more complex than a simple resonator (Section 2.2, [I]). Thereby, the classical theory of broadband matching is not easily applicable to CCE antennas over the wide frequency ranges needed for today’s mobile communications. For single CCE type antennas, multi-resonant matching only
at adjacent frequency bands has been previously demonstrated [114,115].
The multi-band CCE antennas operating at noncontiguous bands have
been based on implementing separate coupling elements for the different frequency bands [57, 85, 116]. However, in the next subsection of this
thesis, novel single-element CCE antennas capable of operation over two
noncontiguous, broad frequency bands will be presented.
3.2.2
Dual-branch matching circuits for a CCE antenna
Although it is easier to implement separate matching circuits for separate
antenna elements, it does not mean that the multi-band matching of a
single CCE is impossible or impractical. In this thesis work [II, III, VIII],
matching circuits for CCE antennas are developed in order to achieve simultaneous multi-band operation at the current mobile communications
frequencies, with performance comparable to (or better than) that of selfresonant antennas of similar size. The idea is based on frequency multiplexing, in this case diplexing. Two matching circuit branches are designed to operate at different frequencies, one with a low-pass frequency
response and the other one with a high-pass response [II,III].
In Figure 3.1a, a dual-branch multiband matching circuit designed for
the CCE in Figure 2.4a is shown [VIII]. The branches are isolated so that
42
Multi-band non-resonant mobile terminal antennas
CCE
L h4
CCE
Ch4
Ch2
L h2
16
24
Ch3
Feed C
h5
Ll4
Feed
Ch1
2
Ll2
Ch3
Ll2
Ll4
6
2
Ch13
24
Cl3
50
24
Ll1
Cl3
Ll1
10
5
(a)
(b)
CCE
Ch4
Lh5
L l4
C h3
Ch13
L l2
Cl3
L l1
Feed 2
Feed 1
L h2
16
6
2
50
24
10
(c)
Figure 3.1. Multiband matching circuits for a CCE antenna with three different feeding configurations. (a): Dual-branch matching circuit with centered feeding
strip, having single antenna feed, from [VIII]. (b): Dual-branch matching circuit with a centered feeding strip and an offset feeding strip, having single
antenna feed, from [II]. (c): Dual-branch matching circuit with a centered
feeding strip and an offset feeding strip, having independent antenna feeds
for the branches, from [III].
they do not significantly affect the behavior of each other, because the
electromagnetic signal at the operating frequency of one branch sees the
other branch as a very high impedance or as a suitable pure reactance,
preferably as an open circuit [II, VIII]. Consequently, the antenna supports simultaneous multiband operation at the lower (700-960 MHz) and
higher (1710-2170 MHz) mobile communication frequencies without need
for frequency tuning.
However, the matching circuit design of Figure 3.1a is not optimal in
terms of achievable bandwidth at the higher frequencies. As was seen
in Figure 2.5, the bandwidth potential of the CCE is better at the high
frequencies when an offset located feeding strip is used. Consequently
in [II, III], [117] the dual-branch matching approach was combined with
optimization of the feed locations for the CCE. The centered feeding strip
and the offset feeding strip were combined to one CCE antenna design
(Figure 3.2). Figure 3.1b shows a schematic figure of the CCE and matching circuit design. The two matching branches operate as a diplexer, and
43
Multi-band non-resonant mobile terminal antennas
5
5
50
10
100
Figure 3.2. Geometry of the CCE and chassis design of [II] with two feeding strips to the
CCE. Dimensions are in mm.
|S11| (dB)
0
-6
-12
CCE, single feeding strip
CCE, two feeding strips
0.5
1
2
1.5
Frequency (GHz)
2.5
3
Figure 3.3. Measured impedance matching level of the multi-band CCE designs shown
in Figure 3.1a and b, [II, VIII].
the two different feeding strip locations on the CCE are selected to provide optimal bandwidth potential both at low and high frequencies of the
mobile 3G and LTE-A systems.
The feeding strip of the higher frequencies is placed similarly to the
one in Figure 2.4b, so that a self-resonance of the CCE-chassis combination is excited at about 2700 MHz. This way, one additional resonance
is obtained compared to the design in [VIII]. Since the resonance has a
low quality factor, it does not harm the antenna operation at other frequencies. Figure 3.3 shows the impedance matching comparison of the
prototype designs in [II] and [VIII] (antenna B in [VIII]) using the otherwise identical coupling elements. It is evident from the result that
the approach using two feeding strips is superior from the bandwidth
point of view. The operation frequencies of this antenna are 698-960 MHz
and 1710-2690 MHz, comprising the conventional 2G and 3G mobile system frequencies (824-960 MHz and 1710-2170 MHz), and additionally the
LTE-A system frequencies 698-798 MHz and 2500-2690 MHz, with also
the 2.45 GHz ISM band used by Bluetooth and WLAN systems. The total efficiency of the antennas is at an acceptable level for mobile terminal
antennas, being better than 50% at their respective operation frequencies
[II, VIII].
In [II] and [VIII], the matching circuit designs rely on the conventional
single antenna feed architecture of the RF front-end, but the dual-branch
44
Multi-band non-resonant mobile terminal antennas
0
|S11| (dB)
|S11| SF
|S11| DF
-6
|S22| DF
-12
-18
0.7 0.96
1.71 2.17 2.69
Frequency (GHz)
3.5
0
100
-1
79
-2
63
-3
50
SF htot
DF 1 htot
DF 2 htot
-4
-5
-6
0.7 0.96
1.71 2.17 2.69
Frequency (GHz)
40
Efficiency (%)
Efficiency (dB)
Figure 3.4. Impedance matching comparison of the single-feed (SF) multi-band CCE antenna of Figure 3.1b, [II] and the dual-feed (DF) multi-band CCE antenna of
Figure 3.1c, [III].
32
3.5
25
Figure 3.5. Total efficiency (ηtot ) comparison of the single feed multi-band CCE antenna
of [II] and the dual-feed multi-band CCE antenna of [III]. SF = single feed,
DF = dual feed. With the dual feed structure, port number 1 refers to the
lower frequency feed, and port number 2 refers to the higher frequency feed.
matching circuit design works at least as fine as with a dual-feed architecture. The circuit of [II] was modified in [III] to be compatible with an RF
front-end interface having a dual antenna feed (Figure 3.1c). With minor
circuit modifications, and without any modifications to the antenna element, the good operation of the antenna at all of its frequency bands was
maintained also in the case of the dual antenna feed. Figure 3.4 shows
the input impedance comparison of the dual-branch matching circuit antennas having the two feeding strips and one or two antenna feeds. In
Figure 3.5, the corresponding comparison of total efficiencies is shown.
It can be seen that while the dual-feed structure achieves slightly better
impedance matching levels, the total efficiencies of the antennas at the
LTE-A frequency bands are very similar.
The performance of the antenna prototypes in [II] and [III] is comparable to or even better than the performance of state-of-the-art mobile terminal antennas of similar size and otherwise comparable structure published recently in the scientific forum [37, 57, 86, 118]. The antennas also
do well in comparison with the top products of some commercial mobile
antenna manufacturers [119, 120]. A comparison of some key character-
45
Multi-band non-resonant mobile terminal antennas
Table 3.1. Comparison of multi-band CCE antenna designs presented in this thesis, antennas published in recent issues of IEEE Transactions on Antennas and Propagation, and antennas provided by commercial antenna manufacturers.
SF = single-feed, DF = dual-feed, ηtot = total efficiency.
Ant. size = dimensions of antenna element including ground clearance
Tot. area = area of handset including antenna element and chassis
BWlow = bandwidth at lower frequency band
BWhigh = bandwidth at higher frequency band
Ant. size
Tot. area
BWlow
BWhigh
ηtot
mm3
mm2
MHz
MHz
%
SF, [II]
10×50×5
110×50
698 – 960
1710 – 2690
55–86
DF, [III]
10×50×5
110×50
698 – 960
1710 – 2690
54–91
Tunable, [III]
10×50×5
110×50
750 – . . .
. . . – 2500
45–69
[37]
10×55×8
115×55
704 – 960
1710 – 2690
63–91
[57]
5×40×5∗
105×40
824 – 960
1710 – 2170
≈30–70
[86]
13×50×5
120×50
698 – 960
1710 – 2170
≈43–80
[118]
5×60×5
105×60
824 – 960
1575 – 2500
≈47–73
[119]
18×50×7
102×50
698 – 960
1710 – 2170
≈35–71
Ref.
2500 – 2690
[120]
∗
13×45×5
107×45
698 – 960
1710 – 2170
≈40–83
) Excluding a small (unknown) gap between the ground plane and the antenna elements.
Including 30×5 mm2 empty space between the two antenna elements, since conductive
materials in this empty space would probably deteriorate the antenna operation.
istics is presented in Table 3.1. In particular, the novel designs of this
thesis achieve better bandwidth and efficiency at the lower frequencies
than the comparable CCE designs having separate elements for lower
and higher frequencies [57, 116]. The single antenna element uses the
total volume allocated for all antennas more efficiently than two separate
elements occupying the same total volume. The whole antenna volume
is conveniently available at the low frequencies where the fundamental
limits of small antennas restrict the performance. In the designs using
separate antenna volumes for different frequencies the antenna volume
is only partly in use at any frequency. Regarding operation at the low frequencies, this would lead to suboptimal efficiency and impedance matching performance.
46
4. Frequency tunable CCE antennas
4.1
Frequency tuning of handset antennas
The relative frequency bandwidth of a communication channel that a mobile handset uses instantaneously is typically quite narrow. For example,
in the 3GPP specification for the LTE communication systems [60], the
maximum bandwidth of a single channel is 20 MHz, or for instance, 2.4%
relative to 830 MHz. In comparison, the combined relative bandwidth of
the LTE frequency bands below 1 GHz is 31.6 % relative to the center
frequency 830 MHz, as the UTRA FDD bands V, VI, VIII and XII–XIV
together comprise 699 – 960 MHz [60]. Since a mobile handheld radio
does not need to cover all of its operation frequencies simultaneously, neither needs the antenna. If it is sufficient to use just a part of the total
bandwidth at a time, frequency tuning can be applied.
From the fundamental limitations of small antennas (Section 2.1), one
can derive the simple relationship: narrowing the instantaneous bandwidth allows decreased antenna size. Therefore, frequency tuning is also
an enabler of antenna size reduction in mobile terminals. It allows, at the
cost of added design complexity, to cover effectively a wide range of frequencies with a compact antenna. On the other hand, the narrow instantaneous bandwidth sets strict requirements for the tuning circuit with
respect to switching speed, sensitivity to impedance variations, and also
efficiency and linearity [121–123], [VI, IV].
4.1.1
Enabling technologies
Electrical tuning of antennas requires the use of components which can
have at least two different states in terms of electrical properties. Semiconductor devices are one large group of components capable of such be-
47
Frequency tunable CCE antennas
havior. Diode switches (pin diodes) [118, 121, 124–130], continuously tunable varactor diodes [124, 131–139], or almost continuously, stepwise tunable reactances (e.g. CMOS), [140], [III] and transistor based switches
(FET, HEMT, CMOS) [7, 123, 124, 128, 137, 141–146], [IV] are common
semiconductor based devices used in tuning circuits for antennas.
Ferroelectrics such as barium strontium titanate (BST) varactors have
been used to tune antennas as well [147–149]. Ferrites and liquid crystals can be used to implement antenna substrates with tunable electrical
properties [129]. Microelectromechanical devices (MEMS) can be used either as RF switches or as tunable capacitors [128, 150–157].
Each technology has its advantages and disadvantages. For example,
pin diodes have good linearity and power handling capability, and they
can also have low RF power loss. On the other hand, the biasing circuit
of the diodes can be rather complex, and the control current consumption
is relatively large [158], which is not a good property for a mobile, battery
driven device. The linearity and power handling of conventional varactor diodes is generally not as good as those properties of the pin diodes,
but the linearity properties of varactors have improved over the past few
years [159]. Another advantage of varactors is that they enable a continuously tunable circuit.
The FET, HEMT and CMOS switches have better switching speed than
the pin diode and they can also have very low bias current consumption.
On the other hand, the insertion loss is slightly higher than with the diode
switches. Today’s transistor-based switches can have quite good linearity and power handling properties. Among the semiconductor switches,
the CMOS switch has the best integration properties, and thanks to the
mass production compatibility, also the lowest unit cost. Stepwise tunable
CMOS capacitors [140] can provide almost continuous tuning with better
linearity than the conventional varactors.
RF MEMS have been a promising technology for antenna tuning for a
couple of decades already [150]. However, they have not gained much popularity in mobile user equipment. MEMS switches and capacitors have
very low insertion loss compared to the semiconductor alternatives, and
also the DC control current consumption is almost zero. RF MEMS can
be electrostatically, piezoelectrically or magnetically actuated. The piezoelectric and magnetic MEMS can have low actuation voltages of a few
Volts [160], [V], like the semiconductor switches. The electrostatically
actuated MEMS need tens of Volts of bias voltage. A typical battery of
48
Frequency tunable CCE antennas
a mobile device only provides less than 5 Volts, so an additional charge
pump would be needed to actuate the electrostatic MEMS. The main disadvantages of the MEMS are related to reliability and packaging challenges [151], which cause MEMS devices to be quite expensive and thus
not a very attractive option for tuning circuits of mobile handsets. MEMS
capacitors at low frequencies also suffer from low tuning ranges [161].
Optimistic views on the RF MEMS exist, too [162]. The integration of RF
MEMS in a CMOS process has been already demonstrated [163].
BST technology can provide good power handling capability [164] and
easy integration [147, 164]. However, the tuning range of the antennas
tuned with BST varactors is fairly narrow [165].
The transistor based tuning components can have both passive and active behavior. An active tuning circuit can have gain, and even non-Foster
reactances (negative slope reactances) can be realized [166–169]. The
main drawbacks of the active and especially the non-Foster tuning circuits include high power consumption, high noise, often poor linearity,
and possible stability issues [170].
4.1.2
Methods for tuning mobile terminal antennas
In the context of antennas, tuning is usually understood as adjusting
the resonance frequency/frequencies of the antenna. This point of view
has been used in microstrip patch antenna design [129, 132, 133], and
consequently in most of the tunable mobile terminal antenna designs as
well. The resonance frequencies have been electrically tuned by changing
the reactive loading of the antenna element [121, 128, 132, 135, 136, 155,
160, 167], or by adjusting the input impedance of the antenna at the RF
feed [123,126,130,138–141,143,154,157,171,172]. These methods, in the
case of self-resonant antennas, often result in quite modest tuning ranges,
up to a few tens of per cent around the inherent resonance frequencies of
the antenna.
Frequency tuning of mobile terminal antennas can also be realized by
dynamically connecting or disconnecting parts of the radiating structure
with reconfigurable elements [118, 127, 134, 152, 156, 173, 174]. Switches
in an antenna slot can be used to change the current paths on the antenna elements to be shorter or longer in order to change the resonance
frequency [127]. Parasitic radiators can be connected to the main antenna element with switches [174, 175]. The width, length or location
of shorting posts or feeding pins of PIFA antennas can also be changed
49
Frequency tunable CCE antennas
by using switches in order to tune the resonance frequencies [176, 177].
These methods can be, of course, also categorized as antenna geometry
based bandwidth enhancement techniques like the ones discussed in Section 3.1.
Much broader tuning ranges can be achieved by modifying the antenna
geometry than by adjusting the impedance of the self-resonant antenna
at the antenna feed. In [156], a tunable ”hybrid” loop/monopole antenna
for a mobile handset was presented. The antenna uses MEMS switches
for selecting one of three operating modes at a time, resulting in a continuous tuning range from 660 MHz to 6 GHz. Unfortunately, no measured
efficiency results of this antenna were presented.
In [129], different techniques for implementing frequency reconfigurability are reviewed with many examples from the literature, mainly regarding patch antennas. Electronic switches and varactors are currently
considered the best suited tuning technologies for the frequency agile antennas [129]. Theoretically, the resonance frequency of mobile terminal
antennas can also be tuned by filling the antenna volume with a material
having tunable permittivity or permeability. Microstrip antennas filled
with magnetized ferrites have been designed [178], but the method has
not been popular among the handset antenna designers. The magnetic
biasing of microwave ferrites requires the use of either permanent magnets or electromagnets, which are relatively large in size. Also the losses
of the ferrites are potentially high. Ferroelectrics or liquid crystals are
not feasible for this purpose at present, either [129].
Tuning has also another interpretation in addition to tuning the frequency. In the case of inherently non-resonant antennas, the natural
resonances do not necessarily exist, and the tuning circuit can include
impedance matching networks which are used essentially to create resonances at almost arbitrary frequencies. In this case, ”impedance tuning”
could be more appropriate term instead of frequency tuning. The tuning of
non-resonant antennas has been demonstrated e.g. in [85, 137, 179, 180].
In this thesis work, tuning is approached from this impedance tuning
point of view. In [IV, V], the input impedance of a CCE antenna is adjusted
to resonance at selected frequencies with a switchable matching circuit,
without a natural resonance being present near the operation frequencies. In [III], a tuning circuit also utilizing the frequency tuning approach
is designed, implementing continuous tuning of multiple resonance frequencies and realized with a tunable capacitor.
50
Frequency tunable CCE antennas
The impedance tuning approach is also natural for adaptive impedance
matching used to match the antenna to the impedance of the receiver (RX)
or the transmitter (TX) [142, 162, 181–186]. This can be done with an
impedance synthesizer, an impedance or power detection method, and a
control unit [162,182–184,187]. An automatic antenna tuning unit (ATU)
can be useful in readjusting the input impedance of an antenna when, for
example, the presence of the user of a mobile terminal causes impedance
detuning [142,162]. The ATU can also be used to simplify the design of the
RF front-end of multi-system radios [187]: A tuning circuit based on an
impedance synthesizer with a feedback system allows a simple method to
match the antenna impedance to the impedance of the RF front-end. With
the limited resources available in the thesis work, the implementation of
an ATU was left out of the scope of this thesis. Such a circuit would be
very costly to realize without the possession of a suitable process line.
4.2
4.2.1
Broadband tuning of CCE antennas
Tunability
As discussed in Section 2.2, a non-resonant CCE can be non-selective at
the frequencies of mobile communications. Since the impedance trace
of the unmatched CCE does not go very close to the edge of the Smith
chart at most frequencies, the CCE can be matched virtually at any frequency above some lower frequency limit. This is a major difference between the non-resonant antennas and most self-resonant antennas. A
self-resonant antenna may have multiple resonance frequencies, but often
the antenna is very selective between the well-matched frequency bands,
the impedance trace being very close to the edge of the Smith chart, like
in the PIFA example presented in Section 2.3 (Figure 2.7). Impedance
matching at these intermediate frequencies is very challenging.
The tuning circuit of a non-resonant antenna can consist of several independent matching circuits which are isolated from each other with an
isolation mechanism, typically RF switches. Figure 4.1 shows a general
tuning topology for CCE antennas. The circuit is entirely located between
the RF feed and the antenna element. If the isolation mechanism works
well, the separate matching circuits do not affect the operation of each
other, and thus the tuning range of the non-resonant antenna is only lim-
51
Frequency tunable CCE antennas
Separate matching circuits 1...m
1
RF in
RF to antenna
2
SPMT
Switch
m
SPMT
Switch
Figure 4.1. Block diagram of the general tuning topology for CCE antennas using single
RF feed and one CCE. Each of the blocks 1...m represents a matching circuit
for a single frequency band.
ited by the efficiency and realizability of the tuning components.
In [137], a tunable CCE antenna for a digital TV system (DVB-H) was
designed. The tuning was based on a varactor having a large capacitance
tuning ratio. Measured frequency tuning range of 1.7:1 was achieved with
Lretn ≥ 7 dB over 442–758 MHz. The radiation efficiency at the DVB-H
frequencies ranged from 25% at 470 MHz to 65% at 710 MHz.
In [179, 180], numerical considerations on the realization of a compact
tunable CCE antenna with operation over 0.1 – 2.5 GHz were presented.
The tuning circuit could be implemented in a very compact way by using LTCC (Low temperature co-fired ceramic) technology to integrate the
matching circuits for each communication system. Additionally, a switch
would be needed to select the system used at each moment. In [85, 188],
simulated results were presented to show that a compact CCE antenna
with two coupling elements on the same chassis would have at least 442
MHz to 2896 MHz tuning range with Lretn ≥ 6 dB. However, a real tunable
antenna was not implemented.
Future communication systems based on the concepts of software-defined
radio (SDR) and cognitive radio may require that the mobile device is operational at different system bandwidths extending even over a decade
frequency range [189, 190]. The non-resonant antenna which lacks inherent selectivity is thus to be reckoned with as a promising antenna choice
for SDR and cognitive radios.
4.2.2
CCE antennas with wide tuning range
In [7, 146], the idea of broadband frequency tuning of CCE antennas was
tested in practice. The objective was to switch the operation band of a compact CCE antenna (chassis dimensions 40×100 mm2 ) between the cellular
900 MHz and 1800 MHz bands, corresponding to one octave tuning range.
The general tuning topology was taken as a starting point. A two-state
tuning circuit using HEMT switches and chip reactances was designed
52
Frequency tunable CCE antennas
L l1
switch 1
switch 1
L h4
C h3
(a)
L h1
L h2
Ci L
1
ZL
Ch4 L
h3
Ch2
Ch1 switch 2
L l4
L l1
Ll2
Cl3
L1
ZL
(b)
Figure 4.2. (a) The tuning circuit topology used in [IV] with a HEMT switch as the reconfigurable element. (b) The tuning circuit topology from [V] used with two
magnetic latching MEMS switches. The CCE with the chassis is represented
as the load impedance ZL .
and a prototype was built. The results show that the CCE can be tuned
over the desired broad frequency range with a few basic components.
In this work [IV], a revision of the circuit in [7, 146] was made in order
to control the losses and nonlinearity of the RF switches. The number of
switches was reduced to one (see Figure 4.2a) by omitting the switch on
the antenna element’s side, as this switch had the highest losses in the
circuit [146]. This decreased the overall losses significantly. The total efficiency of the antenna design of [IV] at the GSM900/1800 frequency bands
was 47 – 66%. Also distortion of the circuit was dramatically decreased
thanks to the more suitable impedance environment of the remaining
switch. One characteristic of CCE antennas is that at low frequencies the
radiation resistance of the CCE is low and the input impedance is highly
capacitive. A switch in this kind of impedance environment is vulnerable
to increased insertion loss and also to voltage distortion [IV, VI].
However, the circuit in [IV] was not anymore of the general tuning circuit type (Figure 4.1) and cannot be directly applied in a circuit combining switches with more than two tuning states. In [V], another revision
of the circuit was presented, this time as a quad-band antenna design for
GSM850/900 and GSM1800/1900 systems, with measured Lretn ≥ 6 dB
and 40% ≤ ηtot ≤ 63% at the operation frequencies. The circuit was now
of the general tuning circuit type and included two magnetically actuated
latching MEMS switches (Figure 4.2b). This MEMS type is more lossy
than the electrostatic MEMS, but the lower actuation voltage allows it to
be more easily adaptable in handset antenna tuning circuits. More importantly, the MEMS switches outperform the semiconductor circuits in both
efficiency and linearity, and should thus be preferred if possible [V].
In [III], a broadband continuously tunable CCE antenna was designed
based on the versatile CCE platform shown in Figure 2.4a. The tuning circuit has one tunable component, a digitally tunable CMOS based capaci-
53
Frequency tunable CCE antennas
0
CCE
|S11| (dB)
3.3 nH 3.3 pF
-6
3.3 pF
C2
-12
-18
-24
4.7 nH
3.9 nH
C 2 = 5.6 pF + DTC in series
0.5
|S 11 |
min (|S11 |)
1
1.5
2
Frequency (GHz)
2.5
3
Figure 4.3. Schematic picture of the tunable matching circuit from [III] with the measured |S11 | of the prototype antenna. The solid lines represent the matching
with the 32 different states of the digitally tunable capacitor and the dashed
line shows the overall achievable |S11 |.
tor (DTC). The capacitance of the DTC is tunable in 32 steps from 1.12 pF
to 5.18 pF. The whole tuning circuit is shown in Figure 4.3 together with
the measured impedance matching result of the antenna. The capacitor
and the fixed-value shunt inductor form a resonator that allows maximizing the tuning range. The antenna achieves Lretn ≥ 6 dB and fairly good
45% ≤ ηtot ≤ 69% at the frequency range 750–2500 MHz.
Overall, the tunable CCE antennas in this thesis represent a practical
proof of concept with regard to broadband frequency tuning of CCE antennas. Although the total efficiencies obtained with the antenna prototypes
are quite moderate, it is probable that in the near future the development of the tuning technologies will produce tunable components with
improved performance with respect to insertion loss [162].
4.3
Challenges set by different impedance environments
Real tuning circuits consist of real components with nonidealities. Fair
comparison between different tuning circuit designs is often difficult because sometimes only idealized simulation results using lossless switches
are presented, or efficiency results are omitted. In order to address the
practical challenges often ignored in tuning circuit design, realistic power
losses of some typical tuning elements with arbitrary load impedances
were studied in [VI]. Specific attention was paid to the role of possible
variations in the input impedance of the antenna due to interactions with
the environment. Linearity of a FET switch in different impedance environments was studied [IV], with a proposed case-specific solution to the
non-linearity problem.
54
Frequency tunable CCE antennas
0
dB
0
dB
2
2
4
4
6
6
>8
>8
(a)
(b)
Figure 4.4. (a) Simulated power loss of a semiconductor switch as a function of load
impedance in series with the switch. (b) Power loss of a digitally tunable
CMOS capacitor (binary state 8 of 32) as a function of load impedance in
parallel with the capacitor at 2 GHz. [VI]
4.3.1
Dissipative losses
Variations in the impedance environment of a tuning component can affect the power loss of the component significantly, as shown in Figure
4.4. Since there can be large variations in the antenna impedance due
to realistic environments of use, such as the user proximity, it is essential to ensure that the power dissipation in the circuit does not cancel
the advantages of the impedance matching [VI]. It is important to identify the challenging impedance environments to which the components
are placed. Equally important is to identify the circuit components which
are the most sensitive to the varying load impedance. Often a trade-off
between acceptable losses and other characteristics, such as the tuning
range have to be made [62, 121, 123, 191].
For example, the losses of the capacitor in the example of Figure 4.4b
reach a minimum when the load impedance parallel to the capacitor is
low and almost real-valued. However, a low load resistance cannot be
tuned much with a shunt reactance, since the parallel connection of a low
resistance and a reactance inevitably result in an impedance having a low
real part and a low imaginary part. In [121], the location of a pin diode
switch on a shorted section of microstrip line loading a patch antenna was
optimized so that the ratio of the relative frequency shift and the relative
losses was maximized.
The power dissipation is crucial in the case of tunable components, but
also non-tunable circuit elements such as chip inductors can be vulnerable. As an example, let us consider the antenna presented in [57]. Very
compact CCEs were used in the design, with 824–960 MHz as the lower
frequencies of operation. The inherent impedance of this kind of antenna
55
Frequency tunable CCE antennas
at the low frequencies is highly capacitive and the radiation resistance
can be very low. From the results of [VI] we can deduce that the matching
circuit of this kind of antenna is prone to increased power losses. Indeed,
the measured total efficiency of the antenna of [57] at these frequencies
was only ηtot ≈ 30 – 40% although only fixed-value reactances were used
as the matching components.
4.3.2
Nonlinear phenomena
In addition to losses, another possible problem with the use of semiconductor tuning components is non-linearity. Distortion is a notable challenge at the highest transmission power levels of mobile communication
systems [121, 123].
In [IV] it was studied how the impedance environment around a semiconductor based switch (HEMT in this case) can affect the intermodulation distortion in the switch. An important conclusion was that the switch
in a high impedance environment caused much higher distortion than a
switch in a low impedance or 50 Ω environment. In order to avoid high
distortion, the switches should thus not be located in high impedance environments in the tuning circuit. This requirement partly contradicts the
requirement for low losses, as the best efficiency of a switch is sometimes
achieved in a high impedance environment [III].
A demonstrator tuning circuit for a CCE antenna in [IV] was built to
show the importance of acknowledging the linearity issue in the design of
the tuning circuit. The circuit was otherwise based on the general tuning topology shown in Figure 4.1, but the switch on the antenna side
was replaced by a filtering circuit. The switch would have been in a
high impedance environment at the low frequencies, like in [146], and
vulnerable to voltage distortion. The third order intermodulation distortion (IMD3 ) of the demonstrator design at 900 MHz was measured and
the corresponding third order input intercept point (IIP3 ) was calculated.
The results were compared to the earlier design of [146]. The IIP3 in
the tuning circuit of the antenna in [IV] was +57 dBm, which was 17 dB
higher than in the circuit of [146]. This improvement was mainly due to
replacing the vulnerable switch with a linear circuit. As described in the
previous section, also the losses of this design were lower than in [146].
However, the linearity of the MEMS switches in [V] was still much better.
The corresponding IMD3 measurement of the antenna and circuit in [V]
indicated an IIP3 of +67 dBm at 900 MHz.
56
5. Antenna as a part of a radio system
Although the antenna is an irreplaceable part of a radio system, it can
never be a device operating on its own. The antenna is a kind of adapter
or transformer which transforms radiowaves travelling in free space (open
medium) to radiowaves travelling in guided medium, and vice versa. Antennas have two kinds of interfaces to the outside world, radiation interfaces and guided wave interfaces.
In a mobile handset, the radiation interface means everything that is
in the proximity of the antenna, including the mobile terminal itself. The
wires and circuits, camera, microphone, speakers, battery, etc. can affect the operation of the antenna or vice versa. Other antennas residing
on the same handset chassis can cause mutual coupling, which is especially challenging to compensate when the antennas are electrically close
to each other and operate at the same frequencies (see [5] and the related
references therein). Actually, in many communication standards today
and especially in the future, the existence of MIMO operation mode is
required. For example, the LTE specification includes MIMO option in
downlink operation [60], and the LTE-A specification will involve MIMO
in both downlink and uplink operations [61].
Fulfilling these specifications will thereby require future work also with
respect to the antennas presented in this thesis. The implementation of
CCE antennas in MIMO handsets is an important current and future research topic which hopefully benefits from the results of this thesis. Some
examples of multi-antenna designs using capacitive coupling elements already exist [192, 193].
Also adjacent antennas operating at different frequencies can cause troubles for the antenna operation, if their presence is not taken into account
in the antenna design. In studies by many different research groups
[57, 85, 116], two CCE antennas intended for different frequencies on the
57
Antenna as a part of a radio system
same chassis were used with frequency selective (filtering) matching circuits to sufficiently isolate the lower and higher frequencies.
Another aspect of the radiation interface is the nearby space around the
handset, which consists of a lot more than just ”free space”. The user of
the mobile handset will very likely touch the mobile terminal while using
it. The ”user effects” can severely disturb the operation of the antenna.
In this thesis [VII,VIII], the mitigation of the input impedance detuning
of the mobile terminal antenna due to user hand effects is studied.
5.1
5.1.1
Mitigation of electromagnetic user interactions
User effects on mobile handset antennas
Antennas for mobile terminals are still often designed to operate optimally in free space conditions, i.e. the handset is assumed to be surrounded by air only. The user of the handset, however, is usually in the
immediate vicinity of the terminal. In the electromagnetic sense, the user
consists of tissues that have high permittivity and moderate effective conductivity. A good example of such material is water, which comprises more
than half of an average human body mass [194]. From electromagnetic
theory it is known that this kind of materials both reflect and absorb radiation.
The vicinity of the user thus causes impedance detuning and power loss
in the antenna, thereby deteriorating the performance of the communication link. If the link quality between a particular mobile terminal and a
base station is poor, the transmit power has to be increased. This leads
to higher energy consumption, shorter battery life, and higher noise, not
to mention the exposure of the user to increased radiated power. Also the
maximum length of the communication link is decreased because of the
user effect. From the antenna point of view, there is thus high motivation to limit the electromagnetic interaction between a mobile terminal
antenna and the user.
The question of the user effect on the performance of an antenna has
been posed already prior to the era of mobile communications [195–198].
Lately the user effect has been a popular research topic, e.g. [199–205].
It has been reported for example that up to 70 % of the radiated power
of the mobile terminal antenna might be absorbed to the user’s hands
58
Antenna as a part of a radio system
[201]. In [VIII], almost 80% absorption loss was measured in the worst
case. Also the impedance mismatch loss due to impedance detuning can
be several decibels [199], [VI]. In some rare conditions the hand effect
can improve the efficiency of an antenna by causing better impedance
matching conditions [206]. In the case of multi-antenna systems, the user
proximity affects the diversity performance in various ways. These effects
can be positive, negative or insignificant with respect to the performance
criteria (isolation, envelope correlation, diversity gain etc.) [193,207,208].
A simulation study in [202] compared four different self-resonant handset antennas in four different hand grip scenarios, and with the head beside the handset. From the results it can be extracted that the antennas
having about the same dimensions and similar feeding structure behaved
similarly in the user proximity, with respect to the absorption of power to
the user’s hand and head. The observed differences were attributed to geometrical details in the antennas. The direct comparison of these results
to the CCE antennas presented e.g. in this thesis is not reasonable since
the handset size and the size of the antennas in [202] differ a lot from the
antennas presented in this thesis. Although no comprehensive user effect
comparisons between self-resonant and non-resonant handset antennas
exist, there is reason to believe that there are no major differences in the
power absorption between the two antenna types. As the handset chassis is anyway the main radiating element especially at the low frequencies, antennas with principally similar structure should operate similarly.
Furthermore, as the non-resonant CCE lacks geometrical details, it can
be used to study e.g. the effect of the antenna dimensions on the antenna
performance in a user’s presence, without the specific antenna geometry
complicating the interpretation of the results [VIII].
There is also another important aspect urging to decrease the antenna–
user interaction: radiation safety, as mentioned in Section 2.1. The ICNIRP guidelines for limiting exposure of general public on electromagnetic fields set limits for the so called Specific Absorption Rate (SAR)
which describes the power rate per unit mass (W/kg) at which the human body tissues absorb RF energy radiated by an antenna [34]. The
SAR quantity is usually expressed as a peak value averaged over 1 gram
or 10 grams of tissue. The governments around the world require that the
consumer products sold in their respective territories satisfy the strict exposure limits set for the general public, including large safety margins.
For example, in EU the maximum allowed peak SAR in the head, aver-
59
Antenna as a part of a radio system
aged over 10 grams of tissue, is 2 W/kg for the general public [35], but 10
W/kg for workers [209]. In this thesis, the SAR values of the prototype
antennas are always only investigated in order to ensure that also the
safety limits are satisfied.
Since the main focus of this work is on studying the impedance matching
of the CCE antennas, the power absorption is only briefly discussed from
the overall antenna performance point of view [VII, VIII]. The mitigation
of particularly the impedance mismatch caused by the proximity of the
user is addressed. In general, the hand causes more severe impedance
detuning than the head [200], so mainly the hand effects are studied.
There are two approaches on the mitigation of the impedance detuning:
Avoiding the detuning from happening or compensating it after detecting
the detuning. Many other studies discuss particularly the minimization
of SAR and/or the power absorption in the user. This can be done with
shielding metallic strips or plates [210–213], or by filling the antenna volume with ferrites [214, 215]. The idea in these studies is to direct the
strongest electromagnetic near fields of the antenna away from the head
of the user, or, as in [213, 216], either away from the head or away from
the hand, depending on the mode of usage.
5.1.2
Compensation of impedance mismatch
The compensation of impedance detuning generally requires detection of
the changed impedance conditions due to the user effect. The detection
can be made directly from the signal by monitoring the signal strength or
the reflection coefficient [184, 186, 217]. Alternatively, an indirect detection system can be used. For example, a separate proximity sensor can be
used to detect a user’s finger close to the antenna element [218, 219]. The
possible sensor should be carefully designed in order to avoid the sensor
affecting the antenna operation [220].
A mismatch compensation mechanism can then be adaptively used. Automatic/adaptive antenna tuning units [182, 184, 186] can be used for the
purpose of compensating the user-originated impedance mismatch [143].
This approach has also the advantage that the antenna tuning unit can
be used as a general tuning circuit for bandwidth enhancement of the
antenna as well [140].
Alternatively, the mismatch compensation does not need to be made
with impedance tuning. Antenna diversity is one possible way to realize
this. Antenna elements can be placed in different locations on the mobile
60
Antenna as a part of a radio system
terminal, and through antenna switching one can choose one of the antennas to operate at a time. The antenna suffering least from the user
proximity would be selected [213,221–223], [VII]. Improvement of several
dB in total efficiency of the antenna system can be achieved with these
methods, at the cost of either increased antenna volumes or decreased
instantaneous bandwidths.
5.1.3
Proactive methods against impedance mismatch
Countermeasures against impedance detuning can also be made passively
without any mismatch detection. These methods aim at a more robust
design of the antenna and the mobile terminal.
A simple way to limit the impedance mismatch caused by a user is to
allocate low-permittivity material around the antenna within the mobile
handset. An example of such design is the plastic ’bumper’ around the
Apple iPhone 4, provided as a fix for the poorly performing antenna when
the user holds the phone in a specific way, a problem widely reported in
the news in 2010. The issue also caught attention among the antenna researchers [139]. Allocating near-empty space around the mobile handset
is probably not the most attractive option from the user experience point
of view, so more elegant methods should be considered.
A distributed antenna system was presented in [224, 225]. This method
resembles the antenna switching approach, but it works passively without
sensing of the impedance environment. The two antenna elements are fed
from a single feed with a 90 degree phase delay introduced in one of the
elements. The antenna elements operate well together in free space, and
are more robust against the finger loading effects than one element alone.
When one of the elements is obstructed by a finger, the other element can
still radiate well [225].
Alternatively, prediction of the user effect could provide a solution to the
mismatch problem with a user’s hand. If the antenna is directly designed
to work both in free space and with some typical use cases, the average
performance loss due to mismatch could be limited to an acceptable level
[VII, VIII].
In the case of non-resonant antennas, the proactive mitigation of the
user effect can be made by optimally selecting the dimensions, shape and
location of the antenna element [VII], or by designing the matching circuit
so that the typical user effect is statistically anticipated during the design
process [VIII]. It was shown in [VII] that by minimizing the probability of
61
Antenna as a part of a radio system
the interaction between the antenna element and the index finger when
the antenna is at the top part of the handset, part of the finger loading
effect could be avoided. This was realized by designing the shape of the
CCE so that the surface of the CCE and the projection of the index finger
on the surface of the handset had as little parallel area as possible. Correspondingly, a small CCE in the bottom corner of the chassis closest to the
thenar muscles was the most vulnerable to the palm effects. Differences
of several decibels in total efficiency were observed between four slightly
different CCE structures placed either at the top or the bottom part of the
handset, measured with three different hand grips. About 0 – 2 dB differences in matching efficiency were obtained, and it was observed that also
the absorption loss is affected by the design of the shape of the antenna
element, with up to 4 dB differences in radiation efficiency between the
different designs.
In [VIII], novel matching circuits for CCE antennas were designed so
that the effect of the user’s hand would not detune the input impedance
on the Smith chart towards the edge, but rather towards the center of
the Smith chart. Experience had shown that the hand and especially the
index finger in the immediate proximity of the antenna will shift the antenna resonances downwards in frequency due to parasitic shunt capacitance, and the impedance trace loops on the Smith chart become smaller
as a result of resistive loading. This assumption is also supported by the
circuit modeling approach discussed in Section 2.2 [47].
The free space frequency response of the circuit in [VIII] was designed
so that it had dual-resonant response at both lower and higher frequencies of mobile communications. The impedance trace formed double loops
on the Smith chart surrounding the center of the chart so that the effect of the hand decreasing the loop size would more probably cause the
impedance matching to improve rather than degrade. It was shown that
this kind of frequency response leads to more robustness than a single
resonant narrowband response. In the case of a typical talking grip with
the index finger touching the top part of the back cover of the handset,
the impedance matching of the multi-band CCE antennas was actually
improved. To obtain these results, the design of the matching circuit was
optimized in parallel for the free space case and the user interaction situation.
62
Antenna as a part of a radio system
5.2
Interface between antenna and transceiver
The other interface of the antenna in contrast to the radiation interface is
the guided wave interface between the antenna and the radio transceiver.
Usually the most important aspect in implementing this interface is to
realize impedance matching over all desired frequencies of operation. Depending on the RF front-end architecture, there may also be other features that should be taken into account. Here the number of antenna
feeds in the RF front-end interface is addressed.
5.2.1
Single-feed and multi-feed architectures
Multi-band operation of non-resonant mobile terminal antennas has been
often realized by using separate antenna elements for lower and higher
frequencies, with each antenna element having its own matching circuit
[57, 116, 226]. The same approach has been used also with some selfresonant antennas [141, 227]. In the case of non-resonant antennas, the
popularity of using multiple antenna elements is probably due to the fact
that it is easier to implement wideband impedance matching separately
for different antenna elements than one matching circuit for single element with non-contiguous passbands.
The multiple antenna elements can be fed from a single combined feed
[57,116], which is the conventional approach also used with single-element
antennas, or the RF feeds for each element can be independent [141, 227].
The conceptual difference between the single-feed and the multi-feed approaches is related to the RF front-end architecture of the multi-system
radio device. Figure 5.1 shows a simplified system block level representation of a conventional RF front-end design in a wireless terminal. The
number of antenna elements can be one or higher, but the key item is
that there is a combined antenna feed for all antennas, and the selection
of the frequency band at the front-end side is made with a single-pole
multi-throw (SPMT) switch. Each frequency band or communication system has its own specific radio frequency front-end hardware: pre-selection
and duplexing filters, power amplifiers (PA), and the RF integrated circuit
(RFIC) part.
The dual-feed or multi-feed configuration has recently gained popularity [228]. A system block diagram of a dual-feed antenna with the RF
front-end is shown in Figure 5.2. The difference between the traditional
approach and the dual-feed approach is that instead of a single SPMT
63
Antenna as a part of a radio system
Front-end
Multiband
Alternative
antenna arrangements
RX 1
Matching
PA
TX 1
RFIC
High f
Low f
SPMT
Switch
RX n
PA
Diplex/
Match.
TX n
Figure 5.1. Traditional RF front-end design in a wireless terminal with single multiband
antenna element, or separate antennas for low and high frequencies.
Switch
(Matching)
RX 1
Antenna(s)
PA
Front-end
TX 1
RFIC
RX n
PA
TX n
Switch
Figure 5.2. Dual-feed RF front-end design.
switch, the selection of the frequency band is made by two switches, or
up to as many switches as there are antenna feeds. This provides a
significant benefit over the single-feed configuration. The multi-feed approach enables inter-band carrier aggregation (CA), as the configuration
supports simultaneous use of multiple transmitters or receivers without
hard-wired matching of the duplexers related to the frequency pairs using
CA [228]. On the other hand, the multi-feed approach does not decrease
the number of RF front-end components from the single-feed case. Each
system usually still needs its own receiver and transmitter chain. Some
systems may partly share the hardware, though [229].
Most of the antennas presented in this thesis have been designed assuming the more conventional single antenna feed interface. In [III] and
Section 3.2 of this thesis, the multi-band CCE antenna with dual branch
matching circuit was successfully implemented both for the single antenna feed and the dual antenna feed RF front-end interfaces. Also the
broadband tunable antennas in [IV, V] would suit the dual-feed front-end
since they also use the dual-branch approach. In the case of using the
circuits of Figure 4.2, the switch 1 in the circuit could be omitted, which
would also decrease the losses of the tuning circuit.
64
Antenna as a part of a radio system
Broadband
&
Multi-band
Antenna
TX
Matching
RFIC
RX
Figure 5.3. Vision of an RF front-end of a software-defined radio.
However, the single-feed configuration should not be discarded, since
it is a natural choice for the software-defined radio [230]. It is possible
that in the future the RF front-end IC can be made flexibly tunable over a
large tuning range, in which case a major part of the frequency-selectivity
would be handled by the IC. Already very wideband SDR receivers with
multi-octave frequency bands have been presented. A concept of tunable
SDR receiver from 800 MHz to 6 GHz without RF pre-selection filtering
is presented in [230, 231]. Also in [232, 233] the receivers exhibit wide
frequency operation across the frequencies of the cellular systems.
Every communication system could ideally share the receiver and transmitter components, as the highly tunable RFIC would replace the switches
and pre-selection filters (Figure 5.3). This would be close to the ideal
for software-defined radio. Of course, this vision is still far from being
practically realizable in mobile communications. The bottlenecks are the
required level of selectivity between transmission and reception, as well
as the suppression of harmonic blocking signals [229]. Currently only
surface and bulk acoustic wave (SAW/BAW) filters provide the needed
steep slope of the frequency response to realize the duplexing in mobile
terminals [234, 235]. These components cannot be implemented on the
transceiver RFIC chip as such, and their tunability is very poor [236]. On
the other hand, the integration of preselect filters in the RFIC by using
alternative technologies has already been demonstrated at selected frequency bands [229]. Another proposed alternative to replace SAW filters
is the single-pole single-zero notch filter with RF MEMS for tuning [237].
In all of the antenna prototypes presented in this thesis, the design
impedance at the RF front-end interface has been 50 Ω for practical reasons. The measurement systems use 50 Ω system impedance, so the antenna matching should be designed to that impedance interface. However,
the optimal receiver or transmitter impedance could be far from 50 Ω. It
could be frequency dependent, and also different for the receiver and the
transmitter sides. Thereby the co-design of the antenna and the RF front-
65
Antenna as a part of a radio system
end should be promoted. As a versatile antenna concept, the CCE antennas would be easily adapted also to these unconventional antenna feed
impedances.
66
6. Summary of publications
[I] Broadband equivalent circuit model for capacitive coupling
element based mobile terminal antenna
The impedance behavior of a simple CCE antenna for mobile handsets
is modeled with a broadband equivalent circuit based on the modeling
of the lowest-order wavemodes of the CCE and the handset chassis as
RLC resonators. The impedances of the model and a simulated antenna
agree very well at the entire UHF band 300-3000 MHz, for which the
model was designed. In addition to the behavioral accuracy, the model
provides insight for analyzing the significance of the different parts on the
radiation and on the impedance and bandwidth potential of the antenna.
[II] Inherently non-resonant mobile terminal antenna with
simultaneous multi-band operation
A novel matching approach for an inherently non-resonant CCE antenna
is presented. The presented matching circuit is based on using two feeding strips and a dual-branch diplexing matching circuit with a single combined feed for a single coupling element. The locations of the feeding
strips are selected to provide the optimal bandwidth both at the lower
(698-960 MHz) and at the higher (1710-2690 MHz) operation frequencies.
The antenna exhibits good impedance matching (6 dB return loss) and
radiation properties (better than 55 % total efficiency).
[III] Capacitive coupling element antennas for multi-standard
mobile handsets
Three multi-band capacitive coupling element antennas based on the same
simple geometrical structure are presented. Experimental results show
how the versatile structure is adapted successfully to single-feed and dual-
67
Summary of publications
feed interfaces between the antenna and an LTE-A transceiver RF frontend. The operation frequencies are 698-960 MHz and 1710-2690 MHz.
The third prototype operates with a single feed interface of a softwaredefined radio receiver interface (operation frequencies 750-2500 MHz).
The performance of the antennas is competitive in comparison with the
state-of-the-art in mobile handset antennas.
[IV] Minimization of power loss and distortion in a tuning
circuit for a mobile terminal antenna
One octave tuning range (900 MHz to 1800 MHz) for a compact CCE antenna is implemented with a band switching tuning circuit designed to
decrease the rather high power loss and distortion observed in an earlier
design. The distortion of a HEMT switch is studied in different impedance
environments and it is concluded that certain load impedances should be
avoided if good linearity is desired. Also the losses of the tuning components depend on the respective impedance environments, and the ideal
impedances from the losses point of view often contradict with the requirements for good linearity. A prototype antenna is built to show that
both linearity and losses of a broadband antenna tuning circuit can be
controlled by placing the tunable components only in favorable impedance
environments.
[V] Frequency-reconfigurable mobile terminal antenna with
MEMS switches
A tunable compact CCE antenna with one octave tuning range for quadband mobile handset operation (GSM850/900 and GSM1800/1900) is presented. The band switching tuning circuit for the CCE antenna is based
on two magnetically actuated latching MEMS switches and two independent matching circuits isolated by the switches. The performance of the
built prototype antenna is compared to similar tuning circuits which used
HEMT based switches. The MEMS switches outperform the semiconductor circuits in both efficiency and linearity.
[VI] Power Dissipation in Mobile Antenna Tuning Circuits under
Varying Impedance Conditions
The paper offers a critical approach in determining the net power dissipation in mobile handset antenna tuning circuits. It is shown how the
68
Summary of publications
power loss of tuning circuit components may be very different from the
datasheet values when the components are placed in different impedance
environments, also including strongly varying load impedances. The paper discusses the importance of identifying the challenging impedance environments in a tuning circuit, as well as the components which are more
sensitive to power losses than alternatives. Countermeasures against the
power losses already in the design stage of the circuits are discussed.
[VII] Avoiding the interaction between hand and capacitive
coupling element based mobile terminal antenna
Interaction between a CCE based mobile terminal antenna and anthropomorphically shaped hand phantoms is investigated at GSM900 and
GSM1800 frequency bands through an experimental study. Coupling elements with slightly different shapes and sizes, as well as CCE locations
at the top or at the bottom part of the chassis are compared. The purpose
is to study the significance of these antenna characteristics on the input impedance detuning and also on absorption loss of the antenna when
the antenna is subject to hand loading effects. Also antenna switching is
tested as an option to compensate the impedance mismatch. The results
show that there can be differences of several decibels in total efficiency
between the slightly different CCE structures, indicating that it is worthwhile to consider the probable hand effects when designing the antenna
elements.
[VIII] Naturally non-selective handset antennas with good
robustness against impedance mistuning
Robustness provided by the matching circuit of a CCE antenna against
the antenna input impedance detuning caused by the hand of a mobile terminal user is studied. The frequencies of interest are those of mobile communications. Experimental data is presented to show the significance of
selecting a matching circuit design that is less sensitive to the impedance
detuning than a generic reference matching circuit. With a naturally nonselective antenna and a suitable broadband matching circuit design, the
impedance detuning can be limited to a moderate level.
69
Summary of publications
70
7. Conclusions
This thesis studies and develops methods to meet the requirements that
the mobile communication systems set on the handset antennas. The antenna structures used in the thesis are based on the concept of inherently
non-resonant capacitive coupling elements. The work contributes to the
mobile antenna research field by strengthening the position of the CCE
antennas as contenders for the primary antenna solution in future mobile
handsets.
The CCE antennas have various advantages. They can be made relatively compact in size, thanks to the efficient utilization of the radiating mobile terminal chassis. Furthermore, the radiating structure can
be made non-selective with respect to the frequency response, which enables very wide tuning ranges for the antenna. However, in order to
make a non-resonant antenna operate well at any frequency, its inherent impedance needs to be matched to the impedance of the transceiver
at each frequency band. This work mainly concentrates on improving the
operation of CCE antennas by developing impedance matching and tuning
solutions that would also suit communication systems of the future such
as software-defined radio and cognitive radio. The properties of non-selfresonant antennas are succesfully utilized in developing novel multi-band
antenna structures for mobile terminals.
The presented work includes a circuit model improving the physical understanding on the operation of a capacitive coupling element antenna
[I]. The model can be utilized e.g. in analyzing the user effect of the antenna, and finally, in improving the CCE design. Geometrically simple
CCE antenna structures are used with various matching circuits capable of simultaneous operation at multiple frequency bands across the cellular communications frequencies [II, III]. These antennas are based on
the novel self-diplexing impedance matching circuit approach and a dis-
71
Conclusions
tributed feeding structure for the CCE developed in this thesis work. The
performance of these antennas is state-of-the-art, comprising a compact
structure, broad operation frequency bands and good efficiency.
Also the broadband tuning of CCE antennas is demonstrated with wide
discrete and continuous tuning ranges. The practical challenges of using real tuning components in varied and varying impedance conditions
for the tuning components are discussed [VI], and realized prototypes
of broadband tunable antennas are presented [III, IV, V]. These results
prove that the CCE antennas can also be used in future mobile handsets
with SDR based communication systems, albeit the improvement of practical tuning components is highly desirable.
Finally, it has to be remembered that an antenna is always just one
part of a communication system, and it is not isolated from the other RF
and non-RF parts of the system. The antennas in a mobile terminal are
subject to electromagnetic interaction with the user of the handset. This
interaction deteriorates the operation of the antenna through absorption
of power and detuning of the antenna impedance. In this work, the problem is tackled by developing methods for avoiding the impedance detuning
[VII, VIII]. Antenna switching diversity is used for decreasing the effect
of the user’s hand on the impedance matching. Coupling element structures and matching circuits producing improved robustness against the
mistuning due to the user effect are developed and studied. Practical issues arising from the system requirements and RF front-end interfaces
should be remembered when designing antennas for specific communication systems [III].
The outcome of each publication in this thesis takes the mobile handset
antenna research step by step closer to the unattainable ideal antenna
mentioned in Section 1.2. Based on the results obtained e.g. in this thesis,
work on the multi-band and multi-standard antennas for wireless devices
continues in the research group. Important topics include multi-band
multi-antenna structures with good diversity performance, co-design of
an antenna and a SDR transceiver with optimized interface, formulating
theoretical background for the multi-band matching of CCE antennas, etc.
As the mobile communication industry evolves rapidly, also totally new
antenna types are considered. Who knows how the new mobile terminals
will look like in 10 years?
72
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