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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS, VOL. 1, NO. 3, SEPTEMBER 2007
A Low-Power Blocking-Capacitor-Free
Charge-Balanced Electrode-Stimulator Chip
With Less Than 6 nA DC Error for 1-mA
Full-Scale Stimulation
Ji-Jon Sit, Member, IEEE, and Rahul Sarpeshkar, Senior Member, IEEE
Abstract—Large dc blocking capacitors are a bottleneck in
reducing the size and cost of neural implants. We describe an
electrode-stimulator chip that removes the need for large dc
blocking capacitors in neural implants by achieving precise
charge-balanced stimulation with 6 nA of dc error. For cochlear
implant patients, this is well below the industry’s safety limit of
25 nA. Charge balance is achieved by dynamic current balancing
to reduce the mismatch between the positive and negative phases
of current to 0.4%, followed by a shorting phase of at least 1ms
between current pulses to further reduce the charge error. On
6 and 9 V rails in a 0.7- m AMI high voltage process, the
power consumption of a single channel of this chip is 47 W when
biasing power is shared by 16 channels.
+
Index Terms—Biphasic current pulse, charge balancing,
cochlear implants, electrode stimulation, neural implants.
I. INTRODUCTION
ANY biomedical implants currently convey information
to nervous tissue by pulsatile current stimulation via an
implanted microelectrode array. Examples of such implants include cochlear implants for the deaf, deep brain stimulators for
Parkinson’s disease and retinal implants for the blind. In order
to avoid irreversible electrochemical reactions leading to toxic
by-product generation and eventual rejection of the implant,
the pulse of current is typically made up of equal-sized negative and positive phases, with the goal of delivering no net
charge through the electrode at the end of the pulse [1]. A perfect balance of zero net charge is difficult to achieve because
current source and sink drivers will typically be mismatched
by 1%–2% in the fabrication process even when great care is
taken to ensure good matching in the layout and design. Studies
have shown that a residual dc current of more than 100 nA is
M
Manuscript received June 29, 2007; revised October 23, 2007. This work was
supported in part by the Packard Foundation and in part by the McGovern Institute for Neuro Technology (MINT) Program at the Massachusetts Institute
of Technology, Cambridge. This paper was recommended by Associate Editor
B.-D. Liu.
J.-J. Sit was with the Analog VLSI and Biological Systems group at the Massachusetts Institute of Technology, Cambridge MA 02139 USA. He is now with
Advanced Bionics Corporation, Sylmar, CA 91342 USA.
R. Sarpeshkar is with the Analog VLSI and Biological Systems Group,
Massachusetts Institute of Technology, Cambridge MA 02139 USA (e-mail:
rahuls@mit.edu).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TBCAS.2007.911631
highly correlated with neural tissue damage [2], and since stimulation currents may need to be as high as 1 mA, a dc error of
0.01% is required. Some means of correcting any charge imbalance such that it presents 100 nA of dc error must therefore
be implemented.
One of the simplest solutions is to insert a large dc blocking
capacitor with a typical shunt impedance of 5G in series
with each electrode, to guarantee that negligible ( 1 nA) dc
current will be able to flow across the electrode. However, to
handle the clinical levels of charge that are required for stimulation, which in cochlear implant users has a median maximum comfortable level around 15 nC/phase [3], the size of a
typical blocking capacitor must be 100 nF or larger so that
the voltage drop across the capacitor does not add appreciably
to the voltage compliance required for driving the electrode
impedance. The size of a large (100 nF) blocking capacitor
is hugely detrimental for implantable electronics, which is required to be small. Hence, safe levels of charge-balance that can
be achieved by capacitor-free precision electronics is attractive.
Since the cost of packaging is often a highly nonlinear function
of the size of the implant, capacitor-free stimulation reduces cost
as well.
Several methods to achieve safely charge-balanced stimulation without using large dc blocking capacitors have been proposed. The first and perhaps simplest solution is to short out
the electrode between current pulses [4]. The degree to which
shorting is effective will depend on the initial charge imbalance,
the electrode time constant of discharge, and the time available
for discharge, set by the minimum inter-pulse interval. At stimulation rates as high as 2000 pulses-per-second (pps)/channel,
shorting has been shown in clinical trials to achieve 100 nA dc
error and safe long-term electrical stimulation [5], [6]. However,
for reasons of patient safety, it is desirable to achieve charge balance which is significantly lower than 100 nA, and more precise
than that achieved by shorting alone. We propose to do this by
reducing the initial charge imbalance below the usual 1%–2%
mismatch error, and getting closer to the level produced by dc
blocking capacitors (on the order of 1 nA dc error).
Besides shorting, the use of many tiny charge dumps from
filling and emptying four small-sized (25 pF) blocking capacitors at 10 MHz has been proposed to reduce but not eliminate the blocking capacitors [7]. The technique was verified in a
discrete implementation and simulated in CADENCE. Experimental evaluation of such ultrahigh-frequency stimulation in actual nerve environments that exhibit refractoriness and lowpass
1932-4545/$25.00 © 2007 IEEE
SIT AND SARPESHKAR: LOW-POWER BLOCKING-CAPACITOR-FREE CHARGE-BALANCED ELECTRODE-STIMULATOR CHIP
filtering is needed, especially to ensure that stimulation thresholds are not adversely affected. Noting that a large dc blocking
capacitor functions as a charge integrator in feedback, an alternative method of charge balancing by charge-metering was proposed in [8], using a switched-capacitor integrator to integrate
the current transacted across a small series resistor, and feeding
back finely discretized charge packets to equilibrate the charge.
As mentioned, the success of this technique then depends on
the ability to minimize the charge-injection, noise, mismatch,
leakage and other sampling errors to achieve highly precise and
stable feedback, which can make for a power-hungry solution
in practice.
Another approach is to perform feedback current DAC calibration by sampling and holding a correction current to improve
the matching of the current drivers [9]. This technique is similar to the dynamic current matching that we will employ, and
achieves 90 nA of dc error, although in this work only current and not charge was matched. Finally, a feedback strategy to
monitor any residual voltage on the electrode capacitance after
each biphasic pulse and then apply stimulation currents until
the residual voltage falls within a safe window ( 100 mV) was
proposed in [10]. The precision of charge balance in this case is
then a function of the electrode capacitance.
In cochlear implants, a typical RC series impedance presented
by an electrode can be as high as 10 k and 10 nF, which
when driven with a 1mA pulse for 30 s/phase, requires at least
V of output compliance on a single rail alone. Thus,
it is crucial for any practical stimulator design to have sufficient
voltage compliance to accommodate both the variability in electrode impedances and the variability in patient thresholds for
comfortable stimulation levels [11]. The particular values used
in our paper are of primary relevance to cochlear implants, but
our technique is widely applicable to all biomedical implants
that employ electrode stimulation.
In this paper, we present a stimulator chip that employs dynamic current matching to reduce the typical 1–2% mismatch
error because the same pMOS device which dynamically stores
an nMOS current magnitude is later used to output the positive current pulse. As the dynamic current matching employed
only provides 0.4% mismatch on its own, we follow the biphasic
pulse with a shorting phase to achieve 6 nA of dc error.
Finally, as in all implanted electronics, fail-safes have to be
designed to protect the user from device failure. These are not
considered in this paper but similar ideas from [7] and [12]
in providing multiple redundant devices and voltage window
checking can and should be implemented.
This paper is organized as follows. Section II provides an
introduction to the electrode model that we use in this paper.
Section III presents our strategy for achieving precise charge
balance. Section IV provides a feedback stability and noise analysis of the dynamic current-balance loop employed in the first
phase of charge balancing. Measurements from our fabricated
stimulator chip are presented in Section V. Section VI concludes
the paper by summarizing our contributions.
II. BACKGROUND ON ELECTRODE STIMULATION
A. Electrode Modeling
When addressing the problem of delivering balanced charge
to an electrode, it is helpful to consider a basic circuit model that
173
Fig. 1. Simple circuit model for a typical electrode in a neural implant.
will govern the electrical response of an electrode to biphasic
current stimulation, shown in Fig. 1.
In this simple model,
is the solution spreading resistance,
which is well determined by the resistivity of the fluid (set by
ionic species in solution).
is the double-layer capacitance,
created by the accumulation of tightly adsorbed ions at the
electrode surface and more loosely attracted ions in a diffuse
layer behind it. Finally,
is the Faradaic resistance, which is
governed by diffusion of reactive species to the electrode for
is indicated as a
charge-transfer reactions. The resistance
time-varying variable resistor because its value varies based
on the dynamics of redox charge-transfer reactions occurring
at the electrode. For the purposes of our analysis, we shall
remains large
M over the time scale of
assume that
a stimulation pulse
s , and can therefore be neglected.
is, however, the
Current through the Faradaic resistance
source of toxicity when there is no long-term charge balance.
It is important to note that the values of
and
will vary
depending on the material and geometry of the electrodes used.
In the cochlear-implant devices that we chose as a reference for
our design, the values of
and
have an upper bound of
20 k and 15 nF, respectively.
B. Biphasic Pulse Generation
In this work, we have chosen to implement only cathodic
(negative) stimulation followed by anodic (positive) stimulation, in accordance with the conventional understanding that the
negative pulse serves to depolarize the nerve and evoke action
potentials by lowering the local extracellular potential relative to
the membrane potential [13]. In contrast, the positive pulse tends
to hyperpolarize the nerve and elevate sensory thresholds, but is
required for charge balance. Most stimulation schemes therefore default to negative-first stimulation, which we also adopt
here.
In practice, many implants will also include a short interphase interval, i.e., a no-stimulus section between the negative
and positive pulses, which can mitigate the hyperpolarizing effect from the trailing positive pulse. The interphase interval is
also helpful in lowering crossbar conduction that wastes implant
power. However, as simulations indicated that charge balance
and power consumption were not significantly affected by the
interphase interval in our design, it was omitted for simplification.
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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS, VOL. 1, NO. 3, SEPTEMBER 2007
Fig. 2. Timing diagram showing the order of operations. Gray blocks indicate a short delay.
Fig. 3. (a) Sample phase in dynamic current balancing: SAMP1 and SAMP switches are closed. (b) Hold phase in dynamic current balancing: SAMP1 opens
first, followed by SAMP, after which HOLD is closed.
III. STRATEGY TO ACHIEVE PRECISE CHARGE BALANCE
A. Overview
Fig. 2 shows that the strategy we employ to achieve charge
balance is two-fold: First, a dynamic current-balancing phase
is performed to match a pMOS current source with an nMOS
current sink. Second, after the biphasic pulse is generated, any
residual charge error left on the electrode is shorted to ground.
A biphasic current pulse is initiated by the falling edge of
, shown in Fig. 2. The use
an asynchronous trigger signal
of Vox enables an asynchronous stimulation strategy, which
encodes phase information for conveying better music and
speech perception in cochlear implants [14]. Immediately after
Vox arrives, a sample-and-hold (S&H) phase begins, shown as
in Fig. 2. In this phase, the pMOS gate voltage
required to support an nMOS current
is sampled on a hold
capacitor as shown in Fig. 3(a), and exploits the same basic
idea as a dynamic current mirror [15].
and
form a source-follower that diode-connects
transistor M1, and is biased with enough current to drive at a
to
reasonable speed. The source-follower also level-shifts
, which keeps the active-cascode formed
a diode drop below
and
in saturation. After the circuit has settled,
by
is opened first.
is then opened followed by the
switch as shown in Fig. 3(b). This circuit
closing of the
of Fig. 3(b) incorporates a closed-loop switched-capacitor
sample-and-hold [16] which attenuates noise and other disturby the gain of the loop around transconductor
bances to
. Errors due to the charge injection of the
switch
switch limit the precision of the sample and
but not the
hold [16].
The sampling phase preceding the hold phase is 4 clock
cycles in duration, counting from the first rising clock edge
. Then, a negative pulse is generated by closing the
after
OUTPUT NEG PULSE switch for 8 clock cycles. A positive
pulse immediately follows the negative pulse via the closing of
the OUTPUT POS PULSE switch for the next 8 clock cycles.
Finally, after a small delay, the SHORT switch is closed to
discharge any residual charge on the electrode. This scheme
was chosen such that an asynchronous stimulation pulse would
have a fixed latency of 4 clock cycles, and an additive variable
. Since
latency between 0–1 clock cycles after the arrival of
1 clock cycle is 4 s, the variable latency is a small fraction of
the output pulsewidth.
B. Low-Leakage Sample and Hold
A conventional differential-mode closed-loop sample-andhold is shown in Fig. 4(a).
Differential sampling allows for partial cancellation of
charge injection errors of the SAMP1 switches in a closed-loop
sample-and-hold. An additional improvement, shown in
Fig. 4(b), includes the addition of a low-leakage sampling
SIT AND SARPESHKAR: LOW-POWER BLOCKING-CAPACITOR-FREE CHARGE-BALANCED ELECTRODE-STIMULATOR CHIP
175
Fig. 4. (a) Conventional differential-mode closed-loop sample-and-hold: Vref is also sampled onto a capacitor. (b) Improved with a low-leakage switch.
Fig. 5. Full circuit of differential, low-leakage analog storage within the dynamic current mirror, showing the use of diffusion-MOS (DMOS) devices with lightly
doped drains (with up to 35 V of breakdown voltage, indicated by a thick drain terminal) to withstand large voltage drops where needed. The reference voltage
Vref is buffered from Vref in by an identical Gmh transconductor and bypassed by a capacitor Cref , shown in gray.
0
switch adapted from the ultralow power analog storage cell of
[17] and [18]. It works to minimize the switch leakage onto
node
by matching source and drain voltages (at
), thus
nullifying leakage diffusion currents between source and drain
[17], [18]. Note that this equality occurs automatically for the
switch, but must be enforced explicitly for the
switch, by switching its rightmost terminal to
during the
phase. The SPDT switch labeled
enforces the equality. The “d” suffix indicates that its switching
time is slightly delayed from the onset of sampling and holding,
the reason for which is explained in the next section.
C. Full Circuit Implementation
The full circuit which implements the differential
low-leakage analog storage within the dynamic current mirror
is shown in Fig. 5.
In order to achieve the high-voltage compliance necessary for
electrode stimulation, DMOS devices available in the Europractice AMI 0.7- m 30 V (I2T30E) process were used to protect
junctions that were exposed to large voltage drops.
Additional circuit details that are required to improve the circuit’s settling time during its sampling phase are as follows.
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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS, VOL. 1, NO. 3, SEPTEMBER 2007
Fig. 6. Timing diagram showing the switching signals required for the various operations in dynamic current balancing.
is low when the circuit is waiting for the
input
signal, and prebiases
to start out near the positive rail. This
initial condition improves the settling time, because at the onset
of sampling, it is faster for the pMOS source-follower to pull
rather than pull up on it. For the same reason,
down on
switch is closed for
to pull
up to the
the
rail when waiting for the
trigger, as it is faster for
to
pull down on
than to rely on a gradually stabilizing current sourced from
to pull up on
. The
switch
is closed to supply current to the active-cascode, whenever
needs to source current. This saves power in the active-cascode
by turning it off when it is not needed.
is simply
, just as
. A third technique used to
shorten the settling time during the sample phase was to delay
the closure of all the feedback loops in Fig. 5 until turn-on transients had died out. Note that when the switch
is closed,
shoots upwards from the negative rail
to equalize
with
near the positive rail, and thereby creates a large spike
on the value of Iin. This spike, while short-lived, can cause
all the feedback loops to take much longer to settle. Hence,
and also
are both
delayed from switching until slightly after the onset of
and
, in order for the initial turn-on spike to die out before
the feedback loops are closed. A timing diagram which summarizes these timing relationships is shown in Fig. 6.
Select capacitor placement was found to be necessary for robust operation. A capacitor
pF was added to bypass
noise on
. The capacitors
and
were both 1 pF
and helped to stabilize the settling behavior of the active cascode. The capacitor
pF was added to dampen a voltage
spike on
when switching from
to
mode. The
capacitor
pF was necessary to compensate the dynamic mirror feedback loop and improve its phase margin. The
stability of this circuit is further described in the next section.
The schematic of the transconductor
is shown in Fig. 7.
D. Shorting
The shorting phase is implemented by a simple nMOS switch
to ground that is sized to have a small on-resistance
2k ,
which is small in comparison with the electrode resistance.
IV. FEEDBACK STABILITY AND NOISE ANALYSIS
A simplified schematic of the dynamic mirror in closed-loop
sampling is shown in Fig. 8.
was not shown in earlier schematics because it is a parasitic capacitance, but is important for this analysis. We will make
the simplifying assumption that the active cascode is fast and
does not contribute to the loop dynamics, and only increases
transistor by a factor of
to
the output resistance of the
. The block diagram for this loop can then be written out
in Fig. 9(a) and simplified to that of Fig. 9(b).
A. Loop Bandwidth
From Fig. 9(b), the transfer function
can be treated
as unity since the source-follower dynamics are comparatively
fast. We can read off the loop bandwidth of
to
be
, as the
feedback term is only significant above crossover. A Bode plot of the forward transmission
together with the inverse of the feedback transmission,
is shown in Fig. 10.
Since the minimum input current
is set by our application
at 1 A, we can expect an upper bound for one time constant to
mV
A
ns. There may be some
be at most 1 pF
degradation in this value due to a lag effect (reduction in loop
gain) from
falling below unity, but the loop should still
be fast enough to achieve full settling within the 4 clock cycles
allotted, which is 16 s in our application. In general, since the
settling is governed solely by
, we can expect
fast first-order settling.
B. Loop Stability
Since
, it is indicated in Fig. 10 by the
continuation of
as a dotted line, having a right-half-plane
(RHP) zero at
. Since the RHP zero occurs after
crossover by a factor of
, it does not adversely affect
loop stability as long as
is not too small compared with
. Thus, we can see that the loop should be stable with nearly
90 of phase margin.
In addition to the usual pole-splitting compensation provided
by , we can identify two additional sources of improvement
SIT AND SARPESHKAR: LOW-POWER BLOCKING-CAPACITOR-FREE CHARGE-BALANCED ELECTRODE-STIMULATOR CHIP
Fig. 7. Topology of the high-output impedance transconductor
swing from
to
.
Vref Vin
177
Gmh. The output is cascoded and protected with DMOS output devices to allow a wide voltage
, which can be quite slow.
added by the source follower at
The capacitor
helps by inserting a zero at
to partially
cancel this nondominant pole. Secondly, the large sampling capacitor , which is sized at 2 pF, would degrade phase margin if
resistor in series with
not for the zero introduced by the
it. At high frequencies, rather than seeing the capacitive roll-off
due to continue forever, the impedance of and
flattens out at
and hence inserts another zero at
.
The closed-loop sample-and-hold topology of Fig. 4 is therefore well suited to allow stable closed-loop sampling within the
dynamic mirror. The use of pole-splitting compensation and the
combination of zeroes at
and
allow the loop to
be stable over a large range of input currents, down to the minimum value of
A.
C. Noise Performance
Fig. 8. Simplified schematic of our dynamic mirror circuit in the sample phase.
in the phase margin: Firstly, without
, a nondominant pole is
From the block diagram in Fig. 9(b), we see that noise sources
and Iin experiences the full bandwidth of the loop
from
and have a frequency dependence just like Iin. However, noise
sources from
,
and
will be strongly attenuated at
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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS, VOL. 1, NO. 3, SEPTEMBER 2007
Fig. 9. (a) Block diagram of the dynamic mirror feedback loop. (b) Simplified block diagram containing unitless transfer functions within the loop. is defined
as C=Gmh.
Fig. 10. Bode plot of the closed loop transfer function G(s) // (1/H(s)) shaded in gray and the loop transmission L(s), which continues after G(s) as a dashed line.
because of the large gain of
at dc in their feedback path to . To understand this attenuation more clearly and
over all frequencies, we can again draw the forward transmission
together with the inverse of the feedback transmission
, as shown in Fig. 11.
The transmission
can be observed to be the same as
except without the zero at
, and
reveals
the strong attenuation of
at low frequencies. At higher frequencies,
intersects with the rolloff from
, hence
is attenuated by the loop over all frethe noise appearing at
quencies. Although current noise from
does not inject directly into
, it is capacitively coupled through , and has a
highpass nature which further attenuates low frequency components. Hence, we can expect the noise contribution from the
,
, and
sources to
to be negligible compared to
and
injections into
, reducing the effective number
of noise sources in this circuit to only
.
SIT AND SARPESHKAR: LOW-POWER BLOCKING-CAPACITOR-FREE CHARGE-BALANCED ELECTRODE-STIMULATOR CHIP
Fig. 11. Bode plot of G(s)==(1=H(s)) shaded in gray, showing the transfer function from Ibs and similar noise sources to Vp. The gain of g
as A and the dominant time constant of (Cc + Cpo)A r is defined as .
179
1
A r is defined
Estimating the noise from only two effective noise sources
and referring it back to
, we get
where
Fig. 12. Die photo of stimulator chip.
pF
mV
pF
At the highest current level, using a maximum
of 1 mA/V,
this sampled noise is converted to 75 nA in current noise. With
a pulsewidth of 32
, the standard deviation in charge
is then 2.4 pC, which is very small compared to the full-scale
charge of 30 nC. This circuit is, therefore, a very low noise
circuit, which is borne out by measurements presented in the
next section.
V. MEASUREMENTS
A 2.3 mm 2.3 mm stimulator chip was fabricated by the
Europractice IC fabrication service in an AMI 0.7- m 30 V
(I2T30E) process. The die photo is shown in Fig. 12.
A. Sample Output Waveforms
The output Vo was loaded with a discrete resistor and a
low-leakage 10 nF Teflon capacitor in series, to simulate a typ) were used to
ical electrode impedance. Seven digital bits (
set the value of
via a 7-bit DAC on chip, and the value of
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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS, VOL. 1, NO. 3, SEPTEMBER 2007
Fig. 14. Residual charge error over the range of DAC currents.
Note that the residual charge error captures the net effect of
any inaccuracies in the positive current pulse causing a mismatch with the negative current pulse in magnitude, duration
or shape. Ultimately, charge balance rather than current balance
determines the safety of current drivers in an implant.
Fig. 13. Output voltage Vo measured with various resistive loads R.
was latched on the falling edge of
. The size of the
LSB could be varied with an external pin control and was nomcorresponds to a 1 mA
inally set to 8 A such that
output. Sample output voltage waveforms from the biphasic current pulse are shown in Fig. 13.
The output voltage compliance exceeds 11 V, but is no greater
than 13 V on 6 and 9 V rails, due to 1 V of headroom
being sacrificed in active cascodes on both the pMOS and nMOS
current drivers. This headroom reduction could be alleviated by
using a differential amplifier to drive the active cascode, rather
than a common-source amplifier, but at the cost of additional
power.
Note that the asymmetry of the output waveforms in Fig. 13
allows us to save power by also making the voltage rails asymmetric. The larger negative-going pulse may result in greater
charge leakage (through ) during the negative phase, and may
also reduce the negative current output by channel-length modulation more than the positive-going pulse reduces the positive
current output. A result of this asymmetry is that the final charge
error then tends to be positive.
C. Verification of Shorting From Capacitor Voltage
Measurements
To verify that the shorting phase was working as designed, the
exponential time constant of discharge during shorting was
measured from scope captures of the capacitor voltage waveform, shown in Fig. 15.
The best-fit time constant of discharge was measured to be
around 40 s. The resistance was 2.2 k for these measurements, confirming that the on-resistance of the shorting tran2 k . If shorting was not performed, the charge
sistor was
error was observed to accumulate over many cycles and raised
the average voltage on the capacitor above ground. Shorting was
thus confirmed to be essential in maintaining precise charge balance.
D. Settling Time Measurements
were made
Measurements of the settling behavior of
to confirm the loop bandwidth and stability predictions of
Section IV-A and B, respectively, and shown in Fig. 16.
Settling was well-behaved over all input levels, showing a
first-order response and reaching final value well within the allotted time of 16 s. A slight bumpiness in the waveforms is
caused by clock switching injection, which is likely the major
cause of the final imbalance in current.
B. Residual Charge Error
E. Noise Measurements
The precision of the dynamic current balance was obtained by
measuring the residual voltage integrated on the 10 nF Teflon
capacitor at the end of the biphasic pulse. A plot of this error
over the range of DAC currents is shown in Fig. 14.
The decrease in error at high currents may be caused by velocity saturation effects in device
, as suggested by simulation, which reduces the transconductance and hence the conversion gain of voltage error on
into output-current error. The
worst-case charge error of 0.12 nC out of a full scale charge of
30 nC thus accounts for a mismatch of 0.4% in our scheme.
To verify the level of circuit noise, multiple readings of the
final capacitor voltage were taken, to compute the variance in
the residual voltage, shown in Fig. 17.
The standard deviation of 0.8 mV translates to 8 pC of
charge on the 10 nF capacitor. The measured noise is slightly
higher than the calculated 2.4 pC in Section IV-C, but is still
very small, on the order of 0.025% of the full scale of 30 nC.
The additional noise likely arises from sources like timing jitter
in the pulsewidth which we have neglected in our calculations.
Fortunately, it is small enough to not be of concern.
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181
Fig. 15. Capacitor voltage scope captures at the end of a biphasic pulse, which allow measurements of residual voltage and to be made.
The power consumption in this work has not been optimized
because we have exceeded the settling time specification of
16 s by a significant margin. We could therefore reduce the
) and source-follower ( )
power in the analog storage (
to achieve perhaps as much as a 50% reduction in the 40 W
of unshared power.
G. Average DC Current Error
Fig. 16. Settling time measurements for Vp over various input levels.
F. Power Consumption
Most of the biasing power (e.g., in the voltage reference and
active cascodes) can be shared between multiple channels in a
multichannel implant. If we amortize this biasing power by 16
channels in a 16-channel implant, the dominant source of power
consumption then lies in Ibs and the analog storage transconductor,
, which cannot be shared. The current
and each
leg of the
transconductor was biased with
1.2 A,
which on 6 and 9 V rails works out to consume 40 W. An
additional 7 W of shared biasing power then yields a total of
47 W per channel.
Dynamic current balancing as we have implemented on its
own achieves 0.4% mismatch by yielding 120 pC imbalance out
of the 30 nC full scale charge. By following dynamic current
balancing with shorting, an additional reduction in the charge
error can be achieved as a function of the electrode impedance
and minimum inter-pulse interval. Assuming a maximum eleck
nF
s (including
trode time constant of
2 k of on-resistance in the shorting switch) and a minimum interval of 1 ms between pulses, the electrode can be discharged
for at least three time constants or achieve a charge reduction
of at least
. The net dc current error is then the
charge error divided by the inter-pulse interval of 1 ms, i.e.,
pC
ms
nA. This level of error puts us well below
the safety limit in neural implants of 25 nA, and starts to approach the level of safety afforded by dc blocking capacitors.
With further refinements in our dynamic current balancer, especially in the area of rejecting charge injection from the clock
and other switching transients, 0.1% mismatch at the same full-
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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS, VOL. 1, NO. 3, SEPTEMBER 2007
Fig. 17. Multiple (N = 1600) scope acquisitions are used to compute the mean () and standard deviation ( ) of the residual error voltage, taken at 80.8 s.
This measurement was taken with Din = 64, which exhibited the worst-case error of 12 mV. The end of the biphasic pulse occurs at 80.5 s, and shorting
begins at
81 s.
TABLE I
PERFORMANCE COMPARISON OF BLOCKING-CAPACITOR-FREE ELECTRODE STIMULATOR DESIGNS
scale current of 1 mA should be achievable, suggesting that future improvements may result in a 1.5 nA error.
VI. CONCLUSION
This paper shows that the two-step technique of cascading
dynamic current balancing followed by shorting can achieve a
level of dc current error of 6 nA and a residual voltage error
of 12 mV, comparable to dc blocking capacitor performance.
Table I summarizes our electrode-stimulator performance relative to other designs.
Our results suggest that a future generation of neural implants
can implement charge balancing in active electronics and remove the need for dc blocking capacitors, thus removing a big
bottleneck in reducing their size.
ACKNOWLEDGMENT
The authors would like to thank M. O’Halloran for helpful
discussions regarding the design and use of the low-leakage
analog storage cell.
SIT AND SARPESHKAR: LOW-POWER BLOCKING-CAPACITOR-FREE CHARGE-BALANCED ELECTRODE-STIMULATOR CHIP
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Ji-Jon Sit received the B.S. degree in electrical engineering and computer science at Yale University,
New Haven, CT, in 2000, and the M.S. degree in
electrical engineering from the Massachussetts Institute of Technology, Cambridge, in 2002, where he
received the Ph.D. degree in electrical engineering,
in the Analog VLSI and Biological Systems Group
in 2007.
He is now with Advanced Bionics Corporation,
Sylmar, CA, and works on emerging cochlear
implant technology.
Rahul Sarpeshkar received the B.S. degrees in electrical engineering and physics at the Massachussetts
Institute of Technology (MIT), Cambridge, and the
Ph.D. degree from California Institute of Technology,
Pasadena.
After completing the Ph.D., he joined Bell Labs
as a member of technical staff in the Department of
Biological Computation within its Physics division.
Since 1999, he has been on the faculty of MIT’s
Electrical Engineering and Computer Science Department where he heads a research group on Analog
VLSI and Biological Systems. He holds over twenty patents and has authored
more than 70 publications including one that was featured on the cover of
Nature. He has given over 100 invited lectures. His research interests include
analog and mixed-signal VLSI, biomedical systems, ultra low power circuits
and systems, biologically inspired circuits and systems, molecular biology,
neuroscience, and control theory.
Prof. Sarpeshkar has received several awards including the Packard Fellow
award given to outstanding young faculty, the ONR Young Investigator Award,
the NSF Career Award, and the Indus Technovator Award. He is currently an
Associate Editor of the IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND
SYSTEMS. He has received the Junior Bose award and the Ruth and Joel Spira
award for excellence in teaching at MIT.