Influence of the properties of magnetic materials
on the size and performance of PLC couplers.
Johannes Binkofski
VACUUMSCHMELZE GmbH & Co. KG
Grüner Weg 37
D-63450 Hanau
Tel.: +49-6181-38-2664
Fax: +49-6181-38-82664
E-mail: Johannes.Binkofski@vacuumschmelze.com
Abstract – Transformers with magnetic core are commonly
used in PLC systems for both capacitive style and inductive style
couplers.
The choice of the magnetic material for such cores greatly
influences the geometric dimensions and the ultimate
performance of the inductive components used in PLC couplers.
VITROPERM, with its high saturation flux density and high
permeability, is a very good choice of core material for such
coupler applications.
I. INTRODUCTION
Soft magnetic materials are commonly described in terms of
a hysteresis loop (Fig. 1). These loops, measured on toroidal
cores approximately 10 mm o.d. and no air-gap, show very
clearly the differences between popular traditional ferrites and
modern nanocrystalline magnetic materials like VITROPERM.
Ferrites are dark grey or black ceramic-like materials [2] with
a typical saturation flux density, BS, about 0.38 T for high
permeability materials (µ ≈ 10,000). For lower permeability
ferrites the saturation flux density can be up to 0.545 T.
Modern nanocrystalline magnetic materials, like VITROPERM, [1], are metallic materials with a typical saturation
flux density, BS, about 1.2 T. The permeability, µ, of VITROPERM can be tightly controlled within a wide range of values
between 20,000 and 100,000. These two magnetic parameters
(saturation flux density and permeability) have an essential
influence on the size and performance of PLC couplers.
II. CAPACITIVE COUPLING
Capacitive coupling mainly consists of a transformer
winding and a capacitor connected in series (Fig. 2). A more
detailed description can be found in [3]. The capacitor
greatly restricts, (but cannot totally eliminate) the flow of the
50/60 Hz mains current in the windings of the transformer.
However, this capacitor readily couples the higher frequency
PLC signals between the transformer windings and the mains
conductors.
Mains
C
PLCTransceiver
1.4
1.2
1.0
I
L
Vitroperm
0.8
Flux density, B (T)
0.6
0.4
0.2
Fig.2. Principle circuit of the capacitive coupling
Ferrite
0.0
Due to the high line-to-line voltages e.g. 230-400 VRMS in low
voltage systems, the mains current conducted through the
capacitor can still achieve considerable values, depending on
the application.
-0.2
-0.4
-0.6
-0.8
-1.0
A. Lower frequency applications
-1.2
-1.4
-80 -70 -60 -50 -40 -30 -20 -10
0
10 20
30 40
50 60
70 80
Applied field, H (A/m)
Fig. 1. Hysteresis loops of magnetic materials
For instance, in AMR (automatic meter reading) and home
automation applications in Europe so-called CENELEC
frequency bands A, B, C and D can be used. In these
frequency bands from 9 kHz to 148.5 kHz the mains current,
limited by the capacitor, can reach some tens of mA. Typical
values for the capacitance C of such capacitors vary from
100 nF to 470 nF. Typical value for the inductance L of the
transformer is 1 mH.
1.4
1.3
1.2
Vitroperm
1.1
Inductance, L (mH)
1.0
0.9
In addition a high level of such harmonics can be the cause of
EMC problems and degraded range of PLC transmission.
The standards and regulations establish restrictions for the
amplitude relationship (in dB) between the PLC signal and its
2nd and 3rd harmonics. Fig. 5 illustrates the amplitude interval
between a 132.5 kHz PLC signal and its 2nd harmonic for two
different PLC transformers. Both transformers use similar
size toroid core, but different magnetic core materials. Note
the degradation of the 2nd harmonic amplitude relationship
with an increase of 50 Hz mains current.
0.8
0.7
90
Vitroperm
Ferrite
0.6
80
0.5
0.4
Limit
70
0.2
0.1
0.0
0
5
10
15
20
25
30
35
40
45
50
55
60
Alternating current (50 Hz), IRMS (mA)
Fig. 3. Inductance versus mains current
The mains current flowing through the windings of the PLC
transformer can drive the magnetic core into saturation.
As shown in Fig. 3, the inductance L of the transformer will
decrease significantly due to saturation of the magnetic core.
As a consequence of this saturation, harmonics of the PLC
signals, exceeding the allowed CENELEC frequency band
will be produced. Fig. 4 shows a typical PLC signal with the
frequency of 132.5 kHz along with its 2nd and 3rd harmonics.
These harmonics exceed the allowed frequency band of 148.5
kHz.
2nd harmonic, ∆K2 (dB)
0.3
60
Ferrite
50
40
30
20
10
0
0
5
10
15
20
25
30
35
40
Alternating current (50 Hz), IRMS (mA)
Fig. 5. Relative amplitude of the 2nd harmonic K2 (265 kHz)
of the PLC signal (132.5 kHz)
In order to meet the 2nd harmonic requirement at higher mains
currents, the ferrite core based transformer of Fig. 5 should
have considerably greater dimensions than the comparable
nanocrystalline core based transformer.
Another way to reduce the harmonics is to use ferrite shapecores with an air-gap spacer between the core-halves. This
approach leads, however, to a significantly higher leakage
inductance of the transformer [4] and [5], which will
negatively influence the insertion loss of the coupling circuit.
B. Higher frequency applications
Fig. 4. Spectrum of the PLC signal due to the saturation
of the transformer
In applications like access technology or home networking,
PLC signals, in higher frequency range (about 1-30 MHz), are
used. In these applications the capacitors used in the coupling
circuits can have much lower capacitances (about 5 nF is
typical). As a consequence, the mains current flowing through
the winding of the transformer has a relatively small value. In
this case the saturation of the transformer core is not the main
issue.
More important is the behaviour of the transformer at higher
frequencies described in terms of insertion loss [4]. This
behaviour is enhanced by smaller leakage inductance [5], and
reduced number of turns. This improvement can be
accomplished using modern magnetic materials (such as
nanocrystalline or amorphous) with higher permeability, µ
(see Fig. 1).
III. INDUCTIVE COUPLING
In inductive coupling systems, [3], the mains current can
flow, unlimited, through the magnetic core clamped over the
mains wire.
The photo of Fig. 6 illustrates the principle of such an
inductive coupling and some companies are making very
successful technical solutions based on this principle.
Magnetic cores used for such systems are so-called cut-cores
with an air-gap, δ. Fig. 7 shows the side view of such a cutcore with the dimensions da (outer diameter), di (inner
diameter), h (height) and δ (air-gap).
The air-gaps, δ, have to be considered on both sides of the
cut-core.
Using these dimensions, simplified formulas for the cross
section AFe, the magnetic path length lFe of the core and VFe
the volume of the magnetic material in the core can be
derived:
AFe = ηFe ⋅
lFe = π ⋅
( da − di )
⋅h
2
(da + di )
2
VFe = AFe · lFe
where ηFe is the filling factor of the magnetic material.
Typical values are 0.8 for metallic alloys and 1 for ferrites.
The effective permeability, µe, of the core decreases
considerably due to the air-gap and can be calculated as
follows:
µe =
1
2 ⋅δ
+
µ
lFe
1
One can readily see that effective permeability µe is
determined (especially for high permeability materials)
mainly by the ratio δ/lFe i.e. the difference between µe of
different magnetic materials is very small.
Fig. 6. Principle of inductive coupling
In such applications, the high values of the mains current are
critical in terms of saturation of the magnetic core. In medium
voltage systems, in which such inductive couplers are
commonly used, the geometric dimensions of the magnetic
core may need to be quite large to avoid the risk of saturation.
The magnetic energy W in the core due to the flowing mains
current I and required inductance L (typical value of 10 µH)
can be expressed by:
B
∫
W = VFe ⋅ H ⋅ dB
0
This expression for W can be developed to:
δ
di
da
W=
1
1
B2
⋅ L ⋅ I 2 = ⋅ VFe ⋅
2
2
µe ⋅ µ 0
from which the required volume of the magnetic material VFe
can be calculated:
h
Fig.7. Toroidal cut-core
VFe = L ⋅ I 2 ⋅ µe ⋅ µ 0 ⋅
1
B2
As the volume of the required magnetic material depends on
the square of the flux density B, the use of magnetic materials
with high saturation flux density BS can help considerably to
reduce the size of the core required for inductive couplers.
For instance, modern nanocrystalline magnetic materials like
VITROPERM with a very high saturation flux density
BS = 1.2 T (compared to BS = 0.4 T for typical ferrite
magnetic materials), allow manufacturers to build inductive
couplers, which require magnetic cores up to 9 times smaller
than similar cores made of ferrites.
IV. CONCLUSIONS
In all these situations, where the saturation of the
transformer core is a crucial issue, a diligent choice of the
magnetic material will reduce the size and the costs of the
PLC coupler.
The most important parameters of magnetic materials used for
PLC couplers are saturation induction, BS, and permeability,
µ, with both factors indicating the ability of the magnetic
material to conduct the magnetic flux.
Especially the saturation induction, BS, determines directly
the volume of the magnetic material needed for inductive
components in PLC couplers. Commonly used ferrite
materials have a relatively low saturation induction compared
with modern nanocrystalline alloys. In practice, selecting
such nanocrystalline alloys can help to reduce the size of the
inductive component by factor of approximately 4-8.
This considerable reduction of the size is accompanied by
better insertion loss at high frequencies, due to the higher
permeability of nanocrystalline alloys.
REFERENCES
[1] J. Petzold, “Applications of nanocrystalline softmagnetic
cores in modern electronics,” Proc. of 16th conference
about soft magnetic materials, Düsseldorf, Germany,
pp. 97-106, September 9-12, 2003.
[2] Ferroxcube, “Soft ferrites and accessories,”
data handbook 2005, September 1, 2004.
[3] O. Bilal, E. Liu, Y. Gao and T. Korhonen, “Design of
broadband coupling circuits for powerline
communication” Proc. of 8th ISPLC-2004, Zaragoza,
Spain, pp. 128-131, March 31-April 2, 2004.
[4] P.A Janse van Rensburg and H.C. Ferreira, “Step-by-step
design of a coupling circuit with bi-directional
transmission capabilities,” Proc. of 8th ISPLC-2004,
Zaragoza, Spain, pp. 238-243, March 31-April 2, 2004.
[5] P.A Janse van Rensburg and H.C. Ferreira, “The role of
magnetizing and leakage inductance in transformer
coupling circuitry,” Proc. of 8th ISPLC-2004,
Zaragoza, Spain, pp. 244-249, March 31-April 2, 2004.