Basic Counting Systems
PULSE ELECTRONICS
The nuclear electronics industry has standardized the signal
definitions, power supply voltages and physical dimensions of basic
nuclear instrumentation modules using the Nuclear Instrumentation
Methods (NIM) standard initiated in the 1960s. This standardization
provides users with the ability to interchange modules, and the
flexibility to reconfigure or expand nuclear counting systems,
as their counting applications change or grow. CANBERRA is a
leading supplier of Nuclear Instrumentation Modules (also called
NIM). In the past several years, the industry trend has been to offer
modular detector electronics with the multichannel analyzer (MCA)
and all supporting instrumentation for spectroscopy with a single
detector combined in a compact, stand-alone enclosure. These
modular MCAs are smaller, lighter and use less power than the
NIM-based counting systems that preceded them. However, their
performance is equal to, or greater than, comparable NIM-based
systems. CANBERRA is also a leading supplier of these modular
detector electronics. Depending on the application and budget, NIM
or modular electronics may be the best counting equipment solution
for the user, and CANBERRA supports both of these form factors
with a wide variety of products.
Basic electronic principals, components and configurations which
are fundamental in solving common nuclear applications are
discussed below.
PREAMPLIFIERS AND AMPLIFIERS
Most detectors can be represented as a capacitor into which a
charge is deposited, (as shown in Figure 1.12). By applying detector
bias, an electric field is created which causes the charge carriers
to migrate and be collected. During the charge collection a small
current flows, and the voltage drop across the bias resistor is the
pulse voltage.
The preamplifier is isolated from the high voltage by a capacitor. The
rise time of the preamplifier’s output pulse is related to the collection
time of the charge, while the decay time of the preamplifier’s output
pulse is the RC time constant characteristic of the preamplifier itself.
Rise times range from a few nanoseconds to a few microseconds,
while decay times are usually set at about 50 microseconds.
Charge-sensitive preamplifiers are commonly used for most solid
state detectors. In charge-sensitive preamplifiers, an output voltage
pulse is produced that is proportional to the input charge. The output
voltage is essentially independent of detector capacitance, which is
especially important in silicon charged particle detection (i.e. PIPS®
detectors), since the detector capacitance depends strongly upon
the bias voltage. However, noise is also affected by the capacitance.
To maximize performance, the preamplifier should be located at
the detector to reduce capacitance of the leads, which can degrade
the rise time as well as lower the effective signal size. Additionally,
the preamplifier also serves to provide a match between the high
impedance of the detector and the low impedance of coaxial cables
to the amplifier, which may be located at great distances from the
preamplifier.
The amplifier serves to shape the pulse as well as further amplify it.
The long delay time of the preamplifier pulse may not be returned
to zero voltage before another pulse occurs, so it is important to
shorten it and only preserve the detector information in the pulse
rise time. The RC clipping technique can be used in which the pulse
is differentiated to remove the slowly varying decay time, and then
integrated somewhat to reduce the noise. The unipolar pulse that
results is much shorter. The actual circuitry used is an active filter
for selected frequencies. A near-Gaussian pulse shape is produced,
yielding optimum signal-to-noise characteristics and count rate
performance.
Figure 1.12 Basic Detector and Amplification
Figure 1.13 Standard Pulse Waveforms
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A second differentiation produces a bipolar pulse. This bipolar pulse
has the advantage of nearly equal amounts of positive and negative
area, so that the net voltage is zero. When a bipolar pulse passes
from one stage of a circuit to another through a capacitor, no charge
is left on the capacitor between pulses. With a unipolar pulse, the
charge must leak off through associated resistance, or be reset to
zero with a baseline restorer.
PULSE HEIGHT ANALYSIS AND COUNTING TECHNIQUES
Pulse Height Analysis may consist of a simple discriminator that
can be set above noise level and which produces a standard logic
pulse (see Figure 1.13) for use in a pulse counter or as gating
signal. However, most data consists of a range of pulse heights of
which only a small portion is of interest. One can employ either of
the following:
High performance gamma spectrometers are often designed today
using Digital Signal Processing (DSP) techniques rather than analog
shaping amplifiers. The shaping functions are then performed in the
digital domain rather than with analog circuitry. This is discussed
later in this section.
1.Single Channel Analyzer and Counter
2.Multichannel Analyzer
Typical preamplifier and amplifier pulses are shown in Figure 1.13.
The dashed line in the unipolar pulse indicates undershoot which can
occur when, at medium to high count rates, a substantial amount of
the amplifier’s output pulses begin to ride on the undershoot of the
previous pulse. If left uncorrected, the measured pulse amplitudes
for these pulses would be too low, and when added to pulses whose
amplitudes are correct, would lead to spectrum broadening of
peaks in acquired spectra. To compensate for this effect, pole/zero
cancellation quickly returns the pulse to the zero baseline voltage.
The bipolar pulse has the further advantage over unipolar in that
the zero crossing point is nearly independent of time (relative to
the start of the pulse) for a wide range of amplitudes. This is very
useful in timing applications such as the ones discussed below.
However, the unipolar pulse has lower noise, and constant fraction
discriminators have been developed for timing with unipolar pulses.
For further discussions on preamplifier and amplifier characteristics,
please refer to each applicable module’s subsection.
The single channel analyzer (SCA) has a lower and an upper level
discriminator, and produces an output logic pulse whenever an
input pulse falls between the discriminator levels. With this device,
all voltage pulses in a specific range can be selected and counted.
If additional voltage ranges are of interest, additional SCAs and
counters (i.e. scalers) can be added as required, but the expense
and complexity of multiple SCAs and counters usually make this
configuration impractical.
If a full voltage (i.e. energy) spectrum is desired, the SCA’s
discriminators can be set to a narrow range (i.e. window) and then
stepped through a range of voltages. If the counts are recorded
and plotted for each step, an energy spectrum will result. In a
typical example of the use of the Model 2030 SCA, the lower level
discriminator (LLD) and window can be manually or externally (for
instance, by a computer) incremented, and the counts recorded for
each step. However, the preferred method of collecting a full energy
spectrum is with a multichannel analyzer.
The multichannel analyzer (MCA), which can be considered as
a series of SCAs with incrementing narrow windows, basically
consists of an analog-to-digital converter (ADC), control logic,
memory and display. The multichannel analyzer collects pulses in
all voltage ranges at once and displays this information in real time,
providing a major improvement over SCA spectrum analysis.
Figure 1.14 Multichannel Analyzer Components with Analog Signal Processing
Figure 1.15
Multichannel Analyzer Components with Digital Signal Processing
Figure 1.14 illustrates a typical MCA block diagram. An input
energy pulse is checked to see if it is within the selected SCA
range, and then passed to the ADC. The ADC converts the pulse
to a number proportional to the energy of the event. This number
is taken to be the address of a memory location, and one count is
added to the contents of that memory location. After collecting data
for some period of time, the memory contains a list of numbers
corresponding to the number of pulses at each discrete voltage.
The memory is accessed by a host computer which is responsible
for spectrum display and analysis as well as control of the MCA.
Depending on the specific model MCA, the host computer may be
either a dedicated, embedded processor or a standard off-the-shelf
computer.
PULSE HEIGHT ANALYSIS WITH DIGITAL SIGNAL
PROCESSORS
Today’s high performance Multichannel Analyzer systems are
designed using Digital Signal Processing (DSP) techniques rather
than the traditional analog methods. DSP filters and processes the
signals using high speed digital calculations rather than manipulation
of the time varying voltage signals in the analog domain. The
preamplifier signal first passes through an analog differentiator,
then is delivered to a high speed digitizing ADC (Figure 1.15). The
output of the ADC is a series of digital values that represent the
differentiated pulse. Those signals are then filtered using highspeed digital calculations within the Digital Signal Processor.
For optimal speed and accuracy in signal processing, a trapezoidal
filter algorithm is deployed in the DSP implementation. Trapezoidal
filtering has been shown to allow for the highest throughput
performance with the least degradation of spectral resolution.
Additionally, the DSP based design is intrinsically more stable,
resulting in better performance over a range of environmental
conditions.
COUNTERS AND RATEMETERS
Counters and ratemeters are used to record the number of logic
pulses, either on an individual basis as in a counter, or as an average
count rate as in a ratemeter. Counters and ratemeters are built with
very high count rate capabilities so that dead times are minimized.
Counters are usually used in combination with a timer (either builtin, or external), so that the number of pulses per unit of time are
recorded. Ratemeters feature a built-in timer, so that the count rate
per unit of time is automatically displayed. Whereas counters have
an LED or LCD for the number of logic pulses, ratemeters have a
mechanical meter for real-time display of the count rate. Typically,
most counters are designed with 8-decade count capacity and offer
an optional external control/output interface, while ratemeters are
designed with linear or log count rate scales, recorder outputs and
optional alarm level presets/outputs. Additional information may be
found in the Counters and Ratemeters Introduction.
SIMPLE COUNTING SYSTEMS
As related above, pulse height analysis can consist of a simple
single channel analyzer and counter, or a multichannel analyzer.
Generally, low resolution/high efficiency detectors (such as
proportional counters and NaI(Tl) detectors) are used in X ray or
low-energy gamma ray applications where only a few peaks occur.
An example of a simple NaI(Tl) detector-based counting system of
this type is illustrated in Figure 1.16.
2015B
Figure 1.16 NaI Detector and Counter/Timer
with Alarm Ratemeter
802-3x3
Osprey
USB
Osprey MCA with Genie 2000 software
Figure 1.17 NaI Detector and MCA Configuration
Figure 1.18 HPGe Detector and Analog MCA Configuration
In this configuration, a Model 2015B Amplifier/SCA is used to
generate a logic pulse for every amplified (detector) pulse that falls
within the SCA’s “energy window”. The logic pulse is then used as
an input to the Model 512 Counter/Timer which provides the user
with a choice of either preset time or preset count operation. The
Model 512 is equipped with an RS-232 interface, which enables
it to be controlled and read out to a computer for data storage or
further analysis.
Alternatively, Model 1481LA Linear/Log Ratemeter is used as the
counter, with an alarm relay that will trigger if the count rate exceeds
a user preset value.
Although counters are still used in some applications, most of
today’s counting systems include a multichannel analyzer (MCA).
Besides being more cost effective than multiple SCA-based
systems, a MCA-based system can provide complete pulse height
analysis such that all nuclides, (i.e., even those not expected), can
be easily viewed and/or analyzed.
NaI(Tl) DETECTORS AND MULTICHANNEL ANALYZERS
The need for a single-input Pulse Height Analysis system for
use with a Sodium Iodide detector is served most simply by a
photomultiplier tube (PMT) base MCA such as the Osprey (Figure
1.17). The Osprey MCA includes a high voltage power supply,
preamplifier, and DSP electronics in addition to its MCA functions,
and thus, there is no need for any NIM modules or a NIM Bin. All of
this capability is provided in an enclosure no larger than a standard
tube base preamplifier, and the computer interface is via a USB or
Ethernet port on the host computer or a network hub.
GERMANIUM DETECTORS AND MULTICHANNEL
ANALYZERS
A typical analog HPGe detector-based gamma spectroscopy
system consists of a HPGe detector, high voltage power supply,
preamplifier (which is usually sold as part of the detector), amplifier,
ADC and multichannel analyzer. As will be discussed in more detail
later, DSP configurations replace the amplifier and ADC with digital
signal processing electronics.
The analog system components are available in several different
types, allowing the system to be tailored to the precise needs of
the application and the available budget. For example, low-end
amplifiers such as the Model 2022 offer basic capabilities, but users
with higher count rate or resolution requirements may consider the
Model 2026 or 2025 with Pileup Rejection/Live Time Correction
(PUR/LTC) feature and both Gaussian and triangular shaping.
Similarly, the ADC chosen for a system including a 556B NIM MCA
could be either an economical Wilkinson ADC like the Model 8701
or a faster Fixed Dead Time (FDT) ADC like Model and 8715.
For more information about selecting specific modules, refer to the
introduction sections for those specific components.
HPGe
Detector
IN
HV
2002
Preamp
H.V. Inh.
Lynx
DSA-10008701
ADC
InSpector 2000
USB
Preamp
Power
Genie 2000 software
Figure 1.19 HPGe Detector with USB-connected Digital Signal Processors
HPGe
Detector
IN
HV
2002
Preamp
H.V. Inh.
Lynx
8701
ADC
Ethernet
Preamp
Power
Genie 2000 software
Figure 1.20 HPGe Detector with DSA-2000 Digital Signal Processor (DSP)
For applications requiring security of the signal processing,
CANBERRA offers a variety of computer controlled electronics
which require access via a host computer, rather than unprotected
front panel for adjustment. For example, the AIM/ICB NIM family
is a network based, computer controlled signal processing line
that can be controlled remotely by a Genie 2000 or Apex-Gamma
spectroscopy workstation.
Spectroscopy systems based on Digital Signal Processing (DSP)
have been widely accepted as the state of the art. In a DSP based
system, the amplifier and ADC are replaced by a set of digital
circuits which implement the filtering functions in high speed digital
calculations. CANBERRA offers several DSP based products, all
of which offer superior environmental stability, higher count rate
throughput performance and better resolution over a range of count
rate conditions. Lynx, DSA-1000, Osprey and the InSpector 2000
all employ this advanced DSP technology.
Figures 1.18, 1.19 and 1.20 show several of the available
Germanium Detectors/MCA configurations. Optional LN2 Monitors,
Level Alarms, and Control Systems are available for most types of
detectors.
LEGe AND Si(Li) DETECTORS WITH MULTICHANNEL
ANALYZERS
Low Energy Germanium (LEGe) and Si(Li) detectors require
special circuitry to provide the long time constants required in the
amplifier to achieve maximum resolution, and to properly handle the
reset signals of their preamplifiers. Although several CANBERRA
amplifiers are suitable, the best resolution for analog MCA systems
will be obtained with the Model 2025 AFT Research Amplifier.
Besides allowing the user to select a long shaping time constant,
the Model 2025 features an enhanced baseline restorer which is
ideal for reset preamplifiers. Any of the CANBERRA Digital Signal
Processing MCAs or components can be used with these detectors
and provide even better throughput and resolution performance.
MULTIPLE INPUT SYSTEMS
CANBERRA offers two solutions for multiple input counting systems
which process the amplified signals from a number of detectors. A
multiple input scenario would typically be considered six or more
detector inputs – or the point at which multiple independent MCA
systems become cost prohibitive for a given counting application.
The Multiport II (Figure 1.18) is the first solution and, also, the
more robust of the two. It offers the capability for up to six totally
independent MCAs and ADCs housed in one double-wide NIM.
Because the MCAs and ADCs are separate from each other, any
combination of detectors and channel number settings may be used
for each input.
Preamp
Power
8224
3102D
Ethernet
556B
2007P
8701
802-3x3
880777 (554) RPI
2022
8224 Mux and 556B MCA
with Genie 2000 software
Figure 1.21 Multiple Input NaI Detector System
The second solution employs a Model 8224 Multiplexer (or Mixer/
Router) to route the signals from multiple detectors to a single ADC
for digitizing and on to a 556B MCA for processing as shown in
Figure 1.21. Since this configuration shares the MCA and ADC
among the detectors, it has a lower cost per input than the Multiport
II – particularly for large numbers of detectors. However, the
Multiplexer configuration has a major drawback due to the single
ADC; the count rate of the individual detectors must be relatively
low to avoid excessive signal pileup. Additionally, a Multiplexer must
allocate the memory of the MCA to its various inputs (same
amount for each input), which decreases the number of channels
available for each individual detector. Within these constraints,
Multiplexers can be quite efficient for applications such as low-level
environmental alpha spectroscopy in which multiple low-intensity
inputs are collected in MCA memory segments of 512 channels or
less. Low-level gamma counting with NaI detectors, which typically
don’t need more than 1024 channels, is another application that can
make use of a Multiplexer. An example configuration is depicted in
Figure 1.21.
It should be noted that the Multiport II and the 8224 Multiplexer
do not include spectroscopy amplifiers or detector bias supplies.
These components must be supplied by other parts of the signal
chain. Also, these two solutions do not include the benefits of
Digital Signal Processing.
Advances in electronics technology have dramatically lowered the
cost of MCAs, so that today, it is frequently more effective to use
multiple complete MCA systems (or the Multiport II) in place of a
Multiplexer.
LOW LEVEL GAMMA RAY COUNTING
Large volume HPGe detectors have become dominant over other
detector types for low level gamma ray spectroscopy because
of their inherently good resolution and linearity. It is only in the
analysis of single radionuclides that NaI(Tl) detectors can compare
in sensitivity with HPGe detectors. Since the majority of all gamma
spectroscopy applications require the analysis of more complex,
multi-radionuclide samples, the following discussion will be limited
to the application of HPGe detectors to low level counting.
The sensitivity of a HPGe spectrometer system depends on
several factors, including detector efficiency, detector resolution,
background radiation, sample constituency, sample geometry and
counting time. The following paragraphs discuss the role these
factors play in low level gamma ray counting.
1.EFFICIENCY: Generally, the sensitivity of a HPGe system
will be in direct proportion to the detector efficiency. HPGe
detectors are almost universally specified for efficiency relative
to a 3 in. NaI(Tl) at 25 cm detector-to-source distance at
1.33 MeV, and from this benchmark one may roughly compute
the efficiency at lower energies. However, for the customer
who is counting weak samples with lower gamma energies,
for instance 100-800 keV, the following subtle considerations
to the detector design are important to system performance:
a.The detector should have an adequate diameter. This
assures that the efficiency at medium and low energies
will be high relative to the efficiency at 1.33 MeV, where it
is bought and paid for.
b.The detector-to-end-cap distance should be minimal – five
millimeters or less. The inverse square law is real and will
affect sensitivity.
c.The detector should be of closed end coaxial geometry, to
assure that the entire front face is active.
2.RESOLUTION: Generally, the superior resolution of a HPGe
detector is sufficient enough to avoid the problem of peak
convolution, (i.e., all peaks are separate and distinct). The
sensitivity of a system improves as the resolution improves
because higher resolution means that spectral line widths
are smaller, and fewer background counts are therefore
involved in calculating peak integrals.
Since the sensitivity is inversely related to the square root of
the background, that is,
Sensitivity =
1
√ Bkg
improvements in resolution will not improve sensitivity as
dramatically as increased efficiency.
3.BACKGROUND RADIATION AND SAMPLE CONSTITUENCY:
Interfering background in gamma spectra originates either
from within the sample being counted (Compton-produced)
or from the environment. If the sample being analyzed has a
high content of high-energy gamma emitting radioisotopes,
the Compton-produced background will easily outweigh the
environmental background. For extremely weak samples,
the environmental background becomes more significant.
Obviously, massive shielding will do little to improve system
sensitivity for low energy gamma rays in the presence of
relatively intense higher energy radiation. However, Comptonsuppression can be very effective in reducing this background.
4.SAMPLE GEOMETRY: An often overlooked aspect of HPGe
detector sensitivity is the sample geometry. For a given
sample size (and the sample size should be as a large as
practicable for maximum sensitivity), the sample should
be distributed so as to minimize the distance between the
sample volume and the detector itself.
GERMANIUM DETECTORS WITH INERT SHIELDS
There are many different types of shield designs that are available, and
because of the difficulty in determining the background contribution of
the materials used in a given shield, it is difficult to assign performance
levels to various types of shields. However, some criteria for shield
designs have evolved over the years, such as:
1.The shield should not be designed to contain unnecessary
components like the Dewar. It will only contribute to
increased background if it is within the walls of the shield, as
well as unnecessarily increase the shield’s size, weight and
cost.
2.The detector should be readily installed and removable from
the shield.
Figure 1.22 Detector located in center of chamber
without requirement for extended end-cap
Pity the person who has to move lead bricks (at 12 kg each)
to disengage a HPGe detector. A HPGe detector and shield
system should have a liquid nitrogen transfer system to avoid
removing the detector for the weekly filling.
3.Sample entry should be convenient to the operator.
4.The shield should accommodate a variety of sample sizes
and configurations.
The HPGe detector should be located in the center of
the shield so as to minimize scatter from the walls. In this
position, the shield must accommodate the largest sample
that is anticipated. Also, sample placement should be
accurately repeatable and easily verified by the operator.
The shield design that has all these features and is moderately
priced is the CANBERRA Model 747 Lead Shield illustrated in
Figures 1.22 and 1.23.
Figure 1.23 Model 747 Lead Shield
The performance of the shield using a CANBERRA HPGe detector
is given below:
Shield Specs:
Inside Dimensions
Wall Thickness
Material
28 cm dia. x 40.5 cm high
10 cm
Low Background Lead
HPGe Specs:
Relative Efficiency
Resolution
12%
1.95 keV FWHM at 1.33 MeV
0.90 keV FWHM at 1.22 keV
Background
Count:
Sensitivity:
2.25 counts/second in the 50 keV–2.7 MeV range
Table 1.4 lists the sensitivities of several single
radioisotopes, assuming a counting time of
50 000 seconds, a 50% error and a detector-topoint-source distance of 1 cm.
Table 1.4 Radioisotope vs. Sensitivity
RADIONUCLIDE
ENERGY
in keV
SENSITIVITY
in pC
57Co
139Ce
137Cs
60Co
122
165
662
133
2
3
6
10
LOW BACKGROUND CRYOSTATS
The design or configuration of the cryostat is another factor in
system performance. Some cryostat/shield designs do not prevent
streaming from the outside environment, nor do they provide selfshielding from their own relatively hot components. Through an
improper choice of material types and/or thicknesses, the cryostat
may appreciably contribute to the background. CANBERRA has
developed sources for select, low-background, materials, and has
invested in the design and fabrication of low-background cryostats.
HPGe COMPTON SUPPRESSION SPECTROMETER
When the ultimate in low level counting is required, a Compton
Suppression Spectrometer, in conjunction with an appropriate
low-background shield/cryostat design, is the answer. In this
configuration, the HPGe detector is surrounded by an active NaI(Tl)
or plastic scintillation guard detector (also known as an annulus
detector), with the electronics configured in an anticoincidence
counting mode. The Compton continuum, which is primarily caused
by gamma rays which sustain one or more inelastic collisions
and escape (i.e. scatter out of) the germanium detector material
without imparting their full energy, can lead to concealment of
low activity peaks. Since this is undesirable in low level counting
applications, a Compton Suppression Spectrometer can be used to
gate (i.e. turn off) data acquisition whenever one of the incompletely
absorbed photons escapes the germanium detector and is “seen”
by the annulus detector. When acquisition is complete, the resultant
spectrum contains only peaks attributed to gamma rays which have
imparted their full energy within the detector material.
It should be pointed out that some radioisotopes (those having
coincident gamma rays) such as 60Co, will not be analyzed properly
by the anticoincidence spectrum from a Compton Suppression
System. Therefore, two spectra are usually obtained from such a
spectrometer – one in the anticoincidence mode, and the other in
the normal (ungated) mode.
Figure 1.24 illustrates a typical example of a Compton Suppression
System.
One type of annulus has six (6) 5.08 cm (2-inch) diameter
photomultiplier tubes (PMTs) on one end, and a 7.62 cm (3-inch)
diameter NaI(Tl) plug with one PMT (which is operated in parallel
with the other PMT) on the other end. A simpler type of annulus
detector uses a 15.24 cm (6-inch) diameter NaI(Tl) well detector
on a single PMT. In either configuration, the annulus must be large
enough to allow the insertion of the HPGe detector’s endcap along
with the sample.
Figure 1.24 Compton Suppression System
While some endorse the use of a fairly complex Timing Chain
to derive the anti-Compton gate signal, CANBERRA has found
that the simplified circuit shown in Figure 1.24 yields equivalent
results.2 The “Incoming Count Rate” signals from the Spectroscopy
Amplifiers are checked for coincidence, and, if it exists, the 2040
Coincidence Analyzer’s output is used as an anti-coincidence input
to the ADC’s Gate. When coincidence occurs, this gate “turns
off” the delayed unipolar signal from the Spectroscopy Amplifier.
Typical Compton Suppression Spectrometer results are illustrated
in Figure 1.25. It can be seen that the ‘figure of merit’ – the value of
the 137Cs peak at 662 keV divided by the average contents of the
Compton continuum (the energy range 358-382 keV) – is on the
order of 1000:1.
Figure 1.25 Ge Spectra with Compton Suppression
HIGH COUNT RATE GAMMA RAY SYSTEMS
High count rate applications require special techniques to assure
good resolution and/or good throughput. In general, “high count
rate” is used to refer to incoming count rate, that is, the number
of events seen by the detector. The term “throughput rate” may be
of more interest to the researcher, being a measure of the rate at
which the system can accurately process these incoming counts.
In high count rate HPGe detector applications, problems such as
the loss of resolution, excessively long counting times, erroneous
peak to background ratios, inaccurate counting statistics or system
shutdown due to overload and saturation begin to appear. In some
experiments, the solution to these problems merely lies in reducing
the incoming count rate to the detector, or by employing electronics
which inhibit the processing of pulses through the electronics
when events are occurring so fast that they are overlapping (pulse
pileup). In this latter solution, system throughput will of course be
reduced, but parameters such as resolution will be enhanced. Table
1.5 indicates the throughput limitations of the major components of
a spectroscopy chain. Note that the term “energy rate limited” refers
to the fact that the component’s performance is not only affected
by the incoming count rate, but by the relative energy (amplitude)
of the incoming counts as well. Each element in the chain can be
optimized for high count rate performance.
2. Compton Suppression Made Easy, Application Note
Table 1.5
Major System Components and their Throughput Limitations
THE DETECTOR
For the detector itself, the charge collection time is the limiting
factor, and this parameter is a function of the detector geometry –
when a photon interaction takes place, charge carriers in the form
of holes and electrons are produced, and the time taken for these
carriers to be swept to the p and n electrodes of the detector is the
time for full energy collection. In a germanium detector, this time is
a function of detector size, as the charge carriers travel about 0.1
mm/ns. As the charge collection time increases, the Amplifier must
take a longer time to process the signal and develop its linear pulse,
or else not all of the incident energy will be reflected in that pulse
(“ballistic deficit”). Thus, larger detectors require longer amplifier
time constants, or more sophisticated peak shapes.
Some ways to address high count rate in the detector include
moving the detector farther away from the source, or collimating
the detector – which in both cases reduces the number of events
seen by the detector – or using a detector of lesser efficiency. The
detector in the latter case will ‘see’ fewer events, and furthermore
will have a lower charge collection time.
THE PREAMPLIFIER
Most Germanium detectors in use today are equipped with RCfeedback, charge sensitive preamplifiers. In the RC-feedback
preamplifier, a feedback resistor discharges the integrator, typically
in one or two milliseconds. If the incoming energy rate (count rate X
energy/count) produces a current that exceeds the capability of the
resistor to bleed it off, the output will increase until, in the extreme,
the preamplifier saturates and ceases to operate. This limit occurs
at approximately 200k MeV/s. The saturated condition remains
until the count rate is reduced. The saturation limit is dependent
on both energy and count rate and is usually specified in terms
of the “energy/rate limit”. The energy/rate limit can be increased
by lowering the value of the feedback resistor, but this in turn
allows more noise to pass through the preamplifier, resulting in a
degradation in resolution.
When a Coaxial Germanium detector is used in applications requiring
high throughput, the Model 2101 Transistor Reset Preamplifier
(TRP) is favored over traditional RC feedback Preamplifiers. The
higher cost of the TRP is justified by its much higher energy rate
capacity, an enhancement obtained by replacing the Feedback
Resistor of a typical RC feedback preamplifier with a special reset
circuit. This circuit monitors the dc level of the preamplifier and
discharges the feedback capacitor whenever the output reaches
a predetermined reset threshold. At moderate to high count rates
(i.e. above 20 000 cps), the absence of the feedback resistor and
its attendant noise and secondary time constant contributions lead
to: 1) lower preamplifier noise contributions, 2) inherently better
resolution and reduced spectrum broadening vs. count rate, 3)
elimination of the need for pole/zero cancellation, and 4) elimination
of ‘lock-up’ due to saturation. Figure 1.26 illustrates the throughout
performance of the two preamplifier styles.
Figure 1.27 Typical Amplifier Pulses
Figure 1.26 Throughput vs. Count Rate:
Throughput Optimization
Although the Model 2101 TRP virtually never shuts down due to
saturation, its reset process and the amplifier overload which it
causes does induce intervals of dead time into the counting system.
The Model 2101 has been designed with a small Charge Gain (50
mV/MeV) and a wide Dynamic Range (4 V) to significantly reduce
the dead time due to resets in comparison to competitive units.
DIGITAL SIGNAL PROCESSOR
As we described in an earlier section, Digital Signal Processors
(DSP) have come to replace the analog shaping amplifier and ADC
in most high performance gamma spectroscopy systems. It is in
applications involving high count rate performance where the
advantages of DSP become most pronounced.
In gamma spectroscopy systems, the DSP replaces the functionality
of both the shaping amplifier and the ADC. The DSP first filters the
signal for optimum signal to noise ratio and to provide gain. It then
detects the peak amplitude of the filtered pulse to calculate the
memory address of the MCA channel into which the event is to be
stored.
In the DSP, the analog signal from the preamplifier is first differentiated
in the analog domain to provide a rapid return to baseline. This is
depicted in Figure 1.27. The resulting time varying voltage signal is
sampled by a high speed sampling analog to digital converter. This
results in a digitized profile of the differentiated preamplifier signal
represented in internal memory of the DSP. From this point on, the
signal is processed in the digital domain by the DSP – essentially
a high speed digital computer executing calculations as opposed
to analog circuits manipulating time varying voltage signals.
Processing the signals digitally allows more sophisticated filtering
functions to be applied to the signal. It also allows greater flexibility
to the user in terms of adjusting filtering parameters – more
possible settings are available because they are handled as digital
commands, not the selection of discrete analog components.
Finally, the use of high speed digital electronics allows the signals
to be processed more rapidly, thus contributing further to the count
rate performance of the system.
CANBERRA’s DSP products deploy a trapezoidal filtering algorithm
as shown in Figure 1.28. Two parameters are available for user
adjustment – the rise/fall time of the trapezoid (hereafter referred to
as rise time) and the flat top time.
Figure 1.28 Trapezoidal Pulse
Waveform as processed in DSP
Adjusting the rise time changes the filter characteristics to optimize
for noise characteristics. The larger the rise time, the better the
signal to noise ratio. Shorter rise times will adversely affect signal
to noise ratio and degrade the resolution of the system. Flat top
adjustments are made to accommodate the variations in pulse rise
time which in turn is proportional to the charge collection time in
the detector. Larger detectors tend to have a larger number of long
rise time (large charger collection time) events, thus requiring a
longer flat top time. Failure to set the DSP rise time long enough to
accommodate the longest charge collection time events results in
degraded resolution, an effect known as ballistic deficit. Note that
for some types of smaller detectors, the flat top time can be set near
or very close to zero, resulting in a triangular shape.
These two parameters together control the total event processing
time. The total processing time for an event processed with the DSP
trapezoidal algorithm is defined by the equation:
Tp = (2Tr) + Tflat top
We see that the settings for both parameters effect the total
processing time, which in turn effects the count rate throughput
of the system. As we noted earlier, setting either parameter too
fast can result in lost resolution. Increasing the settings improve
resolution, but lengthen processing time and sacrifice throughput.
A tradeoff exists (as it did in analog systems) between count rate
throughput and resolution. Higher throughput can be attained with
a loss of resolution and better resolution can be attained at a loss
of throughput – up to the limits imposed by the performance of the
detector and preamplifier components.
These tradeoffs also existed in traditional analog systems, but the
tradeoffs can now be made at a higher level – the DSP provides
both improved throughput and improved resolution as compared
to analog. This is due to a number of factors. First, the trapezoidal
algorithm is simply more efficient and can process the signals more
accurately and rapidly than analog electronics.
Figure 1.29 A comparison of the system throughput as a function
of input count rate for a DSP and an analog system optimized for
high throughput for a small detector (11%)
Secondly, the user has much more flexibility to vary the components
of the processing time. In analog systems, the processing was
controlled by a single parameter – the shaping time. Now with DSP,
two parameters are available – one to accommodate noise level
and one to accommodate detector pulse rise time. By adjusting
these two separately, optimum settings can more readily be attained
resulting, generally, in shorter total processing time to reach the
same resolution result. Additionally, the analog amplifiers typically
were limited to six or fewer shaping time selections. If, say, 2 µs
shaping was too short, the next available selection was usually 4 µs
– twice the processing time. With the CANBERRA DSP products,
the user can typically select from 35 to 40 rise times and 21 flat
top times. Again, this greater granularity of adjustment makes it
possible to more closely optimize the performance.
Note that the CANBERRA DSP products also implement Pile
Up Rejection/Live Time Correction (PUR/LTC). Earlier products
implemented this feature with analog circuitry, but in the DSP this is
incorporated into the digital domain functions. Pulse pileup occurs
when a new pulse from the preamplifier reaches the input stages of
the DSP before the previous pulse is fully processed. In such cases,
the PUR/LTC function a) inhibits the processing of any invalid,
composite pulses and b) turns off the live time clock during the
time pulse processing is gated off. In this manner, piled up events –
which would serve only to distort the spectrum – are rejected before
storage by the MCA and the actual live counting time of the MCA
remains correct.
Figure 1.30 A comparison of the system resolution as a function
of input count rate for a DSP and an analog system optimized
for maximum throughput for a small detector (11%)
The improved performance of the DSP as compared to analog
systems is shown in Figures 1.29 to 1.34. Figures 1.29 and 1.30
show real performance data collected with a DSP and an analog
gated integrator and fast ADC (the fastest available using analog
technology). For this experiment, a Model 2060 DSP was set for rise
time of 0.72 µs and flat top time of 0.68 µs. The analog gated integrator
amplifier (Model 2024) was set for shaping of 0.25 µs and paired with a
800 ns Fixed Dead Time ADC. These settings were chosen
for optimal throughput with a relatively small (≈11% efficient)
germanium detector.
Figure 1.31 A comparison of the system throughput as a function
of input count rate for an analog system optimized for maximum
throughput with a DSP system set for a similar throughput
As we can see from Figure 1.29, the DSP based system provides
higher throughput by approximately 50%. Figure 1.30 shows the
resolution comparison for the same experiment and demonstrates
that the DSP also provides significantly better resolution once the
input count rate exceeds approximately 150 kcps. Note that the
shape of the resolution curve in Figure 1.30 is also much flatter,
indicating that widely varying count rates can be accommodated at
a relatively constant resolution.
Note that with these settings chosen for highest throughput, the
resolution performance at lower count rates is actually slightly
worse with the DSP. However, in an application involving those
count rates, it is unlikely those settings would be used. Figures 1.31
and 1.32 show the same analog data compared to the DSP system
with the rise time extended to 1.24 µs. This reduces the throughput
of the DSP system although it is still superior to that of the analog.
Further, we see now that with these settings, the resolution of the
DSP is superior to the analog across the full range of incoming
count rates.
Figures 1.33 and 1.34 compare a Model 2060 DSP to a Gaussian
analog system consisting of a Model 2025 amplifier and Model
8715 ADC. In this case, the settings of both systems were chosen
to provide optimal resolution under the high incoming count rates.
Analog systems were set for 2 µs and 4 µs Gaussian shaping times
while the DSP settings were 5.6 µs rise time and 0.8 µs flat top.
Figure 1.33 shows that, with these settings, the throughput of the
DSP system is approximately equal to that of the 2 µs Gaussian
system. Yet Figure 1.34 shows the resolution of the DSP system
is superior to the 4 µs Gaussian system. Again, the DSP allows
the spectroscopist to achieve a significantly better tradeoff between
throughput and resolution.
LOSS FREE COUNTING APPLICATIONS
The correction of the Live Time Clock as described above,
effectively extending the counting time to account for those periods
when the system could not accept an input, is adequate for most
samples, in particular those for which the count rate is relatively
constant. However, for short half-lived samples, or samples whose
constituents change (as in a flow monitoring application), this
method will not be accurate. In addition, even if the “counts per
unit time” are accurate using the traditional method for dead time
correction, the “real” counting time will have been extended by an
amount equal to the dead time, which may in fact increase the
actual collection time to an undesirable length.
The principal goal of Loss Free Counting (LFC) is to insure that
at the end of any data acquisition interval, the electronics have
accumulated all of the events that occurred regardless of any dead
time that may have been present in the system. LFC is based on the
concept that by adding “n” counts per event to an MCA’s channel
register, rather than digitizing and storing a single count at a time, a
“zero dead time” energy spectrum can be accumulated that assures
all counts are included in the spectrum. Assuming that “n” is correctly
derived, (“n” should equal “1” plus a “weighting factor” representing
the number of events that were lost since the last event was stored),
and the data is truly random in nature, the concept is statistically
valid. The factor “n” is derived on a continuous basis by examining
the state of the Amplifier and ADC every 200 ns. The proportion of
time during which the Amplifier and ADC are processing a pulse
provides a measure for the magnitude of the weighting factor “n”,
which is updated every 20 µs. Loss free counting requires that the
MCA support “add-n” or multiple “add-one” data transfer; consult
the factory for details.
Figure 1.32 A comparison of the resolution between an analog
system optimized for maximum throughput and a DSP system
set for a similar throughput
Figure 1.33 A comparison of the system throughput as
a function of input count rate for a DSP and two
analog systems optimized for resolution
Figure 1.34 A comparison of the system resolution as
a function of input count rate for a DSP and two
analog systems optimized for resolution
Unfortunately, counting statistics in a Loss Free Counting system
cannot be calculated from the corrected spectrum. One basic
assumption used by all peak fitting algorithms is that of Poisson
counting statistics. That is, the uncertainty of the counts is
proportional to the square root of the number of counts. While this
assumption is true for traditional “add-1” front-ends, it is not true
of the “add-n” Loss Free Counting front-end. This assumption is
especially poor as the weighting factor becomes large. To properly
quantify the uncertainty in each channel’s contents, the peak fitting
program must have access to both the corrected “add-n” and the
uncorrected “add-1” spectra. Therefore, CANBERRA also offers a
“Dual-LFC” feature with the Lynx which allows the collection of both
of these spectra so that statistically correct peak filling can occur.
PIPS DETECTORS AND MULTICHANNEL ANALYZERS
Alpha spectroscopy measurements of low-level samples require
long counting times. A large area PIPS detector, when configured
with a CANBERRA alpha spectrometer and multichannel analyzer,
provides a high resolution, low background, counting system that
will satisfy a multitude of alpha spectroscopy applications.
An example of a single chamber alpha spectroscopy system (that
can easily be upgraded) is illustrated in Figure 1.11. Note that the
Model 7401 Alpha Spectrometer is a complete, self-contained,
double-width NIM module that contains a vacuum chamber, vacuum
gage, detector bias supply, preamplifier/amplifier, SCA, counter/
timer and pulser for setup and test. Multiple Model 7401 Alpha
Spectrometers can be configured with a vacuum system that allows
individual vacuum chambers to be opened and loaded without
affecting the vacuum or data acquisition of the other spectrometers.
However, where numerous samples are counted simultaneously, it
is more cost effective and user efficient to select a system based
on the Alpha Analyst (Figure 1.35). This turn-key system supports
multiple detectors in a comprehensive software environment
featuring full computer control of all vacuum elements and
acquisition electronics.
Ethernet
Network
Client
and/or
Apex-Alpha™
Alpha Spectroscopy
Server
Figure 1.35 Example Large Scale Alpha Spectroscopy System based on the Alpha Analyst
© 2010 Canberra Industries, Inc. All rights reserved.