Modeling of IGBT Resistive and Inductive
Turn-On Behavior
L. Lu, S.G. Pytel, E. Santi
A.T. Bryant
J.L. Hudgins
P.R. Palmer
Department of Electrical
Engineering
University of South Carolina
Columbia, SC 29208
santi@engr.sc.edu
Department of Engineering
University of Cambridge
Trumpington Street
Cambridge, UK
CB2 1PZ
a.t.bryant.97@cantab.net
Department of Electrical
Engineering
University of Nebraska
Lincoln, NE 68588-0511
Department of Electrical and
Computer Engineering
University of British Columbia
2356 Main Mall
Vancouver, BC V6T 1Z4
Canada
Abstract— Although IGBT turn on losses can be comparable to
turn off losses, IGBT turn on has not been as thoroughly studied
in the literature. In the present work IGBT turn on under
resistive and inductive load conditions is studied in detail through
experiments, finite element simulations, and circuit simulations
using physics-based semiconductor models. Under resistive load
conditions, it is critical to accurately model the conductivity
modulation phenomenon. Under clamped inductive load
conditions at turn-on there is strong interaction between the
IGBT and the freewheeling diode undergoing reverse recovery.
Physics-based IGBT and diode models are used that have been
proved accurate in the simulation of IGBT turn-off.
Keywords: IGBT turn-on, interaction, modeling
I.
INTRODUCTION
Accurate IGBT models are desirable in order to simulate
switching waveforms and estimate device stresses, and
switching and conduction losses in converter applications. A
complete physics-based electro-thermal IGBT circuit simulator
model has been presented recently [1,2]. Its high accuracy has
been validated for various IGBT structures, including punchthrough (PT), non-punch-through (NPT), light-punch-through
and field-stop (FS). Its usefulness is enhanced by a practical
parameterization procedure and reasonable simulation speed
[3].
Usually in the study of IGBTs, the attention is focused on
the turn-off behavior, since the IGBT current tail causes
significant losses. The IGBT usually operates under hard
switching (clamped inductive load) conditions. Therefore,
validation for the physics-based model has been performed in
this environment.
Device manufacturers have expended significant effort to
reduce current tail losses, using techniques such as lifetime
control in the buffer layer to optimize device characteristics.
On the other hand, IGBT turn-on losses can be significant, due
to the diode reverse recovery, and may be comparable to turnoff losses. IGBT turn-on behavior so far has received scarce
attention in previous literature. The effect of diode and IGBT
interaction on total device losses has been noted in [4], and
some investigation of IGBT turn-on has been presented in [5].
This work was supported by the U.S. Office of Naval Research under
Grant N00014-02-1-0623.
In this work the physics-based IGBT model is used to
simulate turn-on behavior under resistive and clamped
inductive switching conditions. Accurate modeling of IGBT
turn-on behavior is challenging and the demands that this type
of simulation makes on a model are different from those made
by turn-off simulation.
For resistive switching it is essential to model accurately
the conductivity modulation process at turn-on. This causes a
“voltage tail” in the voltage waveform. For inductive turn-on,
besides the conductivity modulation process, there is a strong
interaction between the IGBT that is turning on and the diode
that is undergoing reverse recovery. For this reason it is
necessary to use accurate physics-based models for both the
IGBT and diode.
In section II the physics-based IGBT model is briefly
reviewed. In section III IGBT turn-on behaviour is discussed in
detail. Finite element simulations are used to provide guidance
to indicate which phenomena are important and to help identify
shortcomings in the analytic models used. Experimental
validation of the model under inductive and resistive switching
conditions for both turn-on and turn-off are presented in section
IV. A discussion of the results is given in section V, including
implications for the simulation of IGBT turn-on and the design
of the circuit.
II.
PHYSICS-BASED IGBT CIRCUIT SIMULATOR MODEL
The behavior of conductivity modulated devices, such as
PIN diodes and IGBTs, depends heavily on the excess carrier
(charge) distribution in the wide lightly-doped drift region. In
modern devices, the charge profile has a one-dimensional form
over most of its volume. Thus, a one-dimensional solution is
adequate for the bulk of the device. Space-charge neutrality is
maintained with the majority carrier profile closely matching
the minority carrier profile (quasi-neutrality). Under these
conditions, and assuming high-level injection, the charge
dynamics are described by the ambipolar diffusion equation:
D
∂ 2 p ( x, t ) p ∂p( x, t )
= +
∂x 2
∂t
τ
(1)
A Fourier-based solution for this equation has been
developed [1,6]. The second-order partial differential equation
is converted into a set of ordinary differential equations with
series coefficients p0…pk…pM-1 forming M equivalent R-C
cells. The representation requires the width of the undepleted
region and the hole and electron currents at the boundaries of
the region (x1 and x2), which provide the boundary conditions.
This model can be used for both diodes and IGBTs provided
appropriate boundary conditions are used. Physics-based
electro-thermal models for diodes and IGBTs using the
Fourier-based solution method have been developed
accordingly and are described in [1,2,7].
III.
In an IGBT, the turn-off behavior is predominantly
dependent on the amount of charge stored in the drift region.
The dV/dt and dI/dt at turn-off and the subsequent current tail
are largely determined by the rate of charge extraction. Turnoff losses are only weakly dependent on the gate circuit [9].
On the other hand, IGBT turn-on is largely a majority
current phenomenon, determined by the MOSFET part of the
IGBT. For this reason, turn-on losses are highly dependent on
the gate drive circuit. A fast gate drive can significantly reduce
losses. However, other considerations such as diode reverse
recovery and short circuit behavior must be considered in the
gate circuit design [9].
A. IGBT Turn-On Under Inductive Load Conditions
The IGBT turn-on process under inductive load conditions
is described in [9]. Fig. 1 shows the chopper cell circuit used
to model inductive switching. The load current commutates
from the freewheeling diode to the IGBT. The diode reverse
recovery current flows through the IGBT causing a significant
overcurrent. After the excess carriers in the diode drift region
have been removed (or have recombined), the diode recovers
its blocking capabilities and the reverse voltage applied to the
diode quickly increases. Depending on the diode construction,
the diode recovery may be abrupt, causing significant
overvoltage and oscillations [10]. Soft recovery diodes are
designed to mitigate this problem.
B. IGBT Inductive Turn-On Operation in Detail
The turn-on process of the IGBT is dominated by the
behaviour of the MOS region. Fig. 2 gives a typical switching
waveform for the turn-on behaviour. The five phases are given
by:
L0
RS
R0
FWD
IL
VDC
LS
LG
IGBT TURN-ON BEHAVIOR
Under hard switching, diode reverse recovery at turn-on
causes significant losses, both in the IGBT and diode, and there
is strong interaction between the devices at this switching event
[4]. Frequently this forces device and circuit designers to slow
down the gate drive in order to mitigate ringing and
electromagnetic interference problems caused by snappy diode
reverse recovery. In [8] extensive turn-on and turn-off losses
are reported for both PT and NPT IGBTs. Turn-on losses are
generally larger than turn-off losses, even without inclusion of
the significant diode turn-off losses that occur during IGBT
turn-on.
CS
RG
IGBT
VGG
LE
Fig. 1. Circuit for IGBT inductive turn-on.
-
Phase 1: gate voltage rise to threshold;
-
Phase 2: collector current increase;
-
Phase 3: forward voltage recovery;
-
Phase 4: gate voltage plateau;
-
Phase 5: final gate voltage rise.
1) Turn-on Phase 1
During phase 1, the gate-emitter voltage VGE is below the
threshold voltage VTH and the device is off. The gate
capacitance consists mostly of the gate-emitter capacitance CGE
since the gate-collector (Miller) capacitance CGC is small.
Hence VGE rises exponentially, at a rate set by the gate
resistance RG and CGE.
2) Turn-on Phase 2
Once VGE reaches VTH at the start of phase 2, the MOS
channel starts to conduct and allows electrons to flow into the
drift region under the gate, through the drift region and into the
IL+IRR
VDC
IL
VGG(on)
VCE
IC
VGE
VTH
Depletion layer voltage
0
time
Drift region
voltage drop
VGG(off)
1
2
3
Fig. 2: Detail of the IGBT turn-on process.
4
5
3) Turn-on Phase 3
The collector current IC therefore increases to a peak of
IL+IRR; at this point, coincident with the start of phase 3, VCE
begins to decrease as the diode voltage is now falling. The
charge level in the drift region continues to build up as the
collector current is high, and the collector voltage decreases
further as the depletion layer shrinks towards the MOS end of
the drift region.
4) Turn-on Phase 4
Once the drift region adjacent to the MOS channel comes
out of depletion and the accumulation layer under the gate
begins to form, the gate-collector (Miller) capacitance CGC
begins to increase. This sharply reduces the rate at which the
depletion layers can shrink, and signals the start of the gate
turn-on plateau during phase 4.
The decrease in collector voltage during the gate plateau is
rarely an observation of only the depletion layer voltage. This
is due to the forward voltage recovery of the drift region as it
becomes conductivity modulated. The level of stored charge is
still low, due to the high lifetime of the IGBT. This results in a
high forward voltage drop across the drift region. As the charge
level builds up, the drift region voltage drop decreases towards
the final on-state value.
The excess ambipolar charge stored adjacent to the
accumulation layer in the IGBT on-state takes a finite time to
increase. As it builds up, it has the effect of decreasing the drift
region voltage drop as it forms, since it allows the minimum
carrier density in the drift region under the gate to increase
from the drift region doping level. Since the drift region
voltage drop mostly occurs across this region at this stage, it
C. ATLAS Simulation Results for Inductive Turn-On
Figs. 3 and 4 show the hole concentration under the gate
and electric field under the P-well respectively for a NPT
planar IGBT during turn-on. Throughout the process the hole
concentration is effectively equal to the ambipolar carrier
density within the drift region. While it is strongly dependent
16
2
x 10
Off-state
0.345µs
0.477µs
0.527µs
0.542µs
0.573µs
0.729µs
2.962µs
15.462µs
1.8
1.6
-3
Hole concentration (cm )
As the diode switches off, the charge stored within it must
be removed before its voltage can reverse. The diode current IA
falls below zero as the charge is extracted. The reverse current
peaks at IA = -IRR, at which point VAK begins to fall below zero
towards the off-state voltage (approximately -VDC). VCE may
rise a little as shown.
5) Turn-on Phase 5
Finally, once the collector voltage has reduced
significantly, the MOS channel voltage decreases to the onset
of linear operation, marking the start of phase 5. Any further
fall in channel voltage results from an increase in gate voltage
VGE, as the load current is approximately constant. The gateemitter voltage VGE now charges to the on-state gate voltage
VGG(on) at a rate set by RG and CGE+CGC.
1.4
1.2
1
0.8
0.6
0.4
0.2
0
0
50
100
150
Position (µm)
200
250
Fig. 3: Detail of the IGBT hole concentration under the gate during the turn-on
process simulated in Silvaco ATLAS for a NPT planar IGBT. The P emitter
(collector terminal) is at x=0 µm and the gate at x=270µm.
4
15
-1
The initial fall of the collector voltage VCE appears across
the stray inductance LS as a back-emf. Since there is little
stored charge in the drift region, VCE decreases at a rate set by
the depletion capacitances and the stray inductance. The
collector current IC increases approximately linearly from zero.
The load current IL is practically constant, so the diode current
IA (not shown) decreases from its forward current (equal to IL)
at an identical rate. The increase in IC corresponds to a rise in
gate voltage VGE from the threshold voltage to a level set by IC.
The collector current consists almost entirely of electron
current at the MOS end of the CSR, so the MOS channel
current can be assumed to equal the collector current. During
this phase the MOS channel is in saturated operation,
depending mostly on the gate-emitter voltage VGE and only
weakly on the collector voltage VCE.
decreases significantly as the sub-accumulation layer charge
builds up.
Electric field (Vcm )
P emitter, where they recombine with holes. This allows hole
injection to take place at the P emitter (collector terminal), and
carrier levels begin to build up in the undepleted drift region.
The depletion layer then contracts, discharging the IGBT
capacitance, and the collector voltage decreases accordingly.
x 10
Off-state
0.345µs
0.477µs
0.527µs
0.542µs
0.573µs
0.729µs
2.962µs
15.462µs
10
5
0
0
50
100
150
Position (µm)
200
250
Fig. 4: Detail of the IGBT electric field under the P-well during the turn-on
process simulated in Silvaco ATLAS for a NPT planar IGBT. The P emitter
(collector terminal) is at x=0 µm and the P-well at x=264µm.
on the collector current at the P emitter end of the drift region,
it takes several microseconds to increase to the on-state profile
throughout the whole drift region. Note the increase in carrier
density under the gate due to the accumulation region: this
starts to appear approximately 0.5µs after the start of turn-on.
This coincides with the start of phase 4, as the accumulation
layer builds up and the gate voltage plateau proceeds. At this
point, however, the drift region voltage drop is significant,
shown by the substantial electric field under the P-well in fig.
4. This demonstrates that the fall in VCE during turn-on obeys
two consecutive stages: the decrease in voltage from the
depletion layer recovery, which is a majority carrier effect due
to the MOSFET, and the peak in voltage from the drift region
forward recovery, which a bipolar effect.
D. IGBT Turn On Under Resistive Load Conditions
IGBT resistive turn on is described in [11,12]. This process
is simpler than in the inductive load case. The resistive load
circuit is shown in fig. 5. The inductance LS represents the
parasitic loop inductance. Waveforms for the resistive turn on
process are shown in fig. 6, characterized by four phases. Since
LS is small, equation (2) is satisfied at all times.
RLOAD
RG
VGG VGE
VDC
IC
VCE
IC
time
0
1
2
3
4
VGG(on)
VTH
0
Fig. 6. Typical waveforms for IGBT resistive turn-on.
At the start of phase 1 (t=0), the gate drive voltage VGG
goes high and the gate-emitter capacitance CGE starts charging.
This phase ends when the gate-emitter voltage reaches the
threshold voltage.
Phase 2 commences as the MOS channel in the IGBT starts
conducting. The collector-emitter voltage VCE decreases
rapidly in response to the voltage drop on resistor RLOAD. The
Miller capacitance CGC acts as feedback to limit the gradient of
the gate-emitter voltage. As a result, the gate-emitter voltage is
approximately constant during this interval. The behavior is
dominated by the MOSFET inside the IGBT.
In phase 3, the ohmic voltage drop in the drift region has a
significant effect on the voltage and current waveforms. The
drift region is initially depleted of charge and consequently it
has a large resistance. At the start of phase 3 this voltage drop
contributes significantly to the collector voltage VCE, and it
slows down the turn-on process. As charge accumulates in the
drift region, the drift region resistance drops due to increasing
conductivity modulation. Therefore the voltage waveform in
this phase is dominated by the charge dynamics in the drift
region. This explains the slow evolution of the voltage. The
gate-emitter voltage remains approximately constant in this
period due to the effect of the Miller capacitance.
E. Influence of Device and Circuit Characteristics
The IGBT turn-on process is critically affected by the
emitter inductance, IGBT MOS characteristics and, in the case
of inductive switching, the diode characteristics. These are
discussed in the following sections.
VCE
VGE
(2)
Again, this demonstrates that turn-on occurs in two stages:
a fast MOSFET stage in which the collector current increases
rapidly, and a slow bipolar stage dominated by conductivity
modulation of the drift region. In the second stage significant
losses may occur.
Fig. 5. Circuit for IGBT resistive turn-on.
VDC
VDC − VCE
RLOAD
Finally, in phase 4 the gradient of the collector voltage VCE
becomes too small to pin the gate-emitter voltage, which
completes its charging up to VGG.
LS
IGBT
IC =
time
1) Importance of the Emitter Inductance
The effect of the Kelvin emitter inductance LE is critical
during inductive turn-on. Typically this is 5-10nH for a 1cm2
IGBT chip. The large positive dIC/dt as the collector current
increases in phase 2 causes a back emf across this inductance,
and reduces the available gate-emitter voltage. This slows
down the rate at which the IGBT turns on. For larger values of
LE the gate voltage is slowed down sufficiently to cause a
significant inflexion in the collector voltage VCE, causing VCE to
actually increase before falling towards its on-state value.
2) Importance of the IGBT MOS characteristics
At the beginning of turn-on, when the IGBT collector
current is increasing, the IGBT behaviour is dominated by the
MOS channel and the gate capacitance. Accurate simulation of
IGBT turn-on waveforms is critically dependent on matching
the parameters controlling the MOS channel – VTH (threshold
200
150
Vge
Vce
100
100
50
Current (A)
Ic
200
0
0
Ia
-50
-100
-100
-200
Vak
-300
-150
-200
-400
-250
-500
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (s)
Fig. 7: Inductive switching waveforms for IGBT turn-on and diode turn-off
using RG=10Ω, at 300V and 100A. The gate voltage VGE is scaled by a factor
of 10.
300V 100A 10ohm IGBT turn-off
700
350
600
300
500
250
200
Voltage (V)
400
Vce
300
150
100
200
Current (A)
RESULTS
A. Inductive Switching
Figs. 7 and 8 give the inductive switching waveforms for
IGBT turn-on and turn-off respectively at 300V, 100A using a
gate resistance RG of 10Ω. Similar waveforms are given in
figs. 9 and 10 for 400V operation. IGBT turn-on waveforms
only are shown in figs. 11 and 12 for 200V and 150V operation
respectively, with a current of 100A and a gate resistance of
10Ω. A reduced gate resistance of 2Ω is used for turn-on in
fig. 13, with conditions of 200V and 100A. Fig. 14 shows the
effect of reduced current, with conditions of 300V, 50A and
10Ω.
250
400
300
50
100
Ic
0
0
-100
-50
Vge
-100
-200
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (s)
Fig. 8: Inductive switching waveforms for IGBT turn-off using RG=10Ω, at
300V and 100A. The gate voltage VGE is scaled by a factor of 10. Diode turnon waveforms are omitted for clarity.
600
400V 100A 10ohm IGBT turn-on/diode turn-off
300
400
200
Ic
200
Voltage (V)
IV.
Experimental and simulation results are given in the
following sections for both inductive and resistive switching
across a range of voltage and current conditions and for
different gate resistances.
The devices used to obtain
experimental waveforms were rated at 600V, 100A. The IGBT
was a fast-switching, avalanche rated NPT device
manufactured by APT, and the diode was a fast-recovery
device manufactured by Powerex. All tests were carried out at
room temperature. A large snubber was placed across the
diode to avoid severe oscillations during reverse recovery,
noting that the IGBT is fast switching. The simulation
waveforms for both the IGBT and diode were obtained using
the Fourier-based solution models implemented in PSpice.
Both models used seven terms in the Fourier series. The model
parameters were extracted using the methods given in [3].
500
100
Vge
Vce
0
0
Ia
-200
Current (A)
3) Importance of the Diode Reverse Recovery
The IGBT collector current, and therefore the IGBT turn-on
power dissipation, depends strongly on the diode reverse
recovery current. Correct modeling of the collector current
requires accurate simulation of the diode reverse recovery;
therefore the diode used in simulation must also be accurate
across a wide range of conditions. The softness of the diode –
the relative duration of the reverse recovery time before and
after the peak in reverse current – is also important, because
this affects the rate of conductivity modulation in the IGBT and
the subsequent decrease in collector voltage towards its onstate value.
300V 100A 10ohm IGBT turn-on/diode turn-off
Voltage (V)
voltage), KP (channel conductance) and λ (channel shortening
parameter) – and those controlling the gate capacitance – COX
(gate oxide capacitance per unit area), lm (intercell half-width),
A (active die area) and ai (intercell area ratio). The IGBT
output capacitance, which interacts with the stray inductances
LS and LE, is also dependent on the undepleted portion of the
drift region; therefore correct determination of the drift region
width WB is necessary.
-100
Vak
B. Resistive Switching
Figs. 15 and 16 show the resistive switching waveforms for
IGBT turn-on and turn-off respectively at 400V, 100A using a
gate resistance of 10Ω. Turn-on waveforms at 200V, 100A,
using a gate resistance of 10Ω, are given in fig. 17. The gate
resistance is reduced to 2Ω for the turn-on waveforms at 400V,
100A in fig. 18.
-400
-200
-600
-300
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (s)
Fig. 9: Inductive switching waveforms for IGBT turn-on and diode turn-off
using RG=10Ω, at 400V and 100A. The gate voltage VGE is scaled by a factor
of 10.
200V 100A 2ohms IGBT turn-on/diode turn-off
400V 100A 10ohm IGBT turn-off
200
600
150
250
400
200
150
300
Ic
200
100
200
50
Voltage (V)
400
Current (A)
Vce
Voltage (V)
500
Vge
Vce
100
0
Ia
100
50
Current (A)
800
0
-50
-100
Vak
Ic
0
0
Vge
-100
-300
-150
-400
-200
-200
-50
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
-250
-500
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (s)
Time (s)
500
200V 100A 10ohm IGBT turn-on/diode turn-off
Fig. 13: Inductive switching waveforms for IGBT turn-on and diode turn-off
using RG=2Ω, at 200V and 100A. The gate voltage VGE is scaled by a factor of
10.
250
600
400
200
450
300
150
0
Ia
-100
-200
-100
-300
-150
Vge
Vce
0
50
0
Ia
-50
-150
-50
Vak
100
150
Voltage (V)
0
50
Current (A)
Vge
Vce
150
Ic
100
100
200
300
Ic
200
300V 50A 10ohm IGBT turn-on/diode turn-off
Current (A)
Fig. 10: Inductive switching waveforms for IGBT turn-off using RG=10Ω, at
400V and 100A. The gate voltage VGE is scaled by a factor of 10. Diode turnon waveforms are omitted for clarity.
Voltage (V)
-200
Vak
-400
-200
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
-300
-100
-450
-150
-600
3.0E-07
Time (s)
5.0E-07
7.0E-07
9.0E-07
1.1E-06
1.3E-06
1.5E-06
1.7E-06
-200
1.9E-06
Time (s)
Fig. 11: Inductive switching waveforms for IGBT turn-on and diode turn-off
using RG=10Ω, at 200V and 100A. The gate voltage VGE is scaled by a factor
of 10.
Fig. 14: Inductive switching waveforms for IGBT turn-on and diode turn-off
using RG=10Ω, at 300V and 50A. The gate voltage VGE is scaled by a factor of
10.
150V 100A 10ohm IGBT turn-on/diode turn-off
250
200
200
600
Vge
150
150
100
100
0
Ia
-50
-50
-100
Vak
-100
-150
-150
-200
-200
-250
-250
-300
3.00E-07 5.00E-07 7.00E-07 9.00E-07 1.10E-06 1.30E-06 1.50E-06 1.70E-06 1.90E-06
-300
Time (s)
Fig. 12: Inductive switching waveforms for IGBT turn-on and diode turn-off
using RG=10Ω, at 150V and 100A. The gate voltage VGE is scaled by a factor
of 10.
125
Ic
400
50
0
150
500
Voltage (V)
Voltage (V)
Vce
Current (A)
Ic
50
400V 100A 10ohm IGBT resistive turn-on
300
100
75
200
Vge
100
50
25
Current (A)
250
Vce
0
0
-100
-25
-200
-50
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (S)
Fig. 15: Resistive switching waveforms for IGBT turn-on using RG=10Ω, at
400V and 100A. The gate voltage VGE is scaled by a factor of 10.
600
400V 100A 10ohm IGBT resistive turn-off
and turn-off across all conditions. Turn-off waveforms are
included in the results in section IV to demonstrate that the
parameters required to give good turn-on matching are also
valid for turn-off.
150
125
500
400
100
300
75
200
50
25
100
Current (A)
Voltage (V)
Vce
Ic
0
0
Vge
-25
-100
-50
-200
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (S)
Fig. 16: Resistive switching waveforms for IGBT turn-off using RG=10Ω, at
400V and 100A. The gate voltage VGE is scaled by a factor of 10.
200V 100A 10ohm IGBT resistive turn-on
150
125
250
Ic
200
75
150
Voltage (V)
100
Vge
50
100
50
Vce
25
Current (A)
300
0
0
-50
-25
-100
-50
-75
-150
2.0E-07 4.0E-07 6.0E-07 8.0E-07 1.0E-06 1.2E-06 1.4E-06 1.6E-06 1.8E-06 2.0E-06
Time (S)
Fig. 17: Resistive switching waveforms for IGBT turn-on using RG=10Ω, at
200V and 100A. The gate voltage VGE is scaled by a factor of 10.
400V 100A 2ohm IGBT resistive turn-on
500
Voltage (V)
120
Ic
100
400
80
300
60
200
Vge
40
20
100
Current (A)
600
Vce
0
0
-20
-100
-40
-200
1.0E-07 2.0E-07 3.0E-07 4.0E-07 5.0E-07 6.0E-07 7.0E-07 8.0E-07 9.0E-07 1.0E-06
Time (s)
Fig. 18: Resistive switching waveforms for IGBT turn-on using RG=2Ω, at
400V and 100A. The gate voltage VGE is scaled by a factor of 10.
V.
DISCUSSION
The results in figs. 7-18 show excellent agreement of
simulation waveforms with those obtained experimentally. It
was necessary to adjust the parameters of both devices and the
circuit carefully to obtain consistent matching at both turn-on
In particular, the emitter inductance LE, IGBT MOS
parameters and diode parameters all strongly affected the IGBT
turn-on waveforms for clamped inductive switching. Accurate
prediction of IGBT turn-on losses is therefore dependent on
correct estimation of these parameters. Although the diode
losses are smaller than those of the IGBT, the diode reverse
recovery current is reflected in the IGBT collector current
waveform at turn-on; therefore correct modeling of the diode is
essential when estimating IGBT turn-on losses.
A. Inductive Switching
The models have successfully captured the diode reverse
recovery waveforms and the inflexion in the IGBT collector
voltage during turn-on in figs. 7, 9, and 11-14. The gate
waveforms are well matched, although there are discrepancies
at the beginning of turn-on and the end of turn-off. It is
suspected that this is caused by extra impedance in the gate
drive not taken into account in the model. This may be due to
a delayed ability to source enough current for switching the
IGBT, thus slowing down the initial rate of change in gate
voltage.
The turn-off waveforms in figs. 8 and 10 are less accurately
matched. This is due to two main factors: the non-linearity in
the gate drive already discussed, and avalanche within the
IGBT. The IGBT is rated at 600V, and the speed at which it is
switched causes the peak in collector voltage at turn-off to
approach or exceed 600V, thus causing avalanche. This is also
evident from the flat shape of the collector voltage peak,
resulting from the increased IGBT capacitance caused by extra
carriers generated during avalanche. The early current fall in
simulation indicates that the small snubber has some
inductance not modeled.
B. Resistive Switching
The resistive switching waveforms in figs. 15-18 show
good matching. All the principal features are captured,
including the gate voltage inflexion at the beginning of turn-on.
The error at negative gate voltages is, as for inductive
switching, due to the gate drive non-linearity not modeled here.
The inflexion in the gate voltage arises from the high dI/dt
appearing in the stray emitter inductance LE. The lack of a
suitable model for MOS injection into the intercell region
accounts for the slow reduction in collector voltage drop during
phases 3 and 4.
C. Interaction of the IGBT and Diode
Comparing the turn-on waveforms in figs. 11 and 13 shows
the effect of the gate resistance RG on the IGBT turn-on and
diode turn-off. With a gate resistance of 10Ω, there is a
noticeable inflexion in the collector voltage as the current
commutates from the diode to the IGBT. With a resistance of
only 2Ω, the initial rate of decrease in collector voltage is
significantly faster, and the collector voltage achieves a low
value quickly. The two-stage turn-on process is now seen
clearly, with the fast initial fall in collector voltage due to the
behaviour of the MOS channel and gate, and the subsequent
reduction in collector voltage due to the bipolar forward
voltage recovery.
The effect on the diode recovery waveforms is that for
faster IGBT switching (reduced gate resistance) the diode peak
reverse recovery current and voltage both increase from 250A
and 350V to 300A and 400V for 10Ω to 2Ω respectively. This
reduction is not as great as might be expected. The effect of a
reduced gate drive resistor is mitigated by the rate of
conductivity modulation and the stray emitter inductance. The
gate resistance must therefore be chosen carefully, emphasising
the need for the IGBT, diode and circuit to be matched [4]. A
strategy to reduce the peak diode reverse recovery voltage
while maintaining low switching losses is described in [13]
using active voltage control of the gate.
D. Modeling of the IGBT Drift Region Voltage Drop
The turn-on waveforms observed at various voltages (figs.
7, 9, 11 and 12) show that the reduction in IGBT drift region
voltage drop in stage 3 of turn-on is less accurately modeled
than the initial fall in collector voltage. This is due to the lack
of a suitable model for electron injection into the drift region
from the MOS channel. As explained in section III, the
behaviour of the IGBT depletion layer and excess carriers is
complex and two-dimensional in this region and is closely
coupled to the diode voltage. The development of suitable
models for MOS injection, excess carrier storage near the gate
accumulation layer and their coupling to the gate capacitance in
this region of the device is essential for accurate simulation of
this stage of IGBT turn-on. It also promises to provide a more
accurate model for the IGBT on-state voltage drop. This work
is currently in progress.
VI.
CONCLUSIONS
This paper has presented an improved understanding of the
IGBT turn-on process, using finite-difference simulation. The
importance of the emitter inductance, IGBT MOS
characteristics and capacitance evolution, and diode recovery
in IGBT turn-on has been explained.
Accurate simulation of turn-on using compact physicsbased models (utilizing the Fourier-based solution method) has
been demonstrated across a wide range of conditions for both
inductive and resistive switching. A full solution of the ADE
was required for the diode as well as the IGBT. Strong
interaction between IGBT and diode at turn-on has been
shown, with the diode significantly affecting IGBT losses.
Turn-off waveforms are also well-matched for the same
parameters, showing that the models are consistent for turn-on
and turn-off. The need for an accurate model of the IGBT
intercell region (encompassing MOS injection into the drift
region and its interaction with the gate capacitance) has been
identified.
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