designfeature By Robert Bell, National Semiconductor Corp
VARIOUS ISOLATED DESIGN APPROACHES HAVE THEIR
OPTIONS, DIFFICULTIES, AND TRADE-OFFS. AN IMPORTANT
DESIGN DECISION IS WHETHER TO USE THE PRIMARY-SIDE
GROUND OR THE SECONDARY-SIDE GROUND AS THE
CONTROLLER’S REFERENCE.
Crossing the boundary:
strategies for feedback
across an isolation barrier
esigners often categorize power converters This decision determines most of the basic configinto two basic types: isolated and nonisolated. uration.
These categories refer to the relationship beOf the two types, the primary-side-referenced
tween the input power ground and the output pow- controller configuration is less complex and the
er ground. Many applications require isoSECONDARY
OUTPUT
INPUT
lation between the two grounds. The
RECTIFICATION FILTER
FILTER
TRANSFORMER
isolation requirement often stems from
+
various safety agencies, and the main
INPUT
POWER
purpose of isolation is to protect personSTART-UP
nel from exposure to dangerous voltage
POWER
levels. In some cases, the grounds must
have sufficient isolation so that
PRIMARY
Figure 1
applying a potential of 1500V or
SWITCH
CONTROLLER
more between them shows no indication
FEEDBACK
of breakdown. The specification that
ISOLATION
quantifies this isolation requirement is a
(a)
small leakage current. An isolated powerSECONDARY
OUTPUT
converter design imposes several extra
INPUT
RECTIFICATION FILTER
FILTER
TRANSFORMER
design challenges on a power-supply designer. Transmitting power or feedback
+
INPUT
information from one ground reference
POWER
to the other is often referred to as “crossing the boundary.”
Given that there are two separate
PRIMARY
grounds, the first design task is to assign
SWITCH
the input- or output-ground reference to
GATE-DRIVE
particular circuits. All switching power
CONTROLLER
ISOLATION
converters include an input filter, an output filter, a transformer, a primary-side
switching element, a secondary-side recSTART-UP
tification element, and a controller cirPOWER
cuit. The center of the converter is some
(b)
type of controller. The reference for the
controller can be either the primary- or The reference for the controller can be the primary-side ground (a) or the
the secondary-side ground (Figure 1). secondary-side ground (b).
D
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+
OUTPUT
POWER
+
OUTPUT
POWER
May 30, 2002 | edn 75
designfeature Crossing the isolation boundary
most common configuration. Both configurations generally use a scheme that
derives bias power for the controller from
REFERENCE
a start-up circuit and then derives a more
VOLTAGE
efficient bias power from an auxiliary
winding during normal operaFigure 2
tion. The efficiency is high because the auxiliary winding steps down
(a)
the bias power from the transformer.
PRIMARY-SIDE CONTROL
AAMP
10,000
AISO
1
APWR
10
VOUT
+
_
AISO
1
VERROR
REFERENCE
VOLTAGE
FEEDBACK
ISOLATION
AAMP
10,000
POWER
STAGE
APWR
10
The problem with the secondary-sideVOUT
+
referenced controller is that the circuit
POWER
must derive bias power from the priSTAGE
FEEDBACK
_
mary-side power—the opposite ground
ISOLATION
VOUT(ISOLATED)
in this case—on initial power-up. You
can overcome this problem in one of two
ways. You can add a separate isolated bias
power supply to supply the few hundred
milliwatts necessary for the secondary- For a power converter with the error amplifier on the secondary side, the static error of the output
side controller. This bias supply can run voltage, assuming an ideal reference and no offsets, is simply equal to 1/(AAMP⫻AISO⫻APWR) (a). For
all the time because it is more efficient a power converter with the error amplifier and reference on the primary side and an isolated copy
than a linear start-up regulator. The sep- of the output voltage crossing the boundary, the isolation amplifier is part of the feedback network
arate-bias-supply approach adds com- and not part of the forward gain (b).
plexity but ensures an orderly start-up
under all starting conditions. The second
VOUT
approach to derive bias power for
REFERENCE
F
i
g
u
r
e
3
the secondary controller is to deGENERATOR
+
sign a scheme that causes the main priR3
ERROR
mary-side switching FET to start switchAMP
OPTOCOUPLER
DRIVER PWM
VOUT
ing immediately at power-up in a
⫺
RETURN
R2
somewhat-controlled manner. As the
+
switching commences, the auxiliary
REF
winding starts to provide the required
R1
ZENER
CONTROLLER
bias power to the controller, which then
takes control of the main switching FET.
(a)
This approach of blindly starting the
VOUT
main switching FET can have problems
with overshoots, short circuits, and exREFERENCE
GENERATOR
cessive loading conditions.
+
R3
You might well ask, then, why a deERROR
AMP
OPTOCOUPLER
DRIVER PWM
signer would ever want to configure a
VOUT
⫺
converter with a secondary-side-referRETURN
R2
+
enced controller. In Figure 1a, an isolatR4
ed feedback signal must cross from the
REF
FB
primary ground to the secondary
R1
LMV431
CONTROLLER
ground. This feedback uses few compoR5
nents but always suffers from some quan(b)
tity of phase delay. This delay limits the
achievable bandwidth and ultimately the
transient response of the converter. Re- A simple approach to deriving the error signal uses a zener diode and an optocoupler (a).
moving the need for the optocoupler that Replacing the zener diode with an LMV431 shunt regulator improves accuracy (b).
isolates the feedback signal ultimately increases the converter reliability; the char- These synchronous FETs require a con- troller can adapt the timing due to operacteristics of these devices tend to shift trol-gate drive. With the controller on the ating conditions and directly control the
with age and temperature. Many con- secondary-side ground, the controller FET gates. Thus, a secondary-side-converters today use FETs instead of diodes can now directly drive the synchronous troller scheme is more complex but can
as a means of secondary rectification, FETs. This situation can lead to opti- perform better than can a primary-sideknown as synchronous rectification. mized control timing, because the con- referenced controller.
76 edn | May 30, 2002
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+
+
If the performance reVOUT
quirements are less
F
i
g
u
r
e
4
COMPENSATION
stringent, then using a
+
NETWORK
primary-side-referenced
controller can make the conVOUT
verter simpler and less exREFERENCE
RETURN
GENERATOR
pensive. In this configura⫺
tion, the main transformer
⫺
OPTOCOUPLER
ERROR
+
AMP
+
transmits power across the
DRIVER PWM
LM358
⫺
boundary, and an optocou+
pler or magnetic pulse trans+
former provides feedback
REF
LM4050
from the output back across
CONTROLLER
the boundary to the primary
side. An optocoupler is the
most common approach due
to its lower cost and complexity. The feedback signal is A dual op amp optimizes the loop compensation and ensures the proper polarity of the feedback signal.
generally not a signal proportional to the output voltage. Rather, proaches with typical assigned gains for ror amplifier and reference on the prithe feedback signal represents the differ- each block. AISO represents the gain of the mary side and an isolated copy of the
ence between the output voltage and a isolation stage, AAMP represents the gain output voltage,VOUT(ISOLATED), crossing the
fixed, precision reference voltage. If you of the error amp, and APWR represents the boundary. The high gain of the error amattempt to bring the output voltage di- gain of the pulse-width modulator and plifier and the power stage continuously
rectly across the boundary, any inaccu- the remainder of the power stage. The keep VOUT(ISOLATED) at the same potential
racy that the isolation circuit causes will only difference between the two ap- as the reference voltage. The isolation
directly affect the regulation. Most opto- proaches is that the error amplifier and amplifier in this case is part of the feedcouplers have wide gain tolerances and the isolation stage are transposed. Figure back network and not just part of the forlarge variations over temperature and 2a represents a power converter with the ward gain. In this case, the initial error
time. Alternatively, if the circuit com- error amplifier on the secondary side. with ideal components is also 0.001%.
pares the output voltage with a fixed ref- The static error of the output voltage, as- However, if the isolation-stage gain deerence and then multiplies the result by suming an ideal reference and no offsets, creases by a factor of two, the system era large gain, the resulting signal will be is simply equal to 1/(AAMP⫻AISO⫻APWR). ror increases to 100%.
Consider an example for which the
just an error signal. When the circuit In this example, the error is 0.001%.
If the gain of the isolation stage, AISO, reference voltage and the output voltage
transmits this error signal across the isolation boundary, the error signal can tie decreases by a factor of two, the overall are initially 3.3V. The reference voltage
directly into the pulse-width modulator. error increases to only 0.002%. Figure 2b and VOUT(ISOLATED) remain locked togethFigure 2 shows two feedback ap- represents a power converter with the er- er even with an isolation-stage-gain deVOUT
COMPENSATION
NETWORK
+
+
+
Figure 5
REFERENCE
GENERATOR
DRIVER PWM
ERROR
AMP
ISOLATION
SAMPLE TRANSFORMER
AND HOLD
⫺
⫺
+
+
+
R1
+
VOUT
RETURN
⫺
+
LM358
C1
REF
+
LM4050
CONTROLLER
OSCILLATOR
TRANSFORMER
A winding from the main power transformer drives a switching (oscillator) transistor that periodically applies the error signal to the isolation transformer.
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May 30, 2002 | edn 77
designfeature Crossing the isolation boundary
crease of a factor of two. However, at that
time, the secondary-side output voltage
will be 6.6V.
DERIVE AN ERROR SIGNAL
Several configurations are possible for
deriving an error signal on the secondary side and bringing that signal across
the isolation boundary. The simplest approach is to use a zener diode and an optocoupler (Figure 3a). An increasing
output voltage increases the current in
the optocoupler diode, which leads to a
reduced output signal of the primary
controller’s error amplifier. This simple,
low-cost configuration is inaccurate due
to the zener-diode and optocouplerdiode tolerances.
A more accurate and more popular
configuration for medium-performance
applications uses the same basic idea in
Figure 3a but replaces the zener diode
with an LMV431 shunt regulator (Figure 3b). The LMV431 shunts current
through the device’s cathode until the
voltage present at the feedback pin is
1.24V. Using this device, you set up a
voltage divider with R4 and R5 such that
VOUT⫽1.24((R4⫹R5)/R5). This configuration is more accurate than a zenerdiode configuration because the initial
tolerance of the shunt regulator is as low
as 0.5%. Also, the voltage drop across the
optocoupler diode is no longer part of
the feedback divider. You can connect
loop compensation between the cathode
and the feedback pins of the LMV431.
Some loop compensation is necessary
for all of these configurations. In some
applications using the LMV431 configuration, optimizing the compensation
can be difficult. Power-supply-rejection
issues can arise because this configuration applies disturbances on the VOUT
line to the cathode. Also, the amount of
voltage feedback at the cathode pin is
limited because the output of the
LMV431 is a current. Adding a dual op
amp provides all of the benefits of Figure 3b with the ability to optimize the
loop compensation (Figure 4). This error-amplifier configuration provides
high gain, high accuracy, and the ability
to compensate the loop. This circuit also
includes a separate temperature-compensated reference, the LM4050.
For all of these configurations, you
have to think through what happens in
start-up mode. When you initially apply
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input power, neither output voltage nor
voltage exists to bias any secondary-side
circuits. In this situation, the feedback
error signal must be the right polarity so
that the circuit requests full power. The
dual op amp in Figure 4 ensures the correct polarity. Another consideration during start-up is soft start. You can slow the
rate at which the output voltage rises by
increasing the capacitor across the reference device.
MAGNETIC SIGNAL TRANSFORMERS
Optocouplers are not the only devices
available to provide signal isolation.
Many military applications prohibit the
use of optocouplers due to age-degeneration effects. Another way to provide the
required isolation is by using a magnetic
signal transformer. Many methods use a
signal transformer to cross the boundary.
Each method requires a circuit that applies the signal to the transformer only
periodically because transformers do not
accommodate dc signals. One method is
basically a form of amplitude modulation
(Figure 5). This approach uses a winding from the main power transformer to
drive a switching transistor that periodically applies the error signal to the isolation transformer. A sample-and-hold circuit connects to the signal transformer’s
output winding. In this design, you need
to pay particular attention to the sampleand-hold circuit’s effect on the system
bandwidth (R1, C1), the duty-cycle range
of the chopper signal, and capacitorloading effects on the error amplifier.
Ultimately, the best design approach
to crossing the isolation boundary varies
with each application. Performance,
complexity, and cost are important considerations. Evaluation of the isolation
circuit against the system objectives is
necessary throughout each stage of the
design. Careful test and measurement is
necessary over all operating conditions,
including fault conditions, such as short
circuits and overloads.왏
Author’s bio graphy
Robert Bell is a principal applications
engineer at National Semiconductor Corp
(www.nsc.com). He has helped to define
several high-performance PWM controllers and associated application circuits.
He holds a BSEE from Fairleigh Dickinson
University (Teaneck, NJ) and enjoys hiking, camping, and tennis.
May 30, 2002 | edn 79