Phase Angle Balance Control for Harmonic Filtering of A Three Phase Shunt
Active Filter System
Souvik Chattopadhyay, V. Ramanarayanan
Power Electronics Group
Department of Electrical Engineering,
Indian Institute of Science, Bangalore 560012, India
souvik@ee.iisc.ernet.in ,vram@ee.iisc.ernet.in
Abstract-This paper proposes a new strategy for harmonic
filtering of a three-phase shunt active filter system. The shunt
harmonic filter’s control objective is defined as: balance the
phase angle of the input current with the phase angle of the
line frequency component of the load current. This objective is
achieved in discreet implementation without sensing the input
voltages. The controller uses a phase shifting method on the
sensed input current and then applies the resistor emulator
type input current shaping strategy on the phase-shifted
current. In implementation Texas Instrument’s DSP based
unit TMS320F240 EVM is used as the digital hardware
platform. The control algorithm is computationally simple yet
the harmonic filtering performance is high. The analysis,
simulation and experimental results of a three-phase shunt
active filter prototype on a 25A non-linear load are presented.
I. INTRODUCTION
Shunt active filter (SAF) is a current controlled voltage
source converter (VSC) of Boost topology ,as can be seen
in Fig.1. We can identify two major functions in the control
of shunt active filter. First : it must generate a current
reference containing only those harmonic components that
are present in the load current and second : a very high
bandwidth current controller needs to be implemented. This
current controller should be able to extract the same actual
current waveshape from the SAF as dictated by the
reference. The unique feature of active filter control is the
method of harmonic extraction. The method should be such
that each harmonic component in the reference is exactly
equal in magnitude and phase to the corresponding
harmonic component in the load current. But it is difficult
input
voltage
a-b-c
vg
3-ph
ig Source
2 Current
Sensors
Input current
OR
Filter current
ih
2 Current
Sensors
to design a harmonic filter that would separate out the
fundamental from the lower order harmonics without any
phase shift. So conventional filtering technique can not be
applied in a straightforward manner. Moreover there can be
unbalanced currents in the phases which may also have to
be filtered out from the input. The harmonic extraction
theories that have addressed these issues and provided
implementable solutions are the Instantaneous Active and
Reactive Power Method [2] ,the Synchronous Reference
Frame Method [3], the Instantaneous Active and Receptive
Current Component Method[4].
All these methods involve computations using the sensed
input voltage. But the input voltage waveform of a real life
system can have distortions. So to filter out the distortions
in the input voltage a phase locked loop (PLL) circuit is a
mandatory requirement for these methods. Design of a high
performance PLL is not easy and is an additional
computational burden on the digital controller.
This paper proposes a new strategy, denoted here as the
Phase Angle Balance (PAB) control technique , for
harmonic filtering of a three-phase shunt active filter
system. The control objective of a shunt active filter is
defined here as: balance the phase angle of the input current
with the phase angle of the line frequency component of the
load current. This objective is achieved in discreet
implementation without sensing the input voltage. The
controller uses a phase shifting method on the sensed input
current and then applies the resistor emulator type input
current shaping strategy on the phase-shifted current. Phase
Angle Balance control has the flexibility to compensate
only for the harmonics of the load current so
il
Shunt Harmonic Filter
Sw1
L
L
L
Sw3
Sw2
Sw4
Sw1
Sw2
Sw5
C
Sw3
Sw5
2 Current
Sensors
Voltage
Sensor
Sw6
Sw4
3-Ph Non-linear
Load
Sw6
Digital implementation (TMS320F240) of phase angle balance (PAB) controller for harmonic filtering
Fig.1 Schematic diagram of a three phase shunt active filter system with the proposed controller
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technique described in [1] is applied on the input current.
For determination of the phase shift load current needs to
be sensed.
Let us assume that the input voltage (v g ) is sinusoidal and
balanced.
β axis
vg
ig
ig ige jϕ
v g = v ga + jv gb = V g1 cos(z f t) + jV g1 sin(z f t)
α axis
ϕs
Control Objective:ig
z f is the angular frequency of the input voltage..(a, b) is a
system of orthogonal and stationary axes as shown in Fig.2.
The variables with suffix a or b represent components along
that particular axis. A line current shaping controller,
similar to the one that has been described in [1] for high
power factor operation of three-phase boost rectifier, can be
vg
Re
Fig.2 Control objective of a three phase shunt active filter
Â
used here to make (i g ) proportional to (v g ). It has been
that the current rating requirement of the converter is kept
lower. However if required this method can compensate for
the harmonic as well as the reactive contents of the load and
thereby shape the input current like input voltage. Any
other resistor emulator type active filtering strategy [5] is
not available with this flexibility.
In section II, we have explained the basic principle of
operation of the proposed Phase Angle Balance controller.
Two methods for the determination of the phase shift of the
input current are also provided. The simulation results of
the proposed control scheme are presented in section III. In
section IV the experimental results of the prototype shunt
active filter system are presented.
II. PHASE ANGLE BALANCE CONTROL
Phase Angle Balance Method is an extension of the current
shaping method that has been described in [1]. For high
power factor rectifier the input current is shaped to follow
the input voltage in each switching period . The difference
for the shunt active filter (SAF) is that the phase shifted
input current is made proportional to the input voltage. The
control objective for a three phase shunt active filter,
defined in Fig. 2, is to make the input current i g devoid of
higher order harmonics that is otherwise present in load
current i l . In order to keep the rating of the active filter
low, current in the shunt filter i h should contain only the
harmonic currents but not the reactive current of the load.
The reactive component of the load current has to be
supplied from the input side. Mathematically the function of
the shunt active filter can be expressed as
ig =
vg
Re
e −jw s
(2)
(1)
i g and v g are input current and voltage vectors respectively.
w s can be positive or negative depending on whether the
load is inductive or capacitive respectively. R e is the
emulated resistance corresponding to active power transfer
in the positive direction. The required phase shift depends
on the load power factor. In implementation either the input
current can be sensed directly or obtained by summation of
the load and the filter currents. The basic current shaping
Â
shown in Fig.2 that from the input current (i g ) , (i g ) can be
obtained by phase shifting by an angle v s . Therefore,
Â
Â
Â
i g = i ga + ji gb = I g1 e j(z f t) = I g1 cos(z f t) + jI g1 sin(z f t) = i g e jw s
= i ga cos(v s ) − i gb sin(v s ) + j(i gb cos(v s ) + i ga sin(v s ))
(3)
Where,
i g = i ga + ji gb = I g1 e j(z f t−v s )
(4)
It can be noted that if we can make the phase shift w s equal
to the phase angle of the positive sequence component of
the fundamental frequency load current then the input
current will consist only of active and reactive current
components of line frequency. This general principle of
filtering the harmonic currents is denoted here as the Phase
Angle Balance (PAB) control technique. The block diagram
of the PAB controller is given in Fig.3. We can use either
Method I , shown in Fig.4(a) or Method II, shown in
Fig.4(b), to get the desired phase angle w s .
Method I: For a three-phase three-wire system (i.e no zero
sequence current) , the non-linear load current (i l ) can be
expressed as a summation of harmonics of positive
sequence and negative sequence current components. It has
been assumed that there is no dc component in the load
current.
i l = S ºk=1 (i lk_p + i lk_n ), where k is the harmonic number (5)
In the suffix of a variable ‘p’and ‘n’ indicate positive and
negative sequence components respectively. The space
phasors in the above equation are decomposed into (a, b)
axis components as (6) and (7).
i la =
º
º
º
S i lka = k=1
S i lka_p + k=1
S i lka_n =
k=1
º
S (I lk_p cos(kz f t + v k_p ) + I lk_n cos(−kz f t + v k_n ))
k=1
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(6)
sin ϕ s
cos ϕ s
igα
igα
igβ
Phase
Shifting
Sw1
Digital Two
Axis Resistor
Emulator
Type Current
Mode
Dc Voltage Sensing Controller
And Space
Vo
Vector
Implementation
igα
ig
β
3-ph to 2-ph
conversion
input current sensing
load current sensing
ih
ig
il
Shunt
Active
Filter
3-ph to 2-ph
conversion
cos ϕ s
Magnitude
Calculation
PI Control ler
ig
ig
β
Sw2
Low Pass
Filter
Sw3
il
Sw4
Sw5
º
igβ
ilα
D-Q Transformation
of Load Current
º
ild
ilq
cos( θ)
sin( θ )
il1_pd
Low Pass
Filter
º
il1_pq
(8)
Il
(9)
The magnitude of the input current is made equal to the
magnitude of the load current by the outer loop PI
controller. Therefore the PI controller ensures that
S ºk=1 (I 2lk_p + I 2lk_n )
(10)
The output of the PI controller gives us the magnitude of
Â
sin(w s ),or sin(w s ) . By monitoring i gb and i lb we can
determine whether the load current is leading or lagging
the input voltage. This information is binary in nature
because it has only two values : m = +1 for lagging current
and m = −1 for leading current.
sin(w s ) = m sin(w s )
(11)
cos(w s ) =
(12)
1 − sin (w s )
2
p ωf
d1_
Ig
Il α θ
Igβ
φs Ig α
β
Il
All the frequency dependent terms are combined together as
fp
. We can eliminate the high frequency components by
a LPF, then
I g1 =x i l x fil =
ig β
β
p
q1_
Ilp_d
º
(I 2lk_p + I 2lk_n ) + f p
S k=1
x i l x fil = S ºk=1 (I 2lk_p + I 2lk_n )
sin ϕs
igα
Computation
of sin( θ )
and cos( θ )
The angular frequency of the fundamental component is z f
. Therefore the magnitude of (i l ) can be written as
i 2la + i 2lb =
cos ϕs
Computation of
cos ϕs and sin ϕs
Fig.4(b) Determination of sin(w s )and cos(w s )by Method II
(7)
S (I lk_p sin(kz f t + v k_p ) + I lk_n sin(−kz f t + v k_n ))
k=1
x i l x=
m
1=leading
-1=lagging
Fig.4(a) Determination of sin(w s )and cos(w s )by Method I
ilα
Fig.3 Block diagram of the Phase Angle Balance (PAB)
control technique.
º
Phase
Information
Magnitude
Calculation
ilβ
ilβ
S i lkb = k=1
S i lkb_p + k=1
S i lkb_n =
k=1
ilβ
ilα
Sw6
ilβ
i lb =
sin ϕ s
-Ilp_q
α
Il1_p (fundamental comp)
(fundamental+harmonics)
Fig.4(c) Load current space phasors (i l ), (i l1_p )in stationary
(a, b) and synchronous (d1_p, q1_p) reference frame
I g1 cos(w s ) = I l1_p cos(v 1_p )
(13)
w 1_p is the phase angle of the positive sequence component
of the fundamental frequency load current with respect to
the input voltage. Therefore
w s = cos −1 (
I l1_p
S ºk=1 (I 2lk_p + I 2lk_n )
cos(w 1_p )) =
(14)
cos −1 (K v cos(w 1_p ))
K w can be expressed as
Kw =
1
1+T 2D
(15)
Where the total distortion coefficient T D is defined as
This method of calculation of phase shift v s is
computationally simple but not very exact. The reason for
that is given below. From the power balance condition
TD =
I 2l1_n +
º
S k=2 (I
2
2
lk_p +I lk_n )
I 2l1_p
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(16)
30
30
30
30
Amp
20
20
20
20
Amp
Amp
10
Iga
Iga
Amp
10
10
10
Iga
0
0
Iga
0
0
-10
-10
-10
-20
-20
-30
-30
-10
-20
-20
-30
0
0.005
0.01
0.015
0.02
0.025
0.03
0.035
0
0
0.04
0.005
0.01
0.015
0.02
time in sec
0.025
0.03
0.035
0.005
0.01
0.015
0.02
0.025
0.03
0.035
0.04
-30
0
time in sec
0.04
0.005
0.01
0.015
5(a)
6(a)
5(b)
20
0.025
0.03
0.035
0.04
6(b)
30
20
30
0.02
time in sec
time in sec
15
15
Amp
20
Amp
20
10
Amp
10
Amp
10
10
5
Iha
0
Ila
5
Ila
Iha 0
0
0
-5
-5
-10
-10
-10
-10
-20
-20
-15
-20
0
0.005
0.01
0.015
0.02
0.03
0.025
0.035
0.04
-15
-30
-20
-30
0
0.005
time in sec
0.01
0.015
0.02
0.025
0.03
0.035
0
0.04
0.005
0.01
0.015
5(c)
0.02
0.025
0.03
0.035
0
0.04
0.005
0.01
0.015
0.02
6(c)
5(d)
0.025
0.03
0.035
0.04
time in sec
time in sec
time in sec
6(d)
30
30
1
1
0.8
20
Ild1
0.8
Cos(Phi)
0.6
Cos(Phi)
0.6
10
20
Amp
Il1d
10
0.4
0.4
Ilq1
0.2
0.2
Sin(Phi)
0
0
0
0
-0.2
Sin(Phi)
Il1q
-0.2
-10
-10
-0.4
-0.4
-0.6
-20
-20
-0.6
-0.8
-0.8
-1
-30
0
0
0.005
0.01
0.015
0.02
0.025
0.03
0.035
0.005
0.01
0.04
0.015
0.02
0.025
0.03
0.035
-30
0.04
-1
0
0
time in sec
0.005
0.01
0.015
time in sec
5(e)
5(f)
2
2
1.5
1.5
1
1
0.5
0
0
-0.5
-0.5
-1
-1
-1.5
-1.5
-2
0.015
0.02
0.025
time in sec
5(g)
0.035
0.005
0.01
0.04
0.03
0.035
0.04
0.005
0.01
0.015
0.02
0.02
0.025
0.03
0.035
2
2
1.5
0.04
1
0
0
-0.5
-0.5
-1
-1
-1.5
-1.5
-2
0
0.005
0.01
5(h)
0.015
0.02
0.025
6(g)
Ideally we would like the controller to make the input
current phase shifted by an amount equal to the phase angle
of the positive sequence component of the load current,
w s = w 1_p . Then the line frequency, positive sequence
current will to be supplied from the input at the optimum
power factor. However it can be seen from (14) that this
principle is not exactly followed in this implementation.
Instead a phase angle error is introduced that depends on
the total distortion coefficient T D of the load current. This
would mean that the input current though free from
0.035
0.5
time in sec
Fig.5. Simulation results of the phase angle balance
controller with load consisting of 20A(peak) diode rectifier
and 10A (peak) reactive current (a) input phase current i ga
Â
(b) i ga (c) filter current i ha (d) load currenti la (e) sin(w s )(f)
i l1_pd , i l1_pd (g)cos(h)(h)sin(h)
0.03
sin(theta)
-2
0.04
time in sec
0.025
time in sec
1.5
1
0
0.015
6(f)
0.5
-2
0.01
0.03
cos(theta)
0.5
0.005
0.025
6(e)
sin(theta)
cos(theta)
0
0.02
time in sec
0.03
0.035
0.04
0
0.005
0.01
0.015
0.02
0.025
0.03
0.035
0.04
time in sec
6(h)
Fig.6. Simulation results of the phase angle balance
controller with 20A(peak) rectifier load (a) input phase
Â
current i ga (b) i ga (c) filter current i ha (d) load current i la (e)
sin(w s )(f)i l1_pd , i l1_pd (g)cos(h)(h)sin(h)
harmonics will not deliver power at the optimum power
factor. However in most cases this error in magnitude of the
input current is insignificant. For example if T D . is 28% as
would be in case of a diode bridge rectifier then the input
current magnitude will be 3.8% more than what is possible
to achieve through perfect compensation.
º
2
2
So if we approximate S k=1 (I lk_p + I lk_n ) as I l1_p then
I g1 = I l1_p
(17)
v s = v 1_p
(18)
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Method II : It is possible to implement perfect
compensation for harmonics with some extra computational
requirements on the controller. The positive sequence
(19)
better dynamics can be extracted by the current controller.
However in this case the choice is dictated by the
limitations of the general purpose TMS320F240 digital
controller that has been used for the hardware
implementation
of
the
control
algorithm.
(20MHz − CPU , 6.6lSec − ADC ) The simulation results
for the method II of Phase Angle Balance control are
presented in Fig.5 and Fig.6. The results obtained from
method I are practically the same. In simulation two cases
are considered : (1) the load is diode rectifier type
non-linear and also has line frequency reactive current
component of 10A peak - is shown in Fig.5. and (2) the
load has no line frequency reactive component but consists
only of 20A peak non-linear current - is shown in Fig.6.
(20)
IV. EXPERIMENTAL VERIFICATION
component of the fundamental current i l1_p can be
extracted from the load current in the synchronously
rotating (d 1_p , q 1_p ) reference frame. For that , it is not
even necessary to sense the input voltage v g as would be
required in other methods. This is because the internal
control variable
Fig.4(c),
i /ga
cos(h) =
sin(h) =
Â
i g is proportional to v g . So, as shown in
(i /ga ) 2 +(i /gb ) 2
i /gb
(i /ga ) 2 +(i /gb ) 2
Therefore,
i l_pd = i la cos(h) + i lb sin(h)
(21)
−i l_pq = −i la sin(h) + i lb cos(h)
(22)
Subsequently we pass i l_pd and i l_pq through low pass filters
to eliminate the ac content in the waveform and get the dc
quantities (i l1_pd , i l1_pq ) corresponding to the positive
sequence fundamental component of the load current. From
(i l1_pd , i l1_pq ) the sin(w s )and cos(w s ) can be computed as
cos(w s ) =
sin(w s ) =
i l1_pd
(i l1_pd ) 2 +(i l1_pq ) 2
i l1_pq
(i l1_pd ) 2 +(i l1_pq ) 2
(23)
(24)
Alternatively the magnitude of i l1_p can be obtained from
(25) and the phase angle can be determined by the closed
loop PI controller as in Method I.
|i l1_p | =
i 2l1_pd + i 2l1_pq
(25)
III. SIMULATION RESULTS
The proposed controller that generates the switching pulses
for the power converter of the shunt active filter is
simulated in the MATLAB-SIMULINK (version 5.3)
simulation package. The non-linear load is of 20A peak.
The shape of the current and its rise and fall times are so
chosen that it approximately models a three-phase diode
bridge rectifier. The per phase inductance of the designed
shunt active filter is 0.75mH . The output dc voltage of
SAF is regulated by an outer loop PI regulator. The
switching frequency of the converter is chosen to be
10KHz. It is desirable that the switching frequency of the
SAF is made as high as possible. At higher switching
frequency the filter inductance can be made lower and
The proposed Phase Angle Balance harmonic filtering
algorithm is implemented on the general purpose digital
hardware platform of Texas Instruments DSP TMS320F240
(20MHz − CPU , 6.6lSec − ADC ). It has three 16-bit
registers [6] - ,CMP1, CMP2 and CMP3 to control the
individual duty cycles of the switches. We sense two phase
currents of the harmonic filter (instead of that the input
current can also be sensed) ,two phase currents of the
non-linear load and the output dc voltage of the shunt
harmonic filter. The phase shifted input current is shaped to
follow the input voltage using the resistor emulator type
digital current mode control algorithm that has been is
described in details in [1]. The power hardware of the
prototype PWM converter is built on IPM (Intelligent
Power Module) switched at 10KHz that corresponds to the
control loop time of T s = 100lSec. The per phase filter
inductance is 0.75mH. The filter and load currents are
measured but input voltages are not sensed. The output of
the shunt harmonic filter is regulated at 375V. A non-linear
load of 12.7A(rms) rating is constructed using a three-phase
diode bridge rectifier connected to the resistive load. For
verifying the performance of the phase shift control
algorithm , described in section II of this paper, a
predominantly reactive load of 24.7A(rms) , is constructed
by adding an induction motor under no load to the already
available nonlinear load of the diode bridge rectifier. The
measured value of THD of the rectifier load current is
25.7% and the THD of the input current with Phase Angle
Balance control is 6.5%. In this case the input voltage is not
an ideal sine wave but itself has 1.8% THD. There is no
substantial difference in the input current THD results
between the two methods of APB control that are proposed
in this paper. The experimental wave forms are shown in
Fig.7 and Fig.8. The harmonic performance can be further
improved by using a lower value of per phase inductance.
However in order to use a lower value of inductance the
switching frequency of the converter has to be increased.
For that a better controller than TMS320F240 needs to be
selected because its ADC conversion time of TMS320F240
is too high for higher frequency operation.
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7(a)
7(b)
7(c)
7(d)
Diode rectifier - ch1: input current, ch2: load current ,
ch3:active filter current ch4: sin(w s ) (from DAC)(d) Diode
rectifier plus IM(no load) - ch1: input current, ch2: load
current , ch3:active filter current ch4: sin(w s ) (e) Diode
rectifier - ch1: input current, ch2: load current , ch3: input
current magnitude i g (from DAC)ch4: filtered load current
magnitude i l fil (f) Diode plus IM (no load) - ch1: input
current, ch2: load current , ch3: input current magnitude
ch4: filtered load current magnitude (g) Diode plus IM (no
load) - without PAB control - for high power factor
operation- ch1: input current, ch2: load current , ch3: active
filter current ch4: sin(w s ) (h) Diode plus IM (no load) without PAB control - for high power factor operation- ch1:
active filter current , ch2:sin(w s ) , ch3: input current
magnitude i g ch4: filtered load current magnitude i l fil
Scale::i ga , i la - 25A/div, i ha - 10A/div, v ga - 170V/div, sin w s
- 1/div , i g , i l fil - 22.5A/div
8(a)
7(e)
7(f)
8(c)
7(g)
8(b)
8(d)
7(h)
Fig. 7. Experimental results of a Shunt Active Filter system
- Phase Angle Balance Control Method I (a) Diode rectifier
- ch1: input voltage (v ga ) , ch2: input current (i ga ), ch3:
load current (i la ) , ch4: active filter current(i ha ) (b) Diode
rectifier plus IM (no load) - ch1: input voltage , ch2: input
current, ch3: load current , ch4:active filter current (c)
8(e)
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8(f)
control is a very simple but effective technique for
achieving the harmonic filtering objective for loads that
support unidirectional power flow from input to load.
REFERENCES
[1] Souvik Chattopadhyay and V. Ramanarayanan ,
“Digital implementation of a line current shaping algorithm
for three phase high power factor Boost rectifier without
input voltage sensing,” in APEC’01 , pp.592-600.
8(g)
8(h)
8. Experimental results of a Shunt Active Filter system Phase Angle Balance Control Method II (a) Diode rectifier
- ch1: input voltage (v ga ) , ch2: input current (i ga ), ch3:
load current (i la ) , ch4: active filter current (i ha ) (b) Diode
and IM(no load) - ch1: input voltage, ch2: input current ,
ch3: load current, ch4:active filter current (c) Diode
rectifier - ch1: input current, ch2: load current , ch3:active
filter current ch4: sin(w s ) (from DAC) (d) Diode plus
IM(no load) - ch1: input current, ch2: load current ,
ch3:active filter current ch4: sin(w s )(e) Diode rectifier ch1: input current, ch2: load current , ch3: d-axis load
currenti l1_pd (from DAC) ch4: q-axis load current (f) Diode
plus IM (no load) - ch1: input current, ch2: load current ,
ch3: d-axis load current ch4: q-axis load current (g) Diode
rectifier - ch1: input current, ch2: load current , ch3 :
cos(h) (from DAC), ch4 : :sin(h) (h) Diode plus IM (no
load) - ch1: input voltage , ch2:input current , ch3: phase
Â
shifted input current i ga (from DAC),ch4: cos(h)
Scale: :i ga , i la - 25A/div, i ha - 10A/div, v ga - 170V/div,
Â
sin w s , cos(h), sin(h)- 1/div , i l1_pd , i l1_pq , i ga - 22.5A/div
[2] H.Akagi, Y. Kanazawa and A. Nabae, “Instantaneous
reactive power compensators comprising switching devices
without energy storage components”, IEEE Trans. on
Industry Applications, Vol. IA-20,No. 3, May/June 1984.
[3] S. Bhattacharya, D. M. Divan and B. Banerjee,
“Synchronous reference harmonic isolator using active
series filter”, EPE Conf. Record, 1991, Florence, Italy.
[4] V. Soares, P. Verdelho and G.D. Marques, “ An
instantaneous active and reactive current component
method for active filters”, IEEE Trans. on Power
Electronics, Vol. 15 Issue 4 , July 2000, pp. 660-669.
[5] K.M. Smedley, L. Zhou, and C. Qjao “Unified constant
-frequency integration control of active power filters steady state and dynamics” , IEEE Trans. on Power
Electronics, Vol. 16, No. 3, May 2001, pp. 428-436.
[6] TMS320C24x DSP Controllers Peripheral Library and
Specific Devices - Reference Set - Volume 2, Literature
Number : SPRU161B December 1997.
VI. CONCLUSION
In this paper phase angle balance (PAB) control for
filtering the harmonic components of the non-linear load is
proposed for shunt active filter. The harmonic filtering
objective is defined as the task of balancing the phase angle
of the input current with the phase angle of the line
frequency component of the load current. To achieve this
objective the input current is sensed, phase shifted by a
specific amount and then made proportional to the input
voltage.Two methods are described in this paper for
determination of the required phase shift. The current mode
control algorithm is input voltage sensorless , without PLL
and suitable for digital implementation with currents being
sampled only once in a switching period. Phase Angle
Balance control has the flexibility to compensate only for
the harmonics of the load current so that the current rating
requirement of the converter can be made lower than the
converter with both reactive and harmonics compensation.
However if required this method can compensate for the
harmonic as well as the reactive contents of the load and
thereby shape the input current like input voltage. In
conclusion it can be said that phase angle balance (PAB)
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