High speed analog-to-digital conversion with silicon
photonics
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Citation
Holzwarth, C. W. et al. “High speed analog-to-digital conversion
with silicon photonics.” Silicon Photonics IV. Ed. Joel A. Kubby &
Graham T. Reed. San Jose, CA, USA: SPIE, 2009. 72200B-15.
© 2009 SPIE--The International Society for Optical Engineering
As Published
http://dx.doi.org/10.1117/12.808952
Publisher
The International Society for Optical Engineering
Version
Final published version
Accessed
Fri May 27 00:16:14 EDT 2016
Citable Link
http://hdl.handle.net/1721.1/52658
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Detailed Terms
Invited Paper
High Speed Analog-to-Digital Conversion with Silicon Photonics
C. W. Holzwarthb,c, R. Amatyaa,b,d, M. Araghchini a,b, J. Birgea,b, H. Byuna,b, J. Chena,b,
M. Dahlema,b, N.A. DiLelloa,d, F. Gana,b, J. L. Hoyta,d, E. P. Ippena,b, F. X. Kärtnera,b,
A. Khiloa,b, J. Kima,b, M. Kima,d, A. Motamedia,b, J. S. Orcutta,b,d, M. Parka,d, M. Perrotta,d,
M. A. Popovića,b, R. J. Rama,b,d, H. I. Smitha,b, and G. R. Zhoub
a
Department of Electrical Engineering and Computer Science, bResearch Laboratory of
Electronics, cDepartment of Material Science and Engineering,
d
Microsystems Technology Laboratory,
Massachusetts Institute of Technology, 77 Massachusetts Avenue, Cambridge, MA 02139
S. J. Spector, T. M. Lyszczarz, M. W. Geis, D. M. Lennon, J. U. Yoon,
M. E. Grein, and R. T. Schulein
Massachusetts Institute of Technology, Lincoln Laboratory, 244 Wood St.,Lexington, MA 02420
S. Frolov, A. Hanjani, and J. Shmulovich
CyOptics, 600 Corporate Court, South Plainfield, NJ 07080, USA
ABSTRACT
Sampling rates of high-performance electronic analog-to-digital converters (ADC) are fundamentally limited by the
timing jitter of the electronic clock. This limit is overcome in photonic ADC’s by taking advantage of the ultra-low
timing jitter of femtosecond lasers. We have developed designs and strategies for a photonic ADC that is capable of
40 GSa/s at a resolution of 8 bits. This system requires a femtosecond laser with a repetition rate of 2 GHz and timing
jitter less than 20 fs. In addition to a femtosecond laser this system calls for the integration of a number of photonic
components including: a broadband modulator, optical filter banks, and photodetectors. Using silicon-on-insulator (SOI)
as the platform we have fabricated these individual components. The silicon optical modulator is based on a MachZehnder interferometer architecture and achieves a VπL of 2 Vcm. The filter banks comprise 40 second-order
microring-resonator filters with a channel spacing of 80 GHz. For the photodetectors we are exploring ion-bombarded
silicon waveguide detectors and germanium films epitaxially grown on silicon utilizing a process that minimizes the
defect density.
Keywords: Electronic photonic integrated circuits, silicon photonics, high index contrast, optical sampling, optical
analog-to-digital conversion
1. INTRODUCTION
Rapid progress in CMOS technology combined with advances in parallel computing architectures has made Teraflop
digital processors a reality. However, the wealth of new system capabilities offered by such processors cannot be fully
exploited due to the limited performance of electronic analog-to-digital converters (ADCs). Performance of ADCs at
high sampling rates is fundamentally limited by the timing jitter of the electronic clocking circuits [1] and the jitter
performance of electronic oscillators, which is currently around 100 fs for state-of-the-art on-chip electronic oscillators.
Fig. 1 illustrates a microwave signal with amplitude V0 and period T0 and the fundamental relationship between signal
resolution ΔV/V0, expressed in number of bits N and timing jitter Δt involved in the sampling process. For a 20 GHz
analog signal, the resolution in number of bits translates into the timing jitter values shown in Table 1 [1]. Current
Silicon Photonics IV, edited by Joel A. Kubby, Graham T. Reed, Proc. of SPIE Vol. 7220,
72200B · © 2009 SPIE · CCC code: 0277-786X/09/$18 · doi: 10.1117/12.808952
Proc. of SPIE Vol. 7220 72200B-1
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electronics can barely achieve 70 fs jitter. Therefore, high resolution, > 6 bit, sampling of microwave signals with
greater than 20 GHz bandwidth is beyond current electronic capabilities.
In this paper, we demonstrate photonic technologies, such as femtosecond lasers, optical integration and multiplexing
enabling the miniaturization of high repetition rate mode-locked lasers, electro-optic conversion, filter and detector
technologies that can be combined to overcome the electronic bottleneck in ADC [2,3,4,5]. Femtosecond lasers provide
a stream of sampling pulses with much reduced timing jitter, when compared to integrated microwave oscillators,
approaching attosecond jitter levels over milliseconds of measurement time [6]. The approach taken here towards a
photonic ADC is based on wavelength division multiplexing to parallelize and demultiplex the stream of sampling
pulses into lower rate channels, that can then be digitized with conventional electronic ADCs [5].
t
icp
Fig. 1. Relationship between achieved signal resolution ΔV/V0,
expressed in number of bits, N, and timing jitter Δt involved in the
sampling process.
Targeted
A/D resolution
Required
sampling jitter
10-bit
4 fs
8-bit
18 fs
6-bit
72 fs
Table 1. Target A/D resolution and required timing jitter
performance of the sampling process for a 20 GHz analog
signal.
.
Although microwave sources with much better timing jitter performance than 70 fs exist, those sources are typically
bulky, because of the necessary high quality microwave cavities involved, which makes them difficult to integrate in a
sampling system. In addition, there is the difficulty of preserving such low jitter during the sampling process itself.
Figs. 2(a) and (b) show the behavior of a microwave signal and optical pulse train, respectively, emitted from an
ensemble of such sources, where the phase of the microwave signal or the pulse position in a mode-locked laser
undergo a random walk due to the fundamental noise sources in the generation process.
In the steady state of a high-quality oscillator, both electrical and
optical cases based on a high-Q resonator, the gain element
compensates for the losses of the cavity including output coupling.
The resonator losses can be characterized by a cavity decay time
τ cav . For a microwave oscillator, the dissipation-fluctuation
Fig. 2. Random walk of (a) the phase in a
microwave oscillator and (b) the pulse position in a
mode-locked laser due to noise sources in the
generation process.
theorem requires that the loss-channels of the cavity couple a noise
energy equivalent to kT into the cavity within one cavity decay
time, otherwise the cavity itself cannot maintain thermal
equilibrium when isolated. An ideal gain element at least doubles
that noise, while compensating for the signal decay due to the
resonator losses. Assuming that this noise energy is due to random
white noise coupling into the cavity, one may easily derive that this
addition of random noise leads to a random walk in phase of the
oscillator or timing jitter of the zero crossings of the microwave
signal with a linear increase in the timing jitter variance given by
Eq. (1). Thus the timing jitter is related to the oscillation period
and the relative noise added to the intracavity energy stored in the
resonator mode, Wmod e , per cavity decay time. In other words, if
one would introduce as much noise into the cavity as there is stored
energy within a cavity decay time, the phase of the signal could be
changed as much as 2π within that time
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T2
d
1 kT
2
< Δt RF
>= 0 2 ⋅
dt
4π Wmod e τ cav
(1)
Therefore, it should not be surprising, that the corresponding timing jitter of an optical pulse train from a mode-locked
laser, which generates a short optical pulse of duration τ and pulse energy W pulse circulating within the optical cavity
(see Fig. 2(b)) obeys Eq.(2) [6,7]
d
π 2τ 2 2 hωc
2
< Δt ML >=
⋅
dt
6 W pulse τ cav
(2)
Of course, real microwave oscillators or mode-locked lasers may have additional noise sources or nonlinear dynamics
that may further increase the timing jitter indicated in Eqs. (1) and (2), but the fundamental scaling should be preserved.
Due to the operation of the laser at optical frequencies, the noise energy added within a cavity decay time is hωc ,
quantum noise, rather than thermal noise scaling with kT. This makes optical amplifiers and sources much more noisy
than their microwave counterparts, because at typical optical frequencies, ωc , corresponding to 1.55 μm, hωc ~ 50kT .
However, for a laser generating τ =100 fs pulses, the timing jitter of the optical source (Eq. (2)) is scaled down by a
factor of 1000 when compared with a 10 GHz microwave oscillator having a period T0 = 100 ps, which easily
overcompensates for the increased noise by more than two orders of magnitude assuming all other parameters stay
constant and similar.
It is this favorable fundamental scaling of the timing jitter of femtosecond lasers with pulse width and cavity finesse and
the possibility of integrating such lasers with superior timing jitter on a chip, that makes photonic ADCs a viable
concept to pursue.
The layout of the photonic ADC pursued at MIT is shown in Fig. 3 and is based on wavelength-demultiplexing to down
scale the overall sampling rate to parallel channels operating at lower rates [5]. A low-jitter pulse train with repetition
rate fR generated by a mode-locked laser passes through a dispersive fiber, such that there is continuous coverage of the
time axis. This chirped pulse train is modulated by a Mach-Zehnder (MZ) modulator whose RF driving voltage V(t) is
the signal to be sampled. The modulator effectively imprints the time dependence of V(t) onto the optical spectrum. The
optical signal is demultiplexed into N channels by a filter bank, so that every pulse is split into N sub-pulses. Each subpulse is detected by a photodetector and digitized by an electronic ADC taking one sample per pulse. This sample
represents the RF signal at time moment tn = τ (ωn), where ωn is the filter center frequency. N samples VADC(tn) are
obtained, which are spaced uniformly across the repetition period TR. This approach allows improvement of the
sampling rate over what is available in electronic ADCs by a factor of N. By using both outputs of the MZ modulator
we can also linearize its transfer function which is otherwise sinusoidal and factor out pulse-to-pulse amplitude
fluctuations [4].
200 ft
200 fs
Bandpass
Bandpass
filter
filter
1. 8T Hz
1.8THz
2km
SMF28
-H
Mode-locked
Laser
VRF
+φ bias
−φ
ModuItor
RF RF
modulator
00 0
00
00...00
ADC
ADC
AX
ADC
Silicon chip
Fig. 3. High-speed, high resolution optical sampling system. An integrated low-jitter femtosecond laser, currently planned on a
separate chip, with repetition rate fR=1/TR is emitting a stream of 100-200 fs pulses that is spectrally dispersed in regular single-mode
fiber. Dispersion is chosen such that the chirped pulses cover the time interval between pulses with a smooth spectrum. The optical
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spectrum is limited by a bandpass filter. The RF-waveform to be sampled is imprinted on the chirped pulse stream via a dual-port
silicon-based electro-optic modulator. The differential optical output is channelized via a dual-WDM-filter bank with precisely-tuned
center frequencies that map each optical frequency component to certain sampling time slots. The signals from each channel
correspond to time interleaved sample sequences are then separately digitized in low-rate high-resolution ADC’s. The silicon
photonics chip may comprise the silicon optical modulator, filter banks, detector arrays, the low-rate electronic ADCs and feedback
circuitry that is necessary to stabilize the optical filter bank.
2. INTEGRATED FEMTOSECOND LASERS
The proposed photonic ADC concept hinges on a compact, if possible, integrated femtosecond laser with low timing
jitter. The approach towards such a device taken here is based on a planar light wave circuit using an Er-doped
waveguide as the amplifier and a III-V semiconductor saturable Bragg reflector (SBR) as the passive mode-locker, see
Fig 4(a). The reason for choosing Er-doped waveguide technology is two-fold. The long upper state lifetime of Erbium,
10 ms, ensures that relative intensity noise of the laser, which may couple to timing jitter, is only relevant in the low
frequency range < 100 kHz. If necessary, fluctuations in that frequency range can be easily combated by feedback
electronics. Second, the host material is glass, where two-photon absorption is absent in contrast to semiconductor
materials, where two-photon absorption and other nonlinear optical effects strongly impact femtosecond pulse
formation at useful average power levels.
Early attempts at passively mode-locked lasers using an erbium-doped waveguide as a gain medium and an SBR as a
mode-locking element were limited to lower repetition rate and picosecond pulse operation [8,9]. Femtosecond
operation of such lasers [10] was only achieved by employing nonlinear polarization rotation modelocking, which is
difficult to integrate especially at high repetition rates. In addition, in all of these cases the cavity comprised free-space
optics in between the SBR and the erbium-doped waveguide. Here, we demonstrate a 394 MHz, self-starting, passively
mode-locked femtosecond laser based on planar silica waveguide technology. The laser generates 440 fs pulses with an
average output power of 1.2 mW for a pump power of 400 mW [11]. The low efficiency is due to currently high internal
losses, which can be minimized in the future.
The schematic laser layout is depicted in Fig. 4(a). The laser cavity consists of a 5 cm section of erbium doped aluminosilicate waveguide with a group-velocity dispersion of 30 fs2/mm. A 20 cm section of phosphorous-doped silica
waveguide with a dispersion of -25 fs2/mm is used to obtain a net anomalous intracavity dispersion to enable soliton
mode-locking [12]. The Er-doped section and the P-doped silica-glass section have an effective mode area of 10 μm2
and 40 μm2, respectively. A loop mirror is used at one end to provide 10% output coupling, while the other end is buttcoupled to an external SBR. The SBR is a commercial device with 14% modulation depth, a 2 ps recovery time, and a
saturation fluence of 25 μJ/cm2. Pump power is provided by an external 980 nm laser diode coupled into the waveguide
chip. The laser was operated with 400 mW of cavity-coupled pump power; the intracavity signal power was measured
to be 12 mW, corresponding to a 30 pJ intracavity pulse energy. The output pulses with an average power of 1.2 mW
are then amplified to 18 mW using an EDFA (980 nm pump, 350 mA), detected using a 10 GHz photodiode, and
measured with a 500 MHz sampling scope and a signal source analyzer (Agilent E5052).
2b
bnwb
roob
1 2O
onbn
1 290
MsAGIGuap
Fig. 4. a) Integrated laser layout. Inset on the left depicts the SBR’s reflection and dispersion spectra. b) Picture of the laser setup.
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S6L 9wb
peoLe swb
1 ao
.
1 2O
1 290
MsAGIGuap
LGd11GUCA
1!WG
CH
22
LGd nGUCA
a2
141 H
1000
1 000
qGI9A
Fig. 5. Measurement traces at 400 MHz: a) normalized optical spectrum before and after amplification, b) Rf spectrum
(3 GHz span, 10 MHz resolution), d) RF spectrum (100 kHz span, 500 Hz resolution), d) 10 second persistence trace, e)
background free autocorrelation trace and f) single side band (SSB) phase noise of the first harmonic of the laser.
Fig. 5 depicts the measurement results. The persistence trace, see Fig. 5(d) shows excellent signal stability, while the RF
spectrum in (c) indicates a side-mode suppression ratio of 80 dB. The 8.4 nm FWHM optical bandwidth before
amplification implies 300 fs duration transform-limited pulses. After amplification, the optical bandwidth decreases to
7.4 nm, corresponding to 340 fs. The autocorrelation measurement yielded a pulse duration of 440 fs. The difference is
attributed to incomplete compensation of the chirp added by the erbium-doped fiber amplifier. The laser was selfstarting; as the pump power is increased, the laser first operates in a mode-locked Q-switching state before transitioning
to a continuous-wave soliton mode-locked state at a pump power of 160 mW. Fig. 5(f) shows the phase noise of the first
harmonic (394 MHz) of the laser. The timing jitter integrated from 20 MHz progressively down to 1 kHz is also shown.
The timing jitter integrated from 10 kHz to 20 MHz is 24 fs, which is close to the noise sensitivity limit of the Agilent
Agilent E5052 signal analyzer. Currently work is in progress to further shorten the pulse duration to the 100-200 fs
range, to post amplify and interleave the pulse train to achieve the necessary output power levels and 2 GHz repetition
rate necessary for the planned photonic ADC. We also expect that the timing jitter can be further reduced to the fewfemtosecond or even sub-femtosecond range [13].
3. SILICON ELECTRO-OPTIC MODULATORS
Another important component of the sampling system is a high-speed silicon optical modulator that transfers the
electronic signal into the optical domain. Fig. 6(a) illustrates the Mach-Zehnder interferometer device architecture used
in these experiments. The interferometer arms contain diode sections (Fig. 6(b)) whose index can be changed by
electronic carrier injection and the resulting plasma effect [14]. The output light intensity is modulated by the phase
shifts induced in each arm of the interferometer by changes in carrier density. This type of modulator can be operated
either as a forward biased PIN diode or as a reverse biased PN diode. In forward biased operation the device is more
sensitive, but under reverse bias the modulator exhibits greater bandwidth [15]. The ADC requires the bandwidth of
reverse bias operation, and we concentrate on those results here.
The modulators were fabricated with a standard CMOS-compatible lithography process. The sidewalls of the
waveguide are moderately doped to a concentration of 1018 cm-3, and the center of the waveguide is doped n-type to a
lighter concentration of 2x1017 cm-3. These doping concentrations were chosen to produce a diode that would
simultaneously cause a reasonable phase shift without excessive optical loss [16]. When operated in reverse bias, a
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depletion region forms at the junction at one side of the waveguide. As the reverse bias voltage is increased, this
depletion region becomes larger and extends into the center of the waveguide. A highly doped (1019 cm-3), 50-nm-thick
silicon layer connects the waveguide to electrical contacts located 1 μm from the waveguide. Even higher doping
concentrations of 1021 cm-3 are used under the metal contacts to assure good ohmic contacts.
Silicon Phase Shifters
N+
P+
N+
P+
DC
Adiabatic
Ch. (1)
3 dB coupler
Ch. (2)
DC/RF
GND
(a)
GND/DC
RF/GND
0.5
Oxide
P
0.05
P++
n
N
P+
0.17
N+
N++
Silicon
Oxide
3.0
1.0
0.05
(b)
Fig. 6. Mach-Zehnder modulator. (a) Interferometer shown in plan view. (b) Cross section of diode structure.
The silicon rib in the center of the micrograph functions as the optical waveguide and the diode. A thin silicon
layer connects the metal contacts to the diode. This arrangement isolates the optical mode from the metal
contacts and minimizes the optical loss.
By employing a relatively short interaction length of 0.5 mm (the travel time of the light is only 5 ps), a simple lumpedelement electrode design can be used to achieve speeds of about 30 GHz. The modulator can be operated in a push-pull
configuration by driving the center electrode (as shown), or a single arm can be driven. An additional thermal phase
shifter (not shown) is fabricated on one arm, to allow the relative optical phase of the two arms of the modulator to be
adjusted independently of the diodes. An adiabatic 4-port 3 dB coupler [17] is used to interfere the two arms at the
output of the Mach-Zehnder interferometer leading to two complementary modulator output ports. To provide efficient
fiber-to-chip coupling, a silicon inverse-taper mode converter combined with a lower index silicon oxynitride
waveguide [18] was used.
To model the carrier concentration in the waveguide, numerical simulations were performed by solving Poisson and
carrier continuity equations. The modeled change in carrier concentration determines a change in the refractive index of
silicon which can then be used to determine the change in the modal index, Neff. The change in the modal index can
then be used to determine the phase shift for a given diode length.
The DC response of the modulator was tested by varying the DC bias voltage on arm #1 while the other arm remained
at the same bias level. Fig. 7 shows the simulated and measured normalized light intensity as a function of the applied
voltage, for the two complementary outputs. The measured and simulated results are in good agreement with each
other. To get a better measure of VπL, a longer device, with 5 mm arm lengths, was also measured. The output of that
modulator (which was fabricated with only one output channel) is also shown in Fig. 7, and gives a VπL of just over
4 Vcm. This is comparable to what had been achieved earlier with a structure requiring more complicated fabrication
[19]. Operating the modulator in a push-pull configuration has also been demonstrated, and cuts the effective VπL for
the device in half to 2 Vcm, as expected.
Fig. 8 shows the small signal frequency response of the modulator after correcting for frequency dependent cable losses.
A 3-dB cut-off frequency of 26 GHz is reached in a 0.5 mm device, which agrees with the estimate calculated above. A
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1.2 5mm
long
Snini long
devicedevice
- --- Simulated Chi (0.5mm)
-Measured Chi (0.5mm)
-Measured Ch2 (0.5mm)
0
- --- Simulated Ch2 (0.5mm)
0.8
-Measured (5mm)
--
Ch.(l(
device
0.5mm 0.5mm
longlong
device
0.4
Normalized Optical Power (dB)
Normalized Intensity (a.u)
shorter device with 0.25 mm long phase shifters was also measured, and exhibits between 30-40 GHz of bandwidth. At
high frequencies, the electrode can no longer be considered a lumped-element device, leading to irregularities such as
the secondary peak at 40 GHz in the 0.5 mm device. The sensitivity of the device was measured by operating the
0.5 mm device in a push-pull configuration. In this configuration, 46 mW of RF power achieves ~22% modulation
depth, (where the modulation depth is defined in [20]), for a 26 GHz sinusoid. In an optically sampled ADC, this
modulation depth is sufficient to achieve a signal to noise ratio of 47 dB [20]. This is only 6 dB less than the theoretical
maximum of 53 dB achieved with a 50% modulation depth under otherwise similar conditions.
Oh. (2)
0.25 mm
0.5 mm
-4
-8
0.0
0
4
8
12
Applied Voltage (V)
16
0.1
Fig. 7. Measured and simulated normalized light transmission
as a function of applied voltage.
1
10
Frequency (GHz)
100
Fig. 8. Measured frequency response of the modulator.
.
4. HIGH INDEX CONTRAST FILTER BANK
After the incoming electric signal has been imprinted onto the femtosecond-laser pulse by the MZ modulator, the signal
traveling out of each output arm must be demultiplexed. Filter banks consisting of microring resonators, utilizing highindex contrast materials, are ideal for this function. High-index-contrast microring resonators have been shown to
achieve the large free-spectral range (FSR) and low loss required for these filter banks, while maintaining wavelength
scale dimensions required for chip-level integration [21,22]. The main challenge in the fabrication of these filter banks
is achieving the dimensional precision, on the tens of picometer scale, required to control the resonant frequency
spacing between each filter of the filter bank. The proposed ADC design requires the output from each arm of the MZ
modulator to pass through identical twenty-channel filter banks comprised of second-order microring resonator filters.
The microring resonators are designed to have 2 THz FSR and 25 GHz bandwidth, allowing for a channel spacing of
80 GHz with less than 30 dB of crosstalk.
Previously, we reported the results of a twenty-channel dual filter bank that used silicon-rich silicon nitride (SiNx),
n =2.2 at 1550 nm, as the core material (Fig. 9) [23]. The resonant frequency spacing of this filter bank was controlled
by changing both the ring radius and average waveguide width of each microring resonator by the appropriate amount.
This required changing the average waveguide width on a scale smaller than the grid size of the scanning-electron-beam
lithography (SEBL) tool, with a precision of tens of picometers. This was achieved by modulating the electron-beam
dose used to write each filter to change the average waveguide width on a scale much finer then the SEBL grid size
[23]. The average channel spacing of the filter bank was measured to be 83 GHz with a standard deviation of 8 GHz.
This 3 GHz error in average spacing corresponds to dimensional precision of the average waveguide width of 75 pm,
much smaller than the 6 nm step size of the SEBL address grid.
Looking at the frequency response of the filter bank in Fig. 9 it is seen that there is a rather large drop loss (~12 dB on
average) and the bandwidth of the filters is much larger than the target 25 GHz. The two reasons for this are that there
is an average frequency mismatch of 29 GHz between the two rings of the same filter and a higher than expected
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propagation loss in the SiNx. The frequency mismatch causes both increased drop loss and widening of the filters
bandwidth. This can be easily corrected though postfabrication thermal trimming using integrated microheaters. Fig.
10 demonstrates how a microheater can be used to reduce this mismatch to less than 1 GHz in a two-channel filter bank.
After thermal trimming the frequency response for the two filters shown are nearly identical. Likewise, if both rings of
the filter are heated simultaneously it is possible to shift the resonant frequency response of the filter, with an efficiency
of 80μW/GHz/filter, to correct for any frequency spacing errors [24]. As shown in Fig. 10 even after correcting for the
frequency mismatch there is still ~8 dB of loss in each filter. This loss is due to material absorption in the SiNx core.
The SiNx used is a non-stoichiometric material optimized for very low stress. The optical properties of this material,
including material absorption, varies from batch to batch. One way to solve this problem is to switch to a SOI platform,
the single crystal silicon layer used for the filters will not vary in its properties and is known to have low optical
absorption.
Normalized Power (dB)
0
thru ports
-10
drop
ports
-20
-30
-40
-1200
-800
-400
0
400
Relative Frequency (GHz)
800
Fig. 9. Measured frequency response for a dual twenty-channel filer bank with all non-filter related losses factored out.
Transmission Power (dB)
0
before
after
-10
-20
-30
-40
-200
-100
0
100 200
Relative Frequency (GHz)
300
Fig. 10. Transmission response of a two channel filter bank before and after thermal trimming.
To make the change from SiNx to Si both the filter design and fabrication process must be altered. The Si filter design
chosen uses a very low aspect ratio, 105 X 600 nm and 105 X 500 nm, for the ring and bus waveguide, respectively,
instead of the typical Si single mode waveguide dimensions of ~220 X ~500 nm. This waveguide design reduces the
filter’s frequency sensitivity to width to 40 GHz/nm, from the high sensitivity of ~100 GHz/nm. This makes the Si
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filter design only slightly more sensitive to changes in width than the SiNx design despite the much higher index
contrast. In addition to the changes in design, the basic fabrication process has to be changed because the process for
the SiNx filters uses a Ni hardmask, formed via a PMMA lift-off process, during the reactive-ion etching (RIE) step. It
is found that using metal hardmasks during RIE of Si can lead to a silicide formation on the waveguide sidewalls,
causing excess propagation loss [25]. The new process uses hydrogen silsequioxane (HSQ), a negative electron-beam
resist, as both the resist and the hardmask. HSQ transforms into a SiO2 like hardmask after being exposed by an
electron-beam and developed in TMAH. The HSQ pattern is then transferred into the Si using RIE with HBr gas, which
has a very high selectivity between SiO2 and Si. After etching, the HSQ is stripped with a dilute HF dip. The devices
are then cleaned with an RCA clean and then overclad with 1 μm of HSQ that is annealed at 400ºC for 1 hr in an
oxygen atmosphere to turn it into SiO2 (Fig. 11(a)) [26]. Titanium resistive heaters are then fabricated above the HSQ
using contact lithography and a lift-off method. The propagation loss of these Si waveguides is measured to be
2.5 dB/cm over the wavelength range of 1520-1610 nm.
HSQ
500 nm
105 nm
342 nm
500 nm
708 nm
Si
600 nm
SiO2
(a)
342 nm
Transmission Power (dB)
0
measured
fitted
-10
-20
-30
-40
-50
-80
0
80
Relative Frequency (GHz)
(b)
(c)
Fig. 11. (a) Cross-section and (b) Top-view scanning-electron micrograph of Si waveguide and second-order filter. (c) Measured
and fitted filter response of a second-order silicon filter.
This low loss in the Si waveguides makes it possible to fabricate the small bandwidth, high FSR filters that are required
for the filter bank while maintaining minimal drop loss. Fig. 11(b) shows the fabricated Si filter design before being
overclad with HSQ. The measured and fitted transmission response for this filter design is shown in Fig. 11(c). The
drop loss of this filter is less than 0.5 dB; much lower than the 8 dB that was measured in the SiNx filters. The fitted
bandwidth is measured to be 21 GHz, only slightly smaller than the designed bandwidth of 25 GHz. The frequency
mismatch between rings in this filter is fitted to be less than 0.25 GHz and the crosstalk between neighboring channels
80 GHz away is less than -30 dB. This filter response meets all the requirements for what is needed for the ADC’s
twenty-channel filter bank.
Since the change to Si required changing electron-beam resists from a positive resist, PMMA, to a negative resist, HSQ,
empirical experiments were performed to find the new frequency dependence on electron-beam dose. These
experiments were performed by writing both a reference filter and a calibration filter that share the same through port
for each variation of dose. The dose used for the reference filter is kept constant while the dose used for the calibration
filters is varied. The frequency dependency on dose is then found by comparing the relative frequency shift between
each reference and calibration filter [23]. From these experiments it is found that the frequency dependence increases
from 14 GHz/% for the SiNx filters to 66 GHz/% for the Si filters. This increase is more than can be explained by the
difference in frequency sensitivity for the two designs. The extra sensitivity is most likely explained by the difference
in contrast of the two e-beam resists. It is also seen that the standard deviation of the fitted frequency calibration data of
the Si filters is approximately a factor of 10 larger than for the SiNx filters. This is not a big concern since the thermal
trimming efficiency of the Si filter is about a factor of 10 more efficient. This means the total power needed to correct
for frequency errors in a Si filter bank should be about the same as for the SiNx filter banks.
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As mentioned earlier the proposed ADC needs two identical filter banks, one for each output arm of the MZ modulator.
One way to insure that the two filter banks are identical is to use the same filter bank for both arms. This can be done
since one ring resonator supports two optical modes with orthogonal propagation directions, clockwise and
counterclockwise. As long as the sidewall roughness of the ring waveguide is low enough there will be no significant
coupling between the two modes [27]. To check the possibility of using this method a single ring filter was fabricated
with the layout shown in Fig. 12(a). The frequency response for the propagating modes as well as the crosstalk of this
filter was measured (Fig. 12(b)). The transmission responses of both modes are exactly the same as is expected.
Looking at the crosstalk at first it seems that there is a large amount of crosstalk between the two counter-propagating
modes. Looking more carefully it is seen that this “crosstalk” is actually the sum of both a through and drop response.
There are two ways that this can happen either the power that is not dropped by the filters bounces off the end-facet and
then travels back towards the filter and is dropped or the power that is dropped bounces of the end-facet travels back to
the filters and hits a through response. Both of these possibilities can be eliminated by reducing the reflections at the
end-facets. To put an upper limit on the crosstalk in the filter the drop response is subtracted from the crosstalk giving a
through response. The dip at the resonant frequency of the through response is below -30 dB showing that the crosstalk
is at least less than -30 dB. This shows that it is possible to use one filter bank (Fig. 12(c)) to demultiplex both arms of
the MZ modulator as long as great care is taken to prevent any reflections in the system.
(a)
Input A
Input B
(b)
-20
Drop A
C
Drop A
Drop B
Drop (B-A)
Drop B
(c)
C)
e4
5
Transmission Power
0
0
-40
100 μm
-60
-500
0
500
Relative Frequency (GHz)
Fig. 12. (a) Layout and (b) transmission response of first-order microring resonator demonstrating the counterpropagating mode technique. (c) Fabricated twenty-channel counter-propagating-mode silicon filter bank.
5. SILICON AND GERMANIUM INFRARED PHOTODIODES
Both germanium and silicon photodiodes are being pursued for this system. The Ge diode technology is well
established, but requires careful attention to process integration issues when used in a silicon CMOS process. In pursuit
of easier process integration, we have developed a novel all-silicon infrared diode device.
In 1959 Fan [28] reported that radiation damaged Si would produce a photocurrent when illuminated with sub bandgap
radiation. Knight [29,30] and others [31,32,33] have used this result to make waveguide optical detectors. Unlike most
other detectors formed in Ge or InGaAs where the optical absorption is > 1000 dB cm-1, the Si waveguide photodiodes
used for the ADC applications discussed here exhibit only ~20 dB cm-1 absorption. A fact that needs special attention.
Here, we summarize the recently achieved device performance; the fabrication, optical absorption, and quantum
efficiency are given in more detail elsewhere [33,34].
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The waveguide photodiodes are of similar structure as shown in Fig. 6, the RC time constant for a 50 Ω load is
~60 ps cm-1, while the optical transient time determined by the group velocity of light is 130 ps cm-1. Thus the optical
absorption of the waveguide detectors directly impacts the tradeoff between the optical response and the electrical
bandwidth due to the necessary device length. As the waveguide length is increased a larger fraction of the incoming
light is absorbed and converted into an electrical signal. However, the increased propagation time of the light through
the longer device reduces the electrical bandwidth. Fig. 13 shows the trade off between the fraction of light absorbed in
the detector and the transient time limited bandwidth for several absorption coefficients. What fraction of the absorbed
light actually generates a photocurrent or is lost to some other mechanism does not effect this bandwidth absorbed light
tradeoff. A traveling wave photodiode structure would have a larger bandwidth, but only half of the photo response. To
absorb half of the incoming light a waveguide-photodiode-length of 1.5 mm is required, which limits the bandwidth to
22 GHz. Although only half of the incoming light is absorbed in the diode, the internal photoresponse of ~ 0.4 to
10 A W-1 implies an external quantum efficiency of 0.2 to 5 A W-1 is achieved. Even if the waveguide length is
increased to 5 mm, which will absorb 90% of the incoming light, the transit-time bandwidth is ~ 10 GHz, an acceptable
value for our ADC application.
Bandwidth (GHz)
100
-1
100 dB cm
40 dB cm
-1
20 dB cm-1
10
10 dB cm
-1
5 dB cm-1
1
0
0.2
0.4
0.6
0.8
1
Fraction of Light Absorbed
Fig. 13. Calculated maximum 3 dB RF power bandwidth as limited by light propagation time though the
waveguide photodetector as a function of the fraction of the light absorbed for several optical absorption
coefficients. An optical group velocity of 7.5x109 cm s-1 was assumed for these calculated results.
0
Power [S21]2 (dB)
To determine the intrinsic photodiode bandwidth, the
frequency response of a 100-μm-long waveguide
photodiode was measured. This short diode absorbs
only 4.5% of the incoming light but the 100 μm length
implies a propagation-time-limited bandwidth of > 200
GHz. The frequency response of the diode was
determined using a measurement system containing a
LiNbO3 Mach-Zehnder optical modulator with a
bandwidth of ~34 GHz and a commercial InGaAs
photodiode with a bandwidth > 50 GHz, used as a
reference. The response of the measurement system
with the Si diode and the standard diode are shown in
Fig. 14. The normalized photodiode response was
obtained by taking the ratio between the measured
response of the Si and the InGaAs-reference detectors
and is shown in Fig. 15. This procedure eliminates the
response of the modulator, RF losses in the cables, and
the bias tee. However the response of the RF probes
used to contact the Si waveguide photodiode cannot be
compensated for. The Si-diode response agrees with the
--Si photodiode, 10 V bias
.InGaAs diode
-10
-20
-30
40
60
20
Frequency (GHz)
Fig. 14. Frequency response of the measurement system with a
100-μm-long waveguide Si photodiode and the commercial
InGaAs photodiode standard. Both curves were set to 0 dB at
1 GHz. The commercial diode has a bandwidth > 50 GHz. The
response decreases with frequency as a result of the limited
bandwidth of the LiNbO3 MZ modulator and cabling losses.
0
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InGaAs-reference within a few dB up to 30 GHz and
stays close even up to 40 and 50 GHz. The calculated
RC-time-constant bandwidth of the contact pad
capacitance and the 50 Ω input impedance of the
network analyzer is ~ 100 GHz. Also a ~90 GHz
bandwidth was calculated considering only the transit
time of holes and electrons at their saturated velocity of
1x107 cm s-1. The RF power to the optical modulator
was increased with operating frequency to compensate
for reduced response. Measurements were made at
different RF power levels to ensure that the modulator
and photodiodes were operating in their linear region.
Power [S21]2 (dB)
0
10
55
00
-5
-10
0.01
1
10
0.1
Frequency (GHz)
100
One challenge of a wideband analog system is that one
must ensure that the second-order distortions are small,
Fig. 15. Measured frequency response of a 100-μm-long
not just the third-order. This is because, by definition, a
waveguide Si photodiode using the difference between a high
wideband system has enough bandwidth for octaves of
bandwidth standard diode and the Si diode. The data for both
the measured signal to be present in the measurement
diodes individually is shown in Fig. 14.
The increase in
bandwidth which will corrupt the signal estimation. We
response between 40 and 50 GHz is believed to result from RF
measured the nonlinear response of the silicon
probes used to contact the Si diode, which are rated to 40 GHz.
waveguide ion-bombarded photodiodes using a threeMore details of this measurement are given elsewhere [34].
tone measurement setup [35] in which three independent
DFB lasers were modulated at f1 (380 MHz), f2 (620 MHz), and f3 (2.000003 GHz). A useful feature of this technique
is that, by judicious choice of frequencies, the second-order nonlinear distortion at f1 + f2 can be placed very close to the
third-order nonlinear distortion f3-(f1+f2) and a narrowband RF filter will pass the distortion components to the
measurement system while removing the strong fundamental components—this is very challenging using only a twotone measurement setup. More importantly, this technique does not rely on any frequency components of type N·f ±M·f
where N and M are integers > 1.
Fig. 16 shows the power of the fundamental compared to the second-and third-order distortion products as a function of
photocurrent for two different bias voltages. The second and third harmonic distortion increased with increasing bias
voltage, and the second harmonic distortion is observed to be more severe than the third harmonic.
40
Output Power (dBm)
In addition to Si photodetectors, germanium
photodiodes are under investigation. Blanket Ge-on-Si
films have a relatively high threading dislocation
density due to the 4% lattice mismatch between Ge and
Si. To reduce the threading dislocation density, Ge can
be grown selectively in oxide openings. For the test
devices in this work, the starting material was 6” p+ Si
wafers with patterned silicon dioxide films. Using an
Applied Materials low-pressure chemical vapor
deposition epitaxial reactor, the germanium was grown
to overfill the openings to ensure that the oxide
openings were completely filled.
A chemicalmechanical polishing step was then performed to
planarize the Ge film. The samples were then
implanted with phosphorus and subjected to a rapid
thermal anneal at 600°C for 5 s to activate the implant.
Contact vias were etched and metal contacts were
deposited to form vertical pin diodes. A crosssectional schematic of the device is shown in the inset
of Fig. 18.
-10
Fundamental
fundamental
second
harmonic
Second
Harmonic
-60
-110
0.01
Third
third
Harmonic
harmonic
0.1
Photocurrent (mA)
1
Fig. 16. Output power of the photocurrent (in a 50 Ohm load)
as a function of photocurrent for the fundamental, second
harmonic (formed by f1+f2), and third harmonic (formed by f3(f1+f2) for 10 V (black) and 12.5 V (red) reverse bias voltage in
a 1-mm Si waveguide ion-bombarded photodetector.
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Fig. 17 shows plan view transmission electron microscopy (TEM) images for various sized Ge features. For the asgrown Ge layers, the threading dislocation density decreases with device width from 5.5 x 108 cm-2 to 2.6 x 108 cm-2 to
1.7 x 108 cm-2 for 5, 1, and 0.65 μm-wide devices respectively. This effect is consistent with previous results on aspect
ratio trapping [36] and a small area effect [37, 38]. Fig. 16 also illustrates that a cyclic anneal reduces the dislocation
density. Analysis of multiple plan view TEM areas indicates that the dislocation density was reduced from
1.7 x 108 cm-2 to 2.6 x 107 cm-2 in the 0.65 μm-wide islands by cyclic annealing.
I urn
5um
O.65um
0.65 um
(a)
(b)
(d)
(c)
Fig. 17. Plan-view TEM images of different sized germanium islands for (a – c) as-grown and (d) cyclically
annealed 0.8 μm-thick Ge grown selectively on Si . The threading dislocation density decreases with the width
of the island. The threading dislocation density also decreases when the film is annealed.
Fewer threading dislocations are expected to improve the yield and reduce the dark current of these devices. Fig. 18
shows current versus voltage characteristics of vertical pin diodes fabricated in 0.35 x 10 μm germanium islands for
both an as-grown Ge film and a film cyclically annealed between 830°C and 450°C. These are representative
characteristics of rectifying devices. After processing, it was found that about 50% of the devices displayed rectifying
behavior and the others were shorted. The impact of material quality and device processing on yield requires further
investigation. The dark current for both devices is ~0.5 nA at -1 V. While the cyclic anneal reduced the threading
dislocation density in these devices, this did not have a significant effect on the dark current. This indicates that the
leakage in these smaller devices is dominated by another mechanism and more work is needed to fully characterize it.
The cyclic anneal did have an effect on the on current, however. The device that received the anneal has ~2x higher on
current at 1 V than the device without the anneal. This is most likely due to a reduction in the series resistance.
-4
10
- n+Ge
-6
Current (A)
10
Metal
.* Si
-8
10
10
10
-10
-UnanneaIe
-Annealed
-12
-1
-0.5
0
0.5
Voltage (V)
1
Fig. 18. Measured current versus voltage characteristics for pin diodes in a Ge film that was annealed and in a
film that was not annealed. The cyclic anneal takes place after germanium growth but before n+ ion
implantation. The anneal has little effect on the dark current of these devices but increases the on current by ~2x.
The germanium island size is 0.35 x 10 μm. Inset: a cross-sectional schematic diagram of the diodes.
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6. CONCLUSIONS
We have presented the system architecture and fabricated key elements of a photonic ADC. The fabricated devices
include: a 394 MHz waveguide femtosecond laser with 24 fs timing jitter, a Si modulator capable of modulating speeds
faster then 25 GHz, high performance Si-microring filters for a demultiplexing filter bank, silicon and germanium
photodiodes. Using such devices for the proposed ADC system allows higher sampling speeds than current pure
electronic systems due to the low timing jitter of the femotsecond laser. The size-scale, materials, and processing
methods for these devices make them suitable for integration on an electronic-photonic silicon-on-insulator platform.
ACKNOWLEDGEMENTS
The MIT Lincoln Laboratory portion of this work was sponsored by the EPIC Program of the Defense Advanced
Research Projects Agency and in part by the Department of the Air Force under Air Force Contract FA8721-05-C-0002.
The MIT Campus portion of this work was sponsored by the DARPA EPIC Program under contract W911NF-04-10431 and DARPA OAWG Program under contract HR0011-05-C-0155. Opinions, interpretations, conclusions, and
recommendations are those of the authors, and do not necessarily represent the view of the United States Government.
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