IF Diversity Receiver
AD6655
Data Sheet
FEATURES
APPLICATIONS
SNR = 74.5 dBc (75.5 dBFS) in a 32.7 MHz BW at
70 MHz at 150 MSPS
SFDR = 80 dBc to 70 MHz at 150 MSPS
1.8 V analog supply operation
1.8 V to 3.3 V CMOS output supply or 1.8 V LVDS
output supply
Integer 1-to-8 input clock divider
Integrated dual-channel ADC
Sample rates up to 150 MSPS
IF sampling frequencies to 450 MHz
Internal ADC voltage reference
Integrated ADC sample-and-hold inputs
Flexible analog input range: 1 V p-p to 2 V p-p
ADC clock duty cycle stabilizer
95 dB channel isolation/crosstalk
Integrated wideband digital downconverter (DDC)
32-bit complex, numerically controlled oscillator (NCO)
Decimating half-band filter and FIR filter
Supports real and complex output modes
Fast attack/threshold detect bits
Composite signal monitor
Energy-saving power-down modes
Communications
Diversity radio systems
Multimode digital receivers (3G)
TD-SCDMA, WiMax, WCDMA,
CDMA2000, GSM, EDGE, LTE
I/Q demodulation systems
Smart antenna systems
General-purpose software radios
Broadband data applications
PRODUCT HIGHLIGHTS
1. Integrated dual, 14-bit, 150 MSPS ADC.
2. Integrated wideband decimation filter and 32-bit
complex NCO.
3. Fast overrange detect and signal monitor with serial output.
4. Proprietary differential input maintains excellent SNR
performance for input frequencies up to 450 MHz.
5. Flexible output modes, including independent CMOS,
interleaved CMOS, IQ mode CMOS, and interleaved LVDS.
6. SYNC input allows synchronization of multiple devices.
7. 3-bit SPI port for register programming and register readback.
FUNCTIONAL BLOCK DIAGRAM
FD[0:3]A
DVDD
DRVDD
FD BITS/THRESHOLD
DETECT
CMOS/LVDS
OUTPUT BUFFER
AD6655
I
SHA
ADC
VIN–A
Q
LP/HP
DECIMATING
HB FILTER +
FIR
VREF
SENSE
CML
RBIAS
SIGNAL
MONITOR
CLK–
fADC /8
NCO
REF
SELECT
DUTY
CYCLE
STABILIZER
DCO
GENERATION
Q
VIN–B
SHA
ADC
VIN+B
I
AGND
MULTI-CHIP
SYNC
FD BITS/THRESHOLD
DETECT
SYNC
FD[0:3]B
LP/HP
DECIMATING
HB FILTER +
FIR
SIGNAL MONITOR
DATA
PROGRAMMING DATA
SIGNAL MONITOR
INTERFACE
SPI
SMI
SMI
SMI
SDFS SCLK/ SDO/
PDWN OEB
SDIO/ SCLK/ CSB
DCS DFS
NOTES
1. PIN NAMES ARE FOR THE CMOS PIN CONFIGURATION ONLY; SEE FIGURE 10 FOR LVDS PIN NAMES.
D0A
CLK+
DIVIDE 1
TO 8
32-BIT
TUNING
NCO
D13A
DCOA
DCOB
D13B
D0B
DRGND
06709-001
VIN+A
CMOS
OUTPUT BUFFER
AVDD
Figure 1.
Rev. B
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Technical Support
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AD6655
Data Sheet
TABLE OF CONTENTS
Features .............................................................................................. 1
Decimating Half-Band Filter and FIR filter ................................ 39
Applications ....................................................................................... 1
Half-Band Filter Coefficients.................................................... 39
Product Highlights ........................................................................... 1
Half-Band Filter Features .......................................................... 39
Functional Block Diagram .............................................................. 1
Fixed-Coefficient FIR Filter ...................................................... 39
Revision History ............................................................................... 3
Synchronization .......................................................................... 40
General Description ......................................................................... 4
Combined Filter Performance .................................................. 40
Specifications..................................................................................... 5
Final NCO ................................................................................... 40
ADC DC Specifications—AD6655-80/AD6655-105 .............. 5
ADC Overrange and Gain Control .............................................. 41
ADC DC Specifications—AD6655-125/AD6655-150 ............ 6
Fast Detect Overview ................................................................. 41
ADC AC Specifications—AD6655-80/AD6655-105 ............... 7
ADC Fast Magnitude ................................................................. 41
ADC AC Specifications—AD6655-125/AD6655-150............. 8
ADC Overrange (OR)................................................................ 42
Digital Specifications—AD6655-80/AD6655-105 ................... 9
Gain Switching ............................................................................ 42
Digital Specifications—AD6655-125/AD6655-150 ............... 11
Signal Monitor ................................................................................ 44
Switching Specifications—AD6655-80/AD6655-105............ 13
Peak Detector Mode................................................................... 44
Switching Specifications—AD6655-125/AD6655-150 ......... 14
RMS/MS Magnitude Mode ....................................................... 44
Timing Specifications ................................................................ 15
Threshold Crossing Mode ......................................................... 45
Absolute Maximum Ratings.......................................................... 18
Additional Control Bits ............................................................. 45
Thermal Characteristics ............................................................ 18
DC Correction ............................................................................ 45
ESD Caution ................................................................................ 18
Signal Monitor SPORT Output ................................................ 46
Pin Configurations and Function Descriptions ......................... 19
Channel/Chip Synchronization .................................................... 47
Equivalent Circuits ......................................................................... 23
Serial Port Interface (SPI) .............................................................. 48
Typical Performance Characteristics ........................................... 24
Configuration Using the SPI ..................................................... 48
Theory of Operation ...................................................................... 29
Hardware Interface..................................................................... 48
ADC Architecture ...................................................................... 29
Configuration Without the SPI ................................................ 49
Analog Input Considerations.................................................... 29
SPI Accessible Features .............................................................. 49
Voltage Reference ....................................................................... 31
Memory Map .................................................................................. 50
Clock Input Considerations ...................................................... 32
Reading the Memory Map Register Table............................... 50
Power Dissipation and Standby Mode ..................................... 34
Memory Map Register Table ..................................................... 51
Digital Outputs ........................................................................... 35
Memory Map Register Description ......................................... 55
Digital Downconverter .................................................................. 37
Applications Information .............................................................. 59
Downconverter Modes .............................................................. 37
Design Guidelines ...................................................................... 59
Numerically Controlled Oscillator (NCO) ............................. 37
Evaluation Board ............................................................................ 61
Half-Band Decimating Filter and FIR Filter ........................... 37
Power Supplies ............................................................................ 61
fADC/8 Fixed-Frequency NCO ................................................... 37
Input Signals................................................................................ 61
Numerically Controlled Oscillator (NCO) ................................. 38
Output Signals ............................................................................ 61
Frequency Translation ............................................................... 38
Default Operation and Jumper Selection Settings ................. 62
NCO Synchronization ............................................................... 38
Alternative Clock Configurations ............................................ 62
Phase Offset ................................................................................. 38
Alternative Analog Input Drive Configuration...................... 63
NCO Amplitude and Phase Dither .......................................... 38
Schematics ................................................................................... 64
Rev. B | Page 2 of 84
Data Sheet
AD6655
Evaluation Board Layouts ..........................................................74
Ordering Guide ........................................................................... 84
Bill of Materials ...........................................................................82
Outline Dimensions ........................................................................84
REVISION HISTORY
1/14—Rev. A to Rev. B
Removed CP-63-3 Package ............................................... Universal
Changes to Address 0x0D, Table 29 ..............................................52
Changes to Memory Map Description Section ...........................55
Updated Outline Dimensions ........................................................84
9/09—Rev. A to Rev. 0
Added Exposed Pad Notation to Figure 9 and Table 12 ............19
Added Exposed Pad Notation to Figure 10 and Table 13 ..........21
Updated Outline Dimensions ........................................................84
Changes to Ordering Guide ...........................................................85
11/07—Revision 0: Initial Version
Rev. B | Page 3 of 84
AD6655
Data Sheet
GENERAL DESCRIPTION
The AD6655 is a mixed-signal intermediate frequency (IF) receiver
consisting of dual 14-bit, 80 MSPS/105 MSPS/125 MSPS/150 MSPS
ADCs and a wideband digital downconverter (DDC). The AD6655
is designed to support communications applications where low
cost, small size, and versatility are desired.
The dual ADC core features a multistage, differential pipelined
architecture with integrated output error correction logic. Each
ADC features wide bandwidth differential sample-and-hold
analog input amplifiers supporting a variety of user-selectable
input ranges. An integrated voltage reference eases design considerations. A duty cycle stabilizer is provided to compensate for
variations in the ADC clock duty cycle, allowing the converters
to maintain excellent performance.
ADC data outputs are internally connected directly to the digital
downconverter (DDC) of the receiver, simplifying layout and
reducing interconnection parasitics. The digital receiver has two
channels and provides processing flexibility. Each receive channel
has four cascaded signal processing stages: a 32-bit frequency
translator (numerically controlled oscillator (NCO)), a halfband decimating filter, a fixed FIR filter, and an fADC/8 fixedfrequency NCO.
In addition to the receiver DDC, the AD6655 has several
functions that simplify the automatic gain control (AGC)
function in the system receiver. The fast detect feature allows
fast overrange detection by outputting four bits of input level
information with short latency.
In addition, the programmable threshold detector allows
monitoring of the incoming signal power using the four fast
detect bits of the ADC with low latency. If the input signal level
exceeds the programmable threshold, the coarse upper threshold
indicator goes high. Because this threshold indicator has low
latency, the user can quickly turn down the system gain to avoid
an overrange condition.
The second AGC-related function is the signal monitor. This
block allows the user to monitor the composite magnitude of the
incoming signal, which aids in setting the gain to optimize the
dynamic range of the overall system.
After digital processing, data can be routed directly to the two
external 14-bit output ports. These outputs can be set from 1.8 V
to 3.3 V CMOS or as 1.8 V LVDS. The CMOS data can also be
output in an interleaved configuration at a double data rate using
only Port A.
The AD6655 receiver digitizes a wide spectrum of IF frequencies.
Each receiver is designed for simultaneous reception of the main
channel and the diversity channel. This IF sampling architecture
greatly reduces component cost and complexity compared with
traditional analog techniques or less integrated digital methods.
Flexible power-down options allow significant power savings,
when desired.
Programming for setup and control is accomplished using a 3-bit
SPI-compatible serial interface.
The AD6655 is available in a 64-lead LFCSP and is specified over
the industrial temperature range of −40°C to +85°C.
Rev. B | Page 4 of 84
Data Sheet
AD6655
SPECIFICATIONS
ADC DC SPECIFICATIONS—AD6655-80/AD6655-105
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference,
DCS enabled, unless otherwise noted.
Table 1.
Parameter
RESOLUTION
ACCURACY
No Missing Codes
Offset Error
Gain Error
MATCHING CHARACTERISTIC
Offset Error
Gain Error
TEMPERATURE DRIFT
Offset Error
Gain Error
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1 V Mode)
Load Regulation at 1.0 mA
INPUT-REFERRED NOISE
VREF = 1.0 V
ANALOG INPUT
Input Span, VREF = 1.0 V
Input Capacitance 1
VREF INPUT RESISTANCE
POWER SUPPLIES
Supply Voltage
AVDD, DVDD
DRVDD (CMOS Mode)
DRVDD (LVDS Mode)
Supply Current
IAVDD 2, 3
IDVDD2, 3
IDRVDD2 (3.3 V CMOS)
IDRVDD2 (1.8 V CMOS)
IDRVDD2 (1.8 V LVDS)
POWER CONSUMPTION
DC Input
Sine Wave Input2 (DRVDD = 1.8 V)
Sine Wave Input2 (DRVDD = 3.3 V)
Standby Power 4
Power-Down Power
Temperature
Full
Full
Full
Full
Min
14
−3.6
AD6655-80
Typ
Max
Guaranteed
±0.2
±0.6
−1.8
−0.1
Min
14
−4.3
Unit
Bits
Guaranteed
±0.2
±0.6
−2.2
−0.5
% FSR
% FSR
±0.2
±0.2
% FSR
% FSR
25°C
25°C
±0.2
±0.2
Full
Full
±15
±95
Full
Full
±5
7
25°C
0.85
0.85
LSB rms
Full
Full
Full
2
8
6
2
8
6
V p-p
pF
kΩ
Full
Full
Full
1.7
1.7
1.7
1.8
3.3
1.8
Full
Full
Full
Full
Full
235
175
18
8
55
Full
Full
Full
Full
Full
470
755
800
52
2.5
±0.6
±0.75
AD6655-105
Typ
Max
±0.6
±0.75
±15
±95
±18
1.9
3.6
1.9
420
490
8
±5
7
1.7
1.7
1.7
1.8
3.3
1.8
315
225
21
11
56
620
995
1040
68
2.5
ppm/°C
ppm/°C
±18
1.9
3.6
1.9
575
650
8
mV
mV
V
V
V
mA
mA
mA
mA
mA
mW
mW
mW
mW
mW
Input capacitance refers to the effective capacitance between one differential input pin and AGND. See Figure 11 for the equivalent analog input structure.
Measured with a 9.7 MHz, full-scale sine wave input, NCO enabled with a frequency of 13 MHz, FIR filter enabled and the fS/8 output mix enabled with approximately
5 pF loading on each output bit.
3
The maximum limit applies to the combination of IAVDD and IDVDD currents.
4
Standby power is measured with a dc input and with the CLK pin inactive (set to AVDD or AGND).
1
2
Rev. B | Page 5 of 84
AD6655
Data Sheet
ADC DC SPECIFICATIONS—AD6655-125/AD6655-150
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference,
DCS enabled, unless otherwise noted.
Table 2.
Parameter
RESOLUTION
ACCURACY
No Missing Codes
Offset Error
Gain Error
MATCHING CHARACTERISTIC
Offset Error
Gain Error
TEMPERATURE DRIFT
Offset Error
Gain Error
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1 V Mode)
Load Regulation at 1.0 mA
INPUT-REFERRED NOISE
VREF = 1.0 V
ANALOG INPUT
Input Span, VREF = 1.0 V
Input Capacitance 1
VREF INPUT RESISTANCE
POWER SUPPLIES
Supply Voltage
AVDD, DVDD
DRVDD (CMOS Mode)
DRVDD (LVDS Mode)
Supply Current
IAVDD 2, 3
IDVDD2, 3
IDRVDD2 (3.3 V CMOS)
IDRVDD2 (1.8 V CMOS)
IDRVDD2 (1.8 V LVDS)
POWER CONSUMPTION
DC Input
Sine Wave Input2 (DRVDD = 1.8 V)
Sine Wave Input2 (DRVDD = 3.3 V)
Standby Power 4
Power-down Power
Temperature
Full
Full
Full
Full
Min
14
−4.7
AD6655-125
Typ
Max
Guaranteed
±0.3
±0.6
−2.7
−0.8
Min
14
−5.1
Unit
Bits
Guaranteed
±0.2
±0.6
−3.2
−1.0
% FSR
% FSR
±0.2
±0.2
% FSR
% FSR
25°C
25°C
±0.3
±0.1
Full
Full
±15
±95
Full
Full
±5
7
25°C
0.85
0.85
LSB rms
Full
Full
Full
2
8
6
2
8
6
V p-p
pF
kΩ
Full
Full
Full
1.7
1.7
1.7
1.8
1.8
1.8
Full
Full
Full
Full
Full
390
270
26
13
57
Full
Full
Full
Full
Full
770
1215
1275
77
2.5
±0.7
±0.7
AD6655-150
Typ
Max
±0.7
±0.8
±15
±95
±18
1.9
3.6
1.9
705
810
8
±5
7
1.7
1.7
1.7
1.8
1.8
1.8
440
320
28
17
57
870
1395
1450
77
2.5
ppm/°C
ppm/°C
±18
1.9
3.6
1.9
805
920
8
mV
mV
V
V
V
mA
mA
mA
mA
mA
mW
mW
mW
mW
mW
Input capacitance refers to the effective capacitance between one differential input pin and AGND. See Figure 11 for the equivalent analog input structure.
Measured with a 9.7 MHz, full-scale sine wave input, NCO enabled with a frequency of 13 MHz, FIR filter enabled and the fS/8 output mix enabled with approximately
5 pF loading on each output bit.
3
The maximum limit applies to the combination of IAVDD and IDVDD currents.
4
Standby power is measured with a dc input, the CLK pin inactive (set to AVDD or AGND).
1
2
Rev. B | Page 6 of 84
Data Sheet
AD6655
ADC AC SPECIFICATIONS—AD6655-80/AD6655-105
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference,
DCS enabled, NCO enabled, half-band filter enabled, FIR filter enabled, unless otherwise noted.
Table 3.
Parameter 1
SIGNAL-TO-NOISE-RATIO (SNR)
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
WORST SECOND OR THIRD HARMONIC
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
SPURIOUS-FREE DYNAMIC RANGE
(SFDR)
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
WORST OTHER HARMONIC OR SPUR 2
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
TWO-TONE SFDR
fIN = 29.12 MHz, 32.12 MHz (−7 dBFS)
fIN = 169.12 MHz, 172.12 MHz (−7 dBFS)
CROSSTALK 3
ANALOG INPUT BANDWIDTH
Temperature
25°C
25°C
Full
25°C
25°C
Min
AD6655-80
Typ
Max
Min
74.9
74.8
AD6655-105
Typ
Max
74.8
74.7
73.0
dB
dB
dB
dB
dB
73.0
74.5
73.4
74.3
73.4
25°C
25°C
Full
25°C
25°C
−86
−85
−86
−85
−84
−83
−84
−83
25°C
25°C
Full
25°C
25°C
86
85
86
85
−74
74
Unit
−74
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
74
84
83
84
83
25°C
25°C
Full
25°C
25°C
−93
−90
−93
−90
−89
−86
−89
−86
dBc
dBc
dBc
dBc
dBc
25°C
25°C
Full
25°C
85
81
95
650
85
81
95
650
dBc
dBc
dB
MHz
−82
See Application Note AN-835, Understanding High Speed ADC Testing and Evaluation, for a complete set of definitions.
See the Applications Information section for more information about the worst other specifications for the AD6655.
3
Crosstalk is measured at 100 MHz with −1 dBFS on one channel and with no input on the alternate channel.
1
2
Rev. B | Page 7 of 84
−82
AD6655
Data Sheet
ADC AC SPECIFICATIONS—AD6655-125/AD6655-150
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference,
DCS enabled, NCO enabled, half-band filter enabled, FIR filter enabled, unless otherwise noted.
Table 4.
Parameter 1
SIGNAL-TO-NOISE-RATIO (SNR)
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
WORST SECOND OR THIRD HARMONIC
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
SPURIOUS-FREE DYNAMIC RANGE (SFDR)
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
WORST OTHER HARMONIC OR SPUR 2
fIN = 2.4 MHz
fIN = 70 MHz
fIN = 140 MHz
fIN = 220 MHz
TWO-TONE SFDR
fIN = 29.12 MHz, 32.12 MHz (−7 dBFS)
fIN = 169.12 MHz, 172.12 MHz (−7 dBFS)
CROSSTALK 3
ANALOG INPUT BANDWIDTH
1
2
3
Temperature
25°C
25°C
Full
25°C
25°C
Min
AD6655-125
Typ
Max
Min
74.7
74.6
AD6655-150
Typ
Max
74.6
74.5
73.0
dB
dB
dB
dB
dB
72.5
74.2
73.3
73.9
73.0
25°C
25°C
Full
25°C
25°C
−86
−85
−85
−84
−84
−83
−83
−77
25°C
25°C
Full
25°C
25°C
86
85
85
80
−73
73
73
Unit
−73
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
84
83
76
74
25°C
25°C
Full
25°C
25°C
−92
−90
−87
−80
−88
−84
−76
−74
dBc
dBc
dBc
dBc
dBc
25°C
25°C
Full
25°C
85
81
95
650
85
81
95
650
dBc
dBc
dB
MHz
−82
See Application Note AN-835, Understanding High Speed ADC Testing and Evaluation, for a complete set of definitions.
See the Applications Information section for more information about the worst other specifications for the AD6655.
Crosstalk is measured at 100 MHz with −1 dBFS on one channel and with no input on the alternate channel.
Rev. B | Page 8 of 84
−80
Data Sheet
AD6655
DIGITAL SPECIFICATIONS—AD6655-80/AD6655-105
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference,
DCS enabled, unless otherwise noted.
Table 5.
Parameter
DIFFERENTIAL CLOCK INPUTS (CLK+, CLK−)
Logic Compliance
Internal Common-Mode Bias
Differential Input Voltage
Input Voltage Range
Input Common-Mode Range
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Capacitance
Input Resistance
SYNC INPUT
Logic Compliance
Internal Bias
Input Voltage Range
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Capacitance
Input Resistance
LOGIC INPUT (CSB) 1
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUT (SCLK/DFS) 2
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUTS (SDIO/DCS, SMI SDFS)1
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
AD6655-80
Typ
Max
Min
Full
Full
Full
Full
Full
Full
Full
Full
Full
Full
CMOS/LVDS/LVPECL
1.2
0.2
6
AVDD − 0.3
AVDD + 1.6
1.1
AVDD
1.2
3.6
0
0.8
−10
+10
−10
+10
4
8
10
12
CMOS/LVDS/LVPECL
1.2
0.2
6
AVDD − 0.3
AVDD + 1.6
1.1
AVDD
1.2
3.6
0
0.8
−10
+10
−10
+10
4
8
10
12
CMOS
1.2
CMOS
1.2
Full
Full
Full
Full
Full
Full
Full
Full
AVDD − 0.3
1.2
0
−10
−10
8
Full
Full
Full
Full
Full
Full
1.22
0
−10
40
Full
Full
Full
Full
Full
Full
1.22
0
−92
−10
Full
Full
Full
Full
Full
Full
1.22
0
−10
38
4
10
Min
AD6655-105
Typ
Max
Temp
AVDD + 1.6
3.6
0.8
+10
+10
AVDD − 0.3
1.2
0
−10
−10
12
8
3.6
0.6
+10
132
1.22
0
−10
40
26
2
1.22
0
−92
−10
26
2
12
1.22
0
−10
38
26
5
V
V p-p
V
V
V
V
µA
µA
pF
kΩ
V
V
V
V
µA
µA
pF
kΩ
3.6
0.6
+10
132
V
V
µA
µA
kΩ
pF
3.6
0.6
−135
+10
V
V
µA
µA
kΩ
pF
3.6
0.6
+10
128
V
V
µA
µA
kΩ
pF
26
2
3.6
0.6
+10
128
Rev. B | Page 9 of 84
4
10
26
2
3.6
0.6
−135
+10
26
5
AVDD + 1.6
3.6
0.8
+10
+10
Unit
AD6655
Parameter
LOGIC INPUTS (SMI SDO/OEB,
SMI SCLK/PDWN)2
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
DIGITAL OUTPUTS
CMOS Mode—DRVDD = 3.3 V
High Level Output Voltage
IOH = 50 µA
IOH = 0.5 mA
Low Level Output Voltage
IOL = 1.6 mA
IOL = 50 µA
CMOS Mode—DRVDD = 1.8 V
High Level Output Voltage
IOH = 50 µA
IOH = 0.5 mA
Low Level Output Voltage
IOL = 1.6 mA
IOL = 50 µA
LVDS Mode, DRVDD = 1.8 V
Differential Output Voltage (VOD),
ANSI Mode
Output Offset Voltage (VOS),
ANSI Mode
Differential Output Voltage (VOD),
Reduced Swing Mode
Output Offset Voltage (VOS),
Reduced Swing Mode
1
2
Data Sheet
Temp
Min
AD6655-80
Typ
Max
Min
AD6655-105
Typ
Max
Full
Full
Full
Full
Full
Full
1.22
0
−90
−10
3.6
0.6
−134
+10
1.22
0
−90
−10
3.6
0.6
−134
+10
Full
Full
3.29
3.25
26
5
3.29
3.25
1.79
1.75
0.2
0.05
1.79
1.75
V
V
V
V
0.2
0.05
Full
Full
V
V
µA
µA
kΩ
pF
V
V
0.2
0.05
Full
Full
Full
Full
26
5
Unit
0.2
0.05
V
V
Full
250
350
450
250
350
450
mV
Full
1.15
1.25
1.35
1.15
1.25
1.35
V
Full
150
200
280
150
200
280
mV
Full
1.15
1.25
1.35
1.15
1.25
1.35
V
Pull up.
Pull down.
Rev. B | Page 10 of 84
Data Sheet
AD6655
DIGITAL SPECIFICATIONS—AD6655-125/AD6655-150
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference,
DCS enabled, unless otherwise noted.
Table 6.
Parameter
DIFFERENTIAL CLOCK INPUTS (CLK+, CLK−)
Logic Compliance
Internal Common-Mode Bias
Differential Input Voltage
Input Voltage Range
Input Common-Mode Range
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Capacitance
Input Resistance
SYNC INPUT
Logic Compliance
Internal Bias
Input Voltage Range
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Capacitance
Input Resistance
LOGIC INPUT (CSB) 1
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUT (SCLK/DFS) 2
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
LOGIC INPUTS (SDIO/DCS, SMI SDFS)1
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
AD6655-125
Typ
Max
Min
Full
Full
Full
Full
Full
Full
Full
Full
Full
Full
CMOS/LVDS/LVPECL
1.2
0.2
6
AVDD − 0.3
AVDD + 1.6
1.1 V
AVDD
1.2
3.6
0
0.8
−10
+10
−10
+10
4
8
10
12
CMOS/LVDS/LVPECL
1.2
0.2
6
AVDD − 0.3
AVDD + 1.6
1.1 V
AVDD
1.2
3.6
0
0.8
−10
+10
−10
+10
4
8
10
12
CMOS
1.2
CMOS
1.2
Full
Full
Full
Full
Full
Full
Full
Full
AVDD − 0.3
1.2
0
−10
−10
8
Full
Full
Full
Full
Full
Full
1.22
0
−10
40
Full
Full
Full
Full
Full
Full
1.22
0
−92
−10
Full
Full
Full
Full
Full
Full
1.22
0
−10
38
4
10
Min
AD6655-150
Typ
Max
Temp
AVDD + 1.6
3.6
0.8
+10
+10
AVDD − 0.3
1.2
0
−10
−10
12
8
3.6
0.6
+10
132
1.22
0
−10
40
26
2
1.22
0
−92
−10
26
2
12
1.22
0
−10
38
26
5
V
V p-p
V
V
V
V
µA
µA
pF
kΩ
V
V
V
V
µA
µA
pF
kΩ
3.6
0.6
+10
132
V
V
µA
µA
kΩ
pF
3.6
0.6
−135
+10
V
V
µA
µA
kΩ
pF
3.6
0.6
+10
128
V
V
µA
µA
kΩ
pF
26
2
3.6
0.6
+10
128
Rev. B | Page 11 of 84
4
10
26
2
3.6
0.6
−135
+10
26
5
AVDD + 1.6
3.6
0.8
+10
+10
Unit
AD6655
Parameter
LOGIC INPUTS (SMI SDO/OEB,
SMI SCLK/PDWN)2
High Level Input Voltage
Low Level Input Voltage
High Level Input Current
Low Level Input Current
Input Resistance
Input Capacitance
DIGITAL OUTPUTS
CMOS Mode—DRVDD = 3.3 V
High Level Output Voltage
IOH = 50 µA
IOH = 0.5 mA
Low Level Output Voltage
IOL = 1.6 mA
IOL = 50 µA
CMOS Mode—DRVDD = 1.8 V
High Level Output Voltage
IOH = 50 µA
IOH = 0.5 mA
Low Level Output Voltage
IOL = 1.6 mA
IOL = 50 µA
LVDS Mode—DRVDD = 1.8 V
Differential Output Voltage (VOD),
ANSI Mode
Output Offset Voltage (VOS),
ANSI Mode
Differential Output Voltage (VOD),
Reduced Swing Mode
Output Offset Voltage (VOS),
Reduced Swing Mode
1
2
Data Sheet
Temp
Min
AD6655-125
Typ
Max
Min
AD6655-150
Typ
Max
Full
Full
Full
Full
Full
Full
1.22
0
−90
−10
3.6
0.6
−134
+10
1.22
0
−90
−10
3.6
0.6
−134
+10
Full
Full
3.29
3.25
26
5
3.29
3.25
Full
Full
Full
Full
26
5
0.2
0.05
1.79
1.75
Full
Full
V
V
µA
µA
kΩ
pF
V
V
0.2
0.05
1.79
1.75
Unit
V
V
V
V
0.2
0.05
0.2
0.05
V
V
Full
250
350
450
250
350
450
mV
Full
1.15
1.25
1.35
1.15
1.25
1.35
V
Full
150
200
280
150
200
280
mV
Full
1.15
1.25
1.35
1.15
1.25
1.35
V
Pull up.
Pull down.
Rev. B | Page 12 of 84
Data Sheet
AD6655
SWITCHING SPECIFICATIONS—AD6655-80/AD6655-105
Table 7.
Parameter
CLOCK INPUT PARAMETERS
Input Clock Rate
Conversion Rate 1
DCS Enabled
DCS Disabled
CLK Period—Divide-by-1 Mode (tCLK)
CLK Pulse Width High (tCLKH)
Divide-by-1 Mode, DCS Enabled
Divide-by-1 Mode DCS Disabled
Divide-by-2 Mode, DCS Enabled
Divide-by-3 Through Divide-by-8 Modes, DCS Enabled
DATA OUTPUT PARAMETERS (DATA, FD)
CMOS Noninterleaved Mode—DRVDD = 1.8 V
Data Propagation Delay (tPD) 2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
CMOS Noninterleaved Mode—DRVDD = 3.3 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
CMOS Interleaved and IQ Mode—DRVDD = 1.8 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
CMOS Interleaved and IQ Mode—DRVDD = 3.3 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
LVDS Mode—DRVDD = 1.8 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Pipeline Delay (Latency) NCO, FIR, fS/8 Mix Disabled
Pipeline Delay (Latency) NCO Enabled, FIR and fS/8 Mix Disabled
(Complex Output Mode)
Pipeline Delay (Latency) NCO, FIR, and fS/8 Mix Enabled
Aperture Delay (tA)
Aperture Uncertainty (Jitter, tJ)
Wake-Up Time 3
OUT-OF-RANGE RECOVERY TIME
Temp
Min
AD6655-80
Typ
Max
Full
Min
AD6655-105
Typ
Max
625
Full
Full
Full
20
10
12.5
Full
Full
Full
Full
3.75
5.63
1.6
0.8
Full
Full
Full
Full
Unit
625
MHz
105
105
MSPS
MSPS
ns
80
80
20
10
9.5
6.25
6.25
8.75
6.88
2.85
4.28
1.6
0.8
4.75
4.75
6.65
5.23
ns
ns
ns
ns
1.6
4.0
3.9
5.4
14.0
11.0
6.2
7.3
1.6
4.0
3.9
5.4
11.0
8.0
6.2
7.3
ns
ns
ns
ns
Full
Full
Full
Full
1.9
4.4
4.1
5.8
14.2
10.8
6.4
7.7
1.9
4.4
4.1
5.8
11.2
7.8
6.4
7.7
ns
ns
ns
ns
Full
Full
Full
Full
1.6
3.4
3.9
4.8
7.15
5.35
6.2
6.7
1.6
3.4
3.9
4.8
5.65
3.85
6.2
6.7
ns
ns
ns
ns
Full
Full
Full
Full
1.9
3.8
4.1
5.2
7.35
5.15
6.4
7.1
1.9
3.8
4.1
5.2
5.85
3.65
6.4
7.1
ns
ns
ns
ns
Full
Full
Full
Full
2.5
3.7
4.8
5.3
38
38
7.0
7.3
2.5
3.7
4.8
5.3
38
38
7.0
7.3
ns
ns
Cycles
Cycles
Full
Full
Full
Full
Full
Conversion rate is the clock rate after the divider.
Output propagation delay is measured from CLK 50% transition to DATA 50% transition, with a 5 pF load.
3
Wake-up time is dependent on the value of the decoupling capacitors.
1
2
Rev. B | Page 13 of 84
109
1.0
0.1
350
2
109
1.0
0.1
350
2
Cycles
ns
ps rms
us
Cycles
AD6655
Data Sheet
SWITCHING SPECIFICATIONS—AD6655-125/AD6655-150
Table 8.
Parameter
CLOCK INPUT PARAMETERS
Input Clock Rate
Conversion Rate 1
DCS Enabled
DCS Disabled
CLK Period—Divide-by-1 Mode (tCLK)
CLK Pulse Width High (tCLKH)
Divide-by-1 Mode, DCS Enabled
Divide-by-1 Mode, DCS Disabled
Divide-by-2 Mode, DCS Enabled
Divide-by-3 Through Divide-by-8 Modes, DCS Enabled
DATA OUTPUT PARAMETERS (DATA, FD)
CMOS Noninterleaved Mode—DRVDD = 1.8 V
Data Propagation Delay (tPD) 2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
CMOS Noninterleaved Mode—DRVDD = 3.3 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
CMOS Interleaved and IQ Mode—DRVDD = 1.8 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
CMOS Interleaved and IQ Mode—DRVDD = 3.3 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Setup Time (tS)
Hold Time (tH)
LVDS Mode—DRVDD = 1.8 V
Data Propagation Delay (tPD)2
DCO Propagation Delay (tDCO)
Pipeline Delay (Latency) NCO, FIR, fS/8 Mix Disabled
Pipeline Delay (Latency) NCO Enabled; FIR and fS/8 Mix Disabled
(Complex Output Mode)
Pipeline Delay (Latency) NCO, FIR, and fS/8 Mix Enabled
Aperture Delay (tA)
Aperture Uncertainty (Jitter, tJ)
Wake-Up Time 3
OUT-OF-RANGE RECOVERY TIME
Temp
AD6655-125
Min
Typ
Max
Full
AD6655-150
Min
Typ
Max
625
Full
Full
Full
20
10
8
Full
Full
Full
Full
2.4
3.6
1.6
0.8
Full
Full
Full
Full
Unit
625
MHz
150
150
MSPS
MSPS
ns
125
125
20
10
6.66
4
4
5.6
4.4
2.0
3.0
1.6
0.8
3.33
3.33
4.66
3.66
ns
ns
ns
ns
1.6
4.0
3.9
5.4
9.5
6.5
6.2
7.3
1.6
4.0
3.9
5.4
8.16
5.16
6.2
7.3
ns
ns
ns
ns
Full
Full
Full
Full
1.9
4.4
4.1
5.8
9.7
6.3
6.4
7.7
1.9
4.4
4.1
5.8
8.36
4.96
6.4
7.7
ns
ns
ns
ns
Full
Full
Full
Full
1.6
3.4
3.9
4.8
4.9
3.1
6.2
6.7
1.6
3.4
3.9
4.8
4.23
2.43
6.2
6.7
ns
ns
ns
ns
Full
Full
Full
Full
1.9
3.8
4.1
5.2
5.1
2.9
6.4
7.1
1.9
3.8
4.1
5.2
4.43
2.23
6.4
7.1
ns
ns
ns
ns
Full
Full
Full
Full
2.5
3.7
4.8
5.3
38
38
7.0
7.3
2.5
3.7
4.8
5.3
38
38
7.0
7.3
ns
ns
Cycles
Cycles
Full
Full
Full
Full
Full
Conversion rate is the clock rate after the divider.
Output propagation delay is measured from CLK 50% transition to DATA 50% transition, with a 5 pF load.
3
Wake-up time is dependent on the value of the decoupling capacitors.
1
2
Rev. B | Page 14 of 84
109
1.0
0.1
350
3
109
1.0
0.1
350
3
Cycles
ns
ps rms
us
Cycles
Data Sheet
AD6655
TIMING SPECIFICATIONS
Table 9.
Parameter
SYNC TIMING REQUIREMENTS
tSSYNC
tHSYNC
SPI TIMING REQUIREMENTS
tDS
tDH
tCLK
tS
tH
tHIGH
tLOW
tEN_SDIO
Conditions
Min
SYNC to the rising edge of CLK setup time
SYNC to the rising edge of CLK hold time
tDIS_SDIO
SPORT TIMING REQUIREMENTS
tCSSCLK
tSSLKSDO
tSSCLKSDFS
Typ
Max
Unit
0.24
0.4
ns
ns
Setup time between the data and the rising edge of SCLK
Hold time between the data and the rising edge of SCLK
Period of the SCLK
Setup time between CSB and SCLK
Hold time between CSB and SCLK
Minimum period that SCLK should be in a logic high state
Minimum period that SCLK should be in a logic low state
Time required for the SDIO pin to switch from an input to an output
relative to the SCLK falling edge
Time required for the SDIO pin to switch from an output to an input
relative to the SCLK rising edge
2
2
40
2
2
10
10
10
ns
ns
ns
ns
ns
ns
ns
ns
10
ns
Delay from rising edge of CLK+ to rising edge of SMI SCLK
Delay from rising edge of SMI SCLK to SMI SDO
Delay from rising edge of SMI SCLK to SMI SDFS
3.2
−0.4
−0.4
4.5
0
0
6.2
+0.4
+0.4
ns
ns
ns
Timing Diagrams
CLK+
tDCO
tPD
CHANNEL A/B
DATA BITS
DECIMATED
CMOS DATA
DECIMATED
FD DATA
CHANNEL A/B
FD BITS
CHANNEL A/B
FD BITS
CHANNEL A/B
DATA BITS
CHANNEL A/B
FD BITS
CHANNEL A/B
FD BITS
CHANNEL A/B
FD BITS
CHANNEL A/B
DATA BITS
CHANNEL A/B
FD BITS
06709-109
tS
DECIMATED
DCOA/DCOB
tH
Figure 2. Decimated Noninterleaved CMOS Mode Data and Fast Detect Output Timing (Fast Detect Mode Select Bits = 000)
CLK+
tPD
tDCO
DECIMATED
CMOS DATA
CHANNEL A/B
DATA BITS
CHANNEL A/B
DATA BITS
CHANNEL A/B
DATA BITS
DECIMATED
FD DATA
CHANNEL A/B
FD BITS
CHANNEL A/B
FD BITS
CHANNEL A/B
FD BITS
DECIMATED
DCOA/DCOB
06709-012
tS
tH
Figure 3. Decimated Noninterleaved CMOS Mode Data and Fast Detect Output Timing (Fast Detect Mode Select Bits = 001 Through Fast Detect Mode Select Bits = 100)
Rev. B | Page 15 of 84
AD6655
Data Sheet
CLK+
tDCO
tPD
DECIMATED
INTERLEAVED
CMOS DATA
CHANNEL A:
DATA
CHANNEL B:
DATA
CHANNEL A:
DATA
CHANNEL B:
DATA
CHANNEL A:
DATA
CHANNEL B:
DATA
DECIMATED
INTERLEAVED
FD DATA
CHANNEL A:
FD BITS
CHANNEL B:
FD BITS
CHANNEL A:
FD BITS
CHANNEL B:
FD BITS
CHANNEL A:
FD BITS
CHANNEL B:
FD BITS
06709-013
tS
DECIMATED
DCO
tH
Figure 4. Decimated Interleaved CMOS Mode Data and Fast Detect Output Timing
CLK+
tPD
tDCO
DECIMATED
CMOS IQ
OUTPUT DATA
CHANNEL A/B:
Q DATA
CHANNEL A/B:
I DATA
CHANNEL A/B:
Q DATA
CHANNEL A/B:
I DATA
CHANNEL A/B:
Q DATA
CHANNEL A/B:
I DATA
CMOS FD
DATA
CHANNEL A/B:
FD BITS
CHANNEL A/B:
FD BITS
CHANNEL A/B:
FD BITS
CHANNEL A/B:
FD BITS
CHANNEL A/B:
FD BITS
CHANNEL A/B:
FD BITS
06709-014
tS
DECIMATED
DCOA/DCOB
tH
Figure 5. Decimated IQ Mode CMOS Data and Fast Detect Output Timing
CLK–
CLK+
tPD
LVDS
DATA
CHANNEL A:
DATA
CHANNEL B:
DATA
CHANNEL A:
DATA
CHANNEL B:
DATA
CHANNEL A:
DATA
LVDS
FAST DET
CHANNEL A:
FD
CHANNEL B:
FD
CHANNEL A:
FD
CHANNEL B:
FD
CHANNEL A:
FD
tDCO
06709-015
DCO–
DCO+
Figure 6. Decimated Interleaved LVDS Mode Data and Fast Detect Output Timing
CLK+
tHSYNC
06709-016
tSSYNC
SYNC
Figure 7. SYNC Timing Inputs
Rev. B | Page 16 of 84
Data Sheet
AD6655
CLK+
CLK–
tCSSCLK
SMI SCLK
tSSCLKSDFS
tSSCLKSDFS
SMI SDO
DATA
Figure 8. Signal Monitor SPORT Output Timing
Rev. B | Page 17 of 84
DATA
06709-017
SMI SDFS
AD6655
Data Sheet
ABSOLUTE MAXIMUM RATINGS
Table 10.
Parameter
ELECTRICAL
AVDD, DVDD to AGND
DRVDD to DRGND
AGND to DRGND
VIN+A/VIN+B, VIN-A/VIN−B to AGND
CLK+, CLK− to AGND
SYNC to AGND
VREF to AGND
SENSE to AGND
CML to AGND
RBIAS to AGND
CSB to AGND
SCLK/DFS to DRGND
SDIO/DCS to DRGND
SMI SDO/OEB to DRGND
SMI SCLK/PDWN to DRGND
SMI SDFS to DRGND
D0A/D0B through D13A/D13B
to DRGND
FD0A/FD0B through FD3A/FD3B to
DRGND
DCOA/DCOB to DRGND
ENVIRONMENTAL
Operating Temperature Range
(Ambient)
Maximum Junction Temperature
Under Bias
Storage Temperature Range
(Ambient)
THERMAL CHARACTERISTICS
Rating
−0.3 V to +2.0 V
−0.3 V to +3.9 V
−0.3 V to +0.3 V
−0.3 V to AVDD + 0.2 V
−0.3 V to +3.9 V
−0.3 V to +3.9 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to AVDD + 0.2 V
−0.3 V to +3.9 V
−0.3 V to +3.9 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
−0.3 V to DRVDD + 0.3 V
The exposed paddle must be soldered to the ground plane for
the LFCSP package. Soldering the exposed paddle to the
customer board increases the reliability of the solder joints,
maximizing the thermal capability of the package.
Table 11. Thermal Resistance
Package
Type
64-Lead LFCSP
9 mm × 9 mm
(CP-64-6)
Airflow
Velocity
(m/s)
0
1.0
2.0
θJA1, 2
18.8
16.5
15.8
θJC1, 3
0.6
θJB1, 4
6.0
Unit
°C/W
°C/W
°C/W
Per JEDEC 51-7, plus JEDEC 25-5 2S2P test board.
Per JEDEC JESD51-2 (still air) or JEDEC JESD51-6 (moving air).
3
Per MIL-Std 883, Method 1012.1.
4
Per JEDEC JESD51-8 (still air).
1
2
Typical θJA is specified for a 4-layer PCB with solid ground
plane. As shown, airflow increases heat dissipation, which
reduces θJA. In addition, metal in direct contact with the
package leads from metal traces, through holes, ground, and
power planes, reduces the θJA.
ESD CAUTION
−40°C to +85°C
150°C
−65°C to +125°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Rev. B | Page 18 of 84
Data Sheet
AD6655
64
63
62
61
60
59
58
57
56
55
54
53
52
51
50
49
DRGND
D5B
D4B
D3B
D2B
D1B
D0B (LSB)
DVDD
FD3B
FD2B
FD1B
FD0B
SYNC
CSB
CLK–
CLK+
PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
PIN 1
INDICATOR
EXPOSED PADDLE, PIN 0
(BOTTOM OF PACKAGE)
AD6655
PARALLEL CMOS
TOP VIEW
(Not to Scale)
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
SCLK/DFS
SDIO/DCS
AVDD
AVDD
VIN+B
VIN–B
RBIAS
CML
SENSE
VREF
VIN–A
VIN+A
AVDD
SMI SDFS
SMI SCLK/PDWN
SMI SDO/OEB
NOTES
1. THE EXPOSED THERMAL PAD ON THE BOTTOM OF THE PACKAGE PROVIDES THE
ANALOG GROUND FOR THE PART. THIS EXPOSED PAD MUST BE CONNECTED TO
GROUND FOR PROPER OPERATION.
06709-002
D5A
D6A
D7A
DRGND
DRVDD
D8A
D9A
DVDD
D10A
D11A
D12A
D13A (MSB)
FD0A
FD1A
FD2A
FD3A
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
DRVDD
D6B
D7B
D8B
D9B
D10B
D11B
D12B
D13B (MSB)
DCOB
DCOA
D0A (LSB)
D1A
D2A
D3A
D4A
Figure 9. LFCSP Parallel CMOS Pin Configuration (Top View)
Table 12. Pin Function Descriptions (Parallel CMOS Mode)
Pin No.
Mnemonic
ADC Power Supplies
20, 64
DRGND
1, 21
DRVDD
24, 57
DVDD
36, 45, 46
AVDD
0
AGND,
Exposed Pad
ADC Analog
37
VIN+A
38
VIN−A
44
VIN+B
43
VIN−B
39
VREF
40
SENSE
42
RBIAS
41
CML
49
CLK+
50
CLK−
ADC Fast Detect Outputs
29
FD0A
30
FD1A
31
FD2A
32
FD3A
53
FD0B
54
FD1B
55
FD2B
56
FD3B
Type
Description
Ground
Supply
Supply
Supply
Ground
Digital Output Ground.
Digital Output Driver Supply (1.8 V to 3.3 V).
Digital Power Supply (1.8 V Nominal).
Analog Power Supply (1.8 V Nominal).
Analog Ground. The exposed thermal pad on the bottom of the package provides the
analog ground for the part. This pad must be connected to ground for proper operation.
Input
Input
Input
Input
Input/Output
Input
Input/Output
Output
Input
Input
Differential Analog Input Pin (+) for Channel A.
Differential Analog Input Pin (−) for Channel A.
Differential Analog Input Pin (+) for Channel B.
Differential Analog Input Pin (−) for Channel B.
Voltage Reference Input/Output.
Voltage Reference Mode Select. (See Table 15 for details.)
External Reference Bias Resistor.
Common-Mode Level Bias Output for Analog Inputs.
ADC Clock Input—True.
ADC Clock Input—Complement.
Output
Output
Output
Output
Output
Output
Output
Output
Channel A Fast Detect Indicator. (See Table 21 for details.)
Channel A Fast Detect Indicator. (See Table 21 for details.)
Channel A Fast Detect Indicator. (See Table 21 for details.)
Channel A Fast Detect Indicator. (See Table 21 for details.)
Channel B Fast Detect Indicator. (See Table 21 for details.)
Channel B Fast Detect Indicator. (See Table 21 for details.)
Channel B Fast Detect Indicator. (See Table 21 for details.)
Channel B Fast Detect Indicator. (See Table 21 for details.)
Rev. B | Page 19 of 84
AD6655
Pin No.
Mnemonic
Digital Input
52
SYNC
Digital Outputs
12
D0A (LSB)
13
D1A
14
D2A
15
D3A
16
D4A
17
D5A
18
D6A
19
D7A
22
D8A
23
D9A
25
D10A
26
D11A
27
D12A
28
D13A (MSB)
58
D0B (LSB)
59
D1B
60
D2B
61
D3B
62
D4B
63
D5B
2
D6B
3
D7B
4
D8B
5
D9B
6
D10B
7
D11B
8
D12B
9
D13B (MSB)
11
DCOA
10
DCOB
SPI Control
48
SCLK/DFS
47
SDIO/DCS
51
CSB
Signal Monitor Port
33
SMI SDO/OEB
35
SMI SDFS
34
SMI SCLK/PDWN
Data Sheet
Type
Description
Input
Digital Synchronization Pin. Slave mode only.
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel A CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel B CMOS Output Data.
Channel A Data Clock Output.
Channel B Data Clock Output.
Input
Input/Output
Input
SPI Serial Clock/Data Format Select Pin in External Pin Mode.
SPI Serial Data I/O/Duty Cycle Stabilizer Pin in External Pin Mode.
SPI Chip Select. Active low.
Input/Output
Output
Input/Output
Signal Monitor Serial Data Output/Output Enable Input (Active Low) in External Pin Mode.
Signal Monitor Serial Data Frame Sync.
Signal Monitor Serial Clock Output/Power-Down Input (Active High) in External Pin Mode.
Rev. B | Page 20 of 84
AD6655
64
63
62
61
60
59
58
57
56
55
54
53
52
51
50
49
DRGND
D0+ (LSB)
D0– (LSB)
FD3+
FD3–
FD2+
FD2–
DVDD
FD1+
FD1–
FD0+
FD0–
SYNC
CSB
CLK–
CLK+
Data Sheet
1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
PIN 1
INDICATOR
EXPOSED PADDLE, PIN 0
(BOTTOM OF PACKAGE)
AD6655
PARALLEL LVDS
TOP VIEW
(Not to Scale)
48
47
46
45
44
43
42
41
40
39
38
37
36
35
34
33
SCLK/DFS
SDIO/DCS
AVDD
AVDD
VIN+B
VIN–B
RBIAS
CML
SENSE
VREF
VIN–A
VIN+A
AVDD
SMI SDFS
SMI SCLK/PDWN
SMI SDO/OEB
NOTES
1. THE EXPOSED THERMAL PAD ON THE BOTTOM OF THE PACKAGE PROVIDES THE
ANALOG GROUND FOR THE PART. THIS EXPOSED PAD MUST BE CONNECTED TO
GROUND FOR PROPER OPERATION.
06709-003
D7+
D8–
D8+
DRGND
DRVDD
D9–
D9+
DVDD
D10–
D10+
D11–
D11+
D12–
D12+
D13– (MSB)
D13+ (MSB)
17
18
19
20
21
22
23
24
25
26
27
28
29
30
31
32
DRVDD
D1–
D1+
D2–
D2+
D3–
D3+
D4–
D4+
DCO–
DCO+
D5–
D5+
D6–
D6+
D7–
Figure 10. LFCSP Interleaved Parallel LVDS Pin Configuration (Top View)
Table 13. Pin Function Descriptions (Interleaved Parallel LVDS Mode)
Pin No.
Mnemonic
ADC Power Supplies
20, 64
DRGND
1, 21
DRVDD
24, 57
DVDD
36, 45, 46 AVDD
0
AGND,
Exposed Pad
ADC Analog
37
VIN+A
38
VIN−A
44
VIN+B
43
VIN−B
39
VREF
40
SENSE
42
RBIAS
41
CML
49
CLK+
50
CLK−
ADC Fast Detect Outputs
54
FD0+
53
FD056
FD1+
55
FD1−
59
FD2+
58
FD2−
61
FD3+
60
FD3−
Type
Description
Ground
Supply
Supply
Supply
Ground
Digital Output Ground.
Digital Output Driver Supply (1.8 V to 3.3 V).
Digital Power Supply (1.8 V Nominal.)
Analog Power Supply (1.8 V Nominal.)
Analog Ground. The exposed thermal pad on the bottom of the package provides the analog
ground for the part. This exposed pad must be connected to ground for proper operation.
Input
Input
Input
Input
Input/Output
Input
Input/Output
Output
Input
Input
Differential Analog Input Pin (+) for Channel A.
Differential Analog Input Pin (−) for Channel A.
Differential Analog Input Pin (+) for Channel B.
Differential Analog Input Pin (−) for Channel B.
Voltage Reference Input/Output.
Voltage Reference Mode Select. See Table 15 for details.
External Reference Bias Resistor.
Common-Mode Level Bias Output for Analog Inputs.
ADC Clock Input—True.
ADC Clock Input—Complement.
Output
Output
Output
Output
Output
Output
Output
Output
Channel A/Channel B LVDS Fast Detect Indicator 0—True. See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 0—Complement. See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 1—True. See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 1—Complement. See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 2—True See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 2—Complement. See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 3—True. See Table 21 for details.
Channel A/Channel B LVDS Fast Detect Indicator 3—Complement. See Table 21 for details.
Rev. B | Page 21 of 84
AD6655
Pin No.
Mnemonic
Digital Input
52
SYNC
Digital Outputs
63
D0+ (LSB)
62
D0− (LSB)
3
D1+
2
D1−
5
D2+
4
D2−
7
D3+
6
D3−
9
D4+
8
D4−
13
D5+
12
D5−
15
D6+
14
D6−
17
D7+
16
D7−
19
D8+
18
D8−
23
D9+
22
D9−
26
D10+
25
D10−
28
D11+
27
D11−
30
D12+
29
D12−
32
D13+ (MSB)
31
D13− (MSB)
11
DCO+
10
DCO−
SPI Control
48
SCLK/DFS
47
SDIO/DCS
51
CSB
Signal Monitor Port
33
SMI SDO/OEB
35
SMI SDFS
34
SMI SCLK/PDWN
Data Sheet
Type
Description
Input
Digital Synchronization Pin. Slave mode only.
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Output
Channel A/Channel B LVDS Output Data 0—True.
Channel A/Channel B LVDS Output Data 0—Complement.
Channel A/Channel B LVDS Output Data 1—True.
Channel A/Channel B LVDS Output Data 1—Complement.
Channel A/Channel B LVDS Output Data 2—True.
Channel A/Channel B LVDS Output Data 2—Complement.
Channel A/Channel B LVDS Output Data 3—True.
Channel A/Channel B LVDS Output Data 3—Complement.
Channel A/Channel B LVDS Output Data 4—True.
Channel A/Channel B LVDS Output Data 4—Complement.
Channel A/Channel B LVDS Output Data 5—True.
Channel A/Channel B LVDS Output Data 5—Complement.
Channel A/Channel B LVDS Output Data 6—True.
Channel A/Channel B LVDS Output Data 6—Complement.
Channel A/Channel B LVDS Output Data 7—True.
Channel A/Channel B LVDS Output Data 7—Complement.
Channel A/Channel B LVDS Output Data 8—True.
Channel A/Channel B LVDS Output Data 8—Complement.
Channel A/Channel B LVDS Output Data 9—True.
Channel A/Channel B LVDS Output Data 9—Complement.
Channel A/Channel B LVDS Output Data 10—True.
Channel A/Channel B LVDS Output Data 10—Complement.
Channel A/Channel B LVDS Output Data 11—True.
Channel A/Channel B LVDS Output Data 11—Complement.
Channel A/Channel B LVDS Output Data 12—True.
Channel A/Channel B LVDS Output Data 12—Complement.
Channel A/Channel B LVDS Output Data 13—True.
Channel A/Channel B LVDS Output Data 13—Complement.
Channel A/Channel B LVDS Data Clock Output—True.
Channel A/Channel B LVDS Data Clock Output—Complement.
Input
Input/Output
Input
SPI Serial Clock/Data Format Select Pin in External Pin Mode.
SPI Serial Data I/O/Duty Cycle Stabilizer in External Pin Mode.
SPI Chip Select (Active Low).
Input/Output
Output
Input/Output
Signal Monitor Serial Data Output/Output Enable Input (Active Low) in External Pin Mode.
Signal Monitor Serial Data Frame Sync.
Signal Monitor Serial Clock Output/Power-Down Input (Active High) in External Pin Mode.
Rev. B | Page 22 of 84
Data Sheet
AD6655
EQUIVALENT CIRCUITS
1kΩ
SCLK/DFS
VIN
06709-008
06709-004
26kΩ
Figure 11. Equivalent Analog Input Circuit
Figure 15. Equivalent SCLK/DFS Input Circuit
AVDD
1.2V
10kΩ
CLK–
06709-009
06709-005
CLK+
1kΩ
SENSE
10kΩ
Figure 12. Equivalent Clock lnput Circuit
Figure 16. Equivalent SENSE Circuit
DRVDD
AVDD
26kΩ
1kΩ
DRGND
06709-010
06709-006
CSB
Figure 17. Equivalent CSB Input Circuit
Figure 13. Equivalent Digital Output Circuit
AVDD
DRVDD
DRVDD
VREF
26kΩ
6kΩ
1kΩ
06709-011
06709-007
SDIO/DCS
Figure 18. Equivalent VREF Circuit
Figure 14. Equivalent SDIO/DCS Circuit or SMI SDFS Circuit
.
Rev. B | Page 23 of 84
AD6655
Data Sheet
TYPICAL PERFORMANCE CHARACTERISTICS
AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, sample rate = 150 MSPS, DCS enabled, 1.0 V internal reference, 2 V p-p differential
input, VIN = −1.0 dBFS, 64k sample, TA = 25°C, NCO enabled, FIR filter enabled, unless otherwise noted. In the FFT plots that follow,
the location of the second and third harmonics is noted when they fall in the pass band of the filter.
–60
SECOND HARMONIC
THIRD HARMONIC
THIRD HARMONIC
–120
–120
10
15
20
25
30
35
FREQUENCY (MHz)
–140
06709-018
5
0
0
AMPLITUDE (dBFS)
–60
–80
20
25
30
35
FREQUENCY (MHz)
–140
06709-019
15
0
0
AMPLITUDE (dBFS)
–60
THIRD HARMONIC
–80
15
20
25
FREQUENCY (MHz)
30
35
06709-020
–120
10
30
35
–80
–120
5
25
–60
–100
0
20
–40
–100
–140
15
150MSPS
332.1MHz @ –1dBFS
SNR = 71.7dBc (72.7dBFS)
SFDR = 95.0dBc
fNCO = 321.5MHz
–20
–40
10
Figure 23. AD6655-150 Single-Tone FFT with fIN = 220.1 MHz,
fNCO = 205 MHz
150MSPS
140.1MHz @ –1dBFS
SNR = 74.3dBc (75.3dBFS)
SFDR = 83.3dBc
fNCO = 56MHz
–20
5
FREQUENCY (MHz)
Figure 20. AD6655-150 Single-Tone FFT with fIN = 30.3 MHz, fNCO = 24 MHz
0
35
THIRD HARMONIC
–120
10
30
–80
–120
5
25
–60
–100
0
20
–40
–100
–140
15
150MSPS
220.1MHz @ –1dBFS
SNR = 71.8dBc (72.8dBFS)
SFDR = 81.4dBc
fNCO = 205MHz
–20
–40
10
Figure 22. AD6655-150 Single-Tone FFT with fIN = 140.1 MHz,
fNCO = 126 MHz
150MSPS
30.3MHz @ –1dBFS
SNR = 74.8dBc (75.8dBFS)
SFDR = 100dBc
fNCO = 24MHz
–20
5
FREQUENCY (MHz)
Figure 19. AD6655-150 Single-Tone FFT with fIN = 2.4 MHz, fNCO = 18.75 MHz
0
SECOND HARMONIC
–80
–100
0
AMPLITUDE (dBFS)
–60
–100
–140
AMPLITUDE (dBFS)
–40
06709-021
AMPLITUDE (dBFS)
–40
–80
150MSPS
140.1MHz @ –1dBFS
SNR = 73.7dBc (74.7dBFS)
SFDR = 82.8dBc
fNCO = 126MHz
–20
06709-022
–20
AMPLITUDE (dBFS)
0
150MSPS
2.4MHz @ –1dBFS
SNR = 74.7dBc (75.7dBFS)
SFDR = 86.5dBc
fNCO = 18.75MHz
Figure 21. AD6655-150 Single-Tone FFT with fIN = 70.1 MHz, fNCO = 56 MHz
Rev. B | Page 24 of 84
–140
0
5
10
15
20
25
30
35
FREQUENCY (MHz)
Figure 24. AD6655-150 Single-Tone FFT with fIN = 332.1 MHz,
fNCO = 321.5 MHz
06709-023
0
Data Sheet
–20
AMPLITUDE (dBFS)
–40
SECOND HARMONIC
–60
THIRD HARMONIC
–80
THIRD HARMONIC
–80
–120
–120
5
20
15
10
25
30
35
FREQUENCY (MHz)
Figure 25. AD6655-150 Single-Tone FFT with fIN = 445.1 MHz, fNCO = 429 MHz
0
–140
0
10
15
AMPLITUDE (dBFS)
SECOND HARMONIC
THIRD HARMONIC
–100
25
30
125MSPS
140.1MHz @ –1dBFS
SNR = 74.1dBc (75.1dBFS)
SFDR = 90.3dBc
fNCO = 142MHz
–20
–60
20
Figure 28. AD6655-125 Single-Tone FFT with fIN = 70.3 MHz, fNCO = 78 MHz
0
–40
–80
5
FREQUENCY (MHz)
125MSPS
2.4MHz @ –1dBFS
SNR = 74.5dBc (75.5dBFS)
SFDR = 87.8dBc
fNCO = 15.75MHz
–20
–40
–60
THIRD HARMONIC
–80
–100
–120
0
5
15
10
20
25
30
FREQUENCY (MHz)
Figure 26. AD6655-125 Single-Tone FFT with fIN =2.4 MHz, fNCO = 15.75 MHz
0
–140
06709-025
–140
0
10
15
AMPLITUDE (dBFS)
THIRD
HARMONIC
–80
–100
30
125MSPS
220.1MHz @ –1dBFS
SNR = 73.4dBc (74.4dBFS)
SFDR = 90.2dBc
fNCO = 231MHz
–20
–60
25
Figure 29. AD6655-125 Single-Tone FFT with fIN = 140.1 MHz, fNCO = 142 MHz
0
–40
20
FREQUENCY (MHz)
125MSPS
30.3MHz @ –1dBFS
SNR = 74.7dBc (75.7dBFS)
SFDR = 89.6dBc
fNCO = 21MHz
–20
5
06709-028
–120
–40
–60
–80
–100
–120
–120
0
5
10
15
20
FREQUENCY (MHz)
25
30
–140
06709-026
–140
Figure 27. AD6655-125 Single-Tone FFT with fIN = 30.3 MHz, fNCO = 21 MHz
0
5
10
15
20
FREQUENCY (MHz)
25
30
06709-029
AMPLITUDE (dBFS)
–60
–100
–140
AMPLITUDE (dBFS)
–40
–100
0
125MSPS
70.3MHz @ –1dBFS
SNR = 74.6dBc (75.6dBFS)
SFDR = 86.1dBc
fNCO = 78MHz
–20
06709-024
AMPLITUDE (dBFS)
0
150MSPS
445.1MHz @ –1dBFS
SNR = 67.4dBc (65.4dBFS)
SFDR = 74.1dBc
fNCO = 429MHz
06709-027
0
AD6655
Figure 30. AD6655-125 Single-Tone FFT with fIN = 220.1 MHz, fNCO = 231 MHz
Rev. B | Page 25 of 84
AD6655
Data Sheet
95
120
80
90
SFDR (dBFS)
SFDR = +85°C
85
SNR (dBFS)
SNR/SFDR (dBc)
SNR/SFDR (dBc AND dBFS)
100
60
40
SFDR (dBc)
SFDR = +25°C
80
SFDR = –40°C
75
SNR = +25°C
SNR = +85°C
SNR = –40°C
70
85dB
REFERENCE LINE
20
65
–80
–70
–60
–50
–40
–30
–20
–10
0
INPUT AMPLITUDE (dBFS)
60
06709-030
0
–90
Figure 31. AD6655-150 Single-Tone SNR/SFDR vs. Input Amplitude (AIN) with
fIN = 2.4 MHz, fNCO = 18.75 MHz
0
50
100
150
200
250
300
350
400
450
INPUT FREQUENCY (MHz)
06709-033
SNR (dBc)
Figure 34. AD6655-125 Single-Tone SNR/SFDR vs. Input Frequency (fIN) and
Temperature with DRVDD = 3.3 V
120
–1.5
0.5
–2.0
0.4
GAIN ERROR (%FSR)
SNR/SFDR (dBc AND dBFS)
SNR (dBFS)
80
60
SFDR (dBc)
40
85dB
REFERENCE LINE
–2.5
0.3
OFFSET
–3.0
0.2
GAIN
0.1
–3.5
20
OFFSET ERROR (%FSR)
SFDR (dBFS)
100
SNR (dBc)
–70
–60
–50
–40
–30
–20
–10
0
INPUT AMPLITUDE (dBFS)
–4.0
–40
0
20
40
60
80
TEMPERATURE (°C)
Figure 32. AD6655-150 Single-Tone SNR/SFDR vs. Input Amplitude (AIN) with
fIN = 98.12 MHz, fNCO = 100.49 MHz
Figure 35. AD6655-150 Gain and Offset vs. Temperature
95
0
90
–20
SFDR/IMD3 (dBc AND dBFS)
SFDR = +85°C
85
SFDR = +25°C
80
SFDR = –40°C
75
SNR = +25°C
SNR = +85°C
SNR = –40°C
70
SFDR (dBc)
–40
IMD3 (dBc)
–60
–80
IMD3 (dBFS)
SFDR (dBFS)
–100
65
0
50
100
150
200
250
300
INPUT FREQUENCY (MHz)
350
400
450
–120
–90
06709-032
60
Figure 33. AD6655-125 Single-Tone SNR/SFDR vs. Input Frequency (fIN) and
Temperature with DRVDD = 1.8 V
–78
–66
–54
–42
–30
INPUT AMPLITUDE (dBFS)
–18
–6
06709-035
SNR/SFDR (dBc)
0
–20
06709-034
–80
06709-031
0
–90
Figure 36. AD6655-150 Two-Tone SFDR/IMD3 vs. Input Amplitude (AIN) with
fIN1 = 29.12 MHz, fIN2 = 32.12 MHz, fS = 150 MSPS, fNCO = 22 MHz
Rev. B | Page 26 of 84
Data Sheet
AD6655
0
0
–20
SFDR (dBc)
AMPLITUDE (dBFS)
–40
IMD3 (dBc)
–60
–80
–40
–60
–80
–100
IMD3 (dBFS)
SFDR (dBFS)
–100
–120
–66
–54
–42
–30
–18
–6
INPUT AMPLITUDE (dBFS)
–140
0
–20
–20
–40
–40
AMPLITUDE (dBFS)
0
–120
06709-037
–120
5
10
15
20
25
0
7.5
15.0
22.5
30.0
37.5
FREQUENCY (MHz)
Figure 41. AD6655-150 Noise Power Ratio (NPR)
95
150MSPS
29.12MHz @ –7dBFS
32.12MHz @ –7dBFS
SFDR = 89.1dBc (96.1dBFS)
fNCO = 22MHz
SFDR (dBc)
–40
SNR/SFDR (dBc)
AMPLITUDE (dBFS)
35
–140
30
Figure 38. AD6655-125, Two 64k WCDMA Carriers with fIN = 170 MHz,
fS = 122.88 MHz, fNCO = 168.96 MHz
–20
30
NPR = 64.5dBc
NOTCH @ 18.5MHz
NOTCH WIDTH = 3MHz
FREQUENCY (MHz)
0
25
–80
–100
0
20
–60
–100
–140
15
Figure 40. AD6655-150 Two Tone FFT with fIN1 = 169.12 MHz,
fIN2 = 172.12 MHz, fS = 150 MSPS, fNCO = 177 MHz
0
–80
10
FREQUENCY (MHz)
Figure 37. AD6655-150 Two-Tone SFDR/IMD3 vs. Input Amplitude (AIN) with
fIN1 = 169.12 MHz, fIN2 = 172.12 MHz, fS = 150 MSPS, fNCO = 177 MHz
–60
5
06709-039
–78
06709-036
–120
–90
06709-040
SFDR/IMD3 (dBc AND dBFS)
–20
AMPLITUDE (dBFS)
150MSPS
169.12MHz @ –7dBFS
172.12MHz @ –7dBFS
SFDR = 85.5dBc (92.5dBFS)
fNCO = 177MHz
–60
–80
85
SNR (dBc)
75
–100
0
5
10
15
20
25
FREQUENCY (MHz)
30
35
65
06709-038
–140
Figure 39. AD6655-150 Two-Tone FFT with fIN1 = 29.12 MHz, fIN2 = 32.12 MHz,
fS = 150 MSPS, fNCO = 22 MHz
06709-041
–120
0
25
50
75
100
125
150
SAMPLE RATE (MSPS)
Figure 42. AD6655-150 Single-Tone SNR/SFDR vs. Sample Rate (fs) with
fIN = 2.3 MHz
Rev. B | Page 27 of 84
AD6655
Data Sheet
12
90
0.85 LSB rms
85
SFDR
8
SNR/SFDR (dBc)
6
4
N–3
N–2
N–1
N
N+1
N+2
SNR
65
0.2
06709-042
0
N+3
OUTPUT CODE
Figure 43. AD6655 Grounded Input Histogram
85
SFDR DCS ON
SFDR DCS OFF
80
SNR DCS ON
75
40
50
60
DUTY CYCLE (%)
70
80
06709-043
SNR DCS OFF
30
0.4
0.6
0.8
1.0
1.2
INPUT COMMON-MODE VOLTAGE (V)
1.4
Figure 45. AD6655-150 SNR/SFDR vs. Input Common Mode (VCM) with
fIN = 30.3 MHz, fNCO = 45 MHz
90
SNR/SFDR (dBc)
75
70
2
70
20
80
06709-044
NUMBER OF HITS (1M)
10
Figure 44. AD6655-150 SNR/SFDR vs. Duty Cycle with fIN = 30.3 MHz,
fNCO = 45 MHz
Rev. B | Page 28 of 84
Data Sheet
AD6655
THEORY OF OPERATION
The AD6655 has two analog input channels, two decimating
channels, and two digital output channels. The intermediate
frequency (IF) input signal passes through several stages before
appearing at the output port(s) as a filtered, decimated digital
signal.
The dual ADC design can be used for diversity reception of
signals, where the ADCs operate identically on the same carrier
but from two separate antennae. The ADCs can also be operated
with independent analog inputs. The user can sample any fS/2
frequency segment from dc to 150 MHz using appropriate lowpass or band-pass filtering at the ADC inputs with little loss
in ADC performance. Operation to 450 MHz analog input is
permitted but occurs at the expense of increased ADC noise and
distortion.
In nondiversity applications, the AD6655 can be used as a baseband receiver, where one ADC is used for I input data, and the
other is used for Q input data.
Synchronization capability is provided to allow synchronized
timing between multiple channels or multiple devices. The
NCO phase can be set to produce a known offset relative to
another channel or device.
Programming and control of the AD6655 are accomplished
using a 3-bit SPI-compatible serial interface.
ADC ARCHITECTURE
ANALOG INPUT CONSIDERATIONS
The analog input to the AD6655 is a differential switchedcapacitor SHA that has been designed for optimum performance
while processing a differential input signal.
The clock signal alternatively switches the SHA between sample
mode and hold mode (see Figure 46). When the SHA is switched
into sample mode, the signal source must be capable of charging
the sample capacitors and settling within 1/2 of a clock cycle.
A small resistor in series with each input can help reduce the
peak transient current required from the output stage of the
driving source. A shunt capacitor can be placed across the inputs
to provide dynamic charging currents. This passive network creates
a low-pass filter at the ADC input; therefore, the precise values
are dependent on the application.
In IF undersampling applications, any shunt capacitors should be
reduced. In combination with the driving source impedance,
the shunt capacitors limit the input bandwidth. Refer to Application Note AN-742, Frequency Domain Response of SwitchedCapacitor ADCs; Application Note AN-827, A Resonant Approach
to Interfacing Amplifiers to Switched-Capacitor ADCs; and the
Analog Dialogue article, “Transformer-Coupled Front-End for
Wideband A/D Converters,” for more information on this subject
(see www.analog.com). In general, the precise values are dependent
on the application.
S
AD6655 architecture consists of a front-end sample-and-hold
amplifier (SHA) followed by a pipelined, switched-capacitor ADC.
The quantized outputs from each stage are combined into a final
14-bit result in the digital correction logic. The pipelined architecture permits the first stage to operate on a new input sample
and the remaining stages to operate on the preceding samples.
Sampling occurs on the rising edge of the clock.
The input stage of each channel contains a differential SHA that
can be ac- or dc-coupled in differential or single-ended modes.
The output staging block aligns the data, corrects errors, and
passes the data to the output buffers. The output buffers are
powered from a separate supply, allowing adjustment of the
output voltage swing. During power-down, the output buffers
go into a high impedance state.
CS
VIN+
CPIN, PAR
S
H
CS
VIN–
CH
CPIN, PAR
S
06709-048
Each stage of the pipeline, excluding the last, consists of a low
resolution flash ADC connected to a switched-capacitor digitalto-analog converter (DAC) and an interstage residue amplifier
(MDAC). The residue amplifier magnifies the difference between
the reconstructed DAC output and the flash input for the next
stage in the pipeline. One bit of redundancy is used in each stage
to facilitate digital correction of flash errors. The last stage
simply consists of a flash ADC.
CH
S
Figure 46. Switched-Capacitor SHA Input
For best dynamic performance, the source impedances driving
VIN+ and VIN− should be matched such that common-mode
settling errors are symmetrical. These errors are reduced by the
common-mode rejection of the ADC.
An internal differential reference buffer creates positive and
negative reference voltages that define the input span of the
ADC core. The output common mode of the reference buffer is
set to VCMREF (approximately 1.6 V).
Input Common Mode
The analog inputs of the AD6655 are not internally dc biased.
In ac-coupled applications, the user must provide this bias
externally. Setting the device so that VCM = 0.55 × AVDD is
recommended for optimum performance, but the device functions
over a wider range with reasonable performance (see Figure 45).
Rev. B | Page 29 of 84
AD6655
Data Sheet
An on-board common-mode voltage reference is included in
the design and is available from the CML pin. Optimum performance is achieved when the common-mode voltage of the analog
input is set by the CML pin voltage (typically 0.55 × AVDD).
The signal characteristics must be considered when selecting
a transformer. Most RF transformers saturate at frequencies
below a few megahertz (MHz). Excessive signal power can also
cause core saturation, which leads to distortion.
Differential Input Configurations
At input frequencies in the second Nyquist zone and above, the
noise performance of most amplifiers is not adequate to achieve
the true SNR performance of the AD6655. For applications where
SNR is a key parameter, differential double balun coupling is
the recommended input configuration (see Figure 49).
Optimum performance is achieved while driving the AD6655
in a differential input configuration. For baseband applications,
the AD8138, ADA4937-2, and ADA4938-2 differential drivers
provide excellent performance and a flexible interface to the
ADC. The output common-mode voltage of the AD8138 is
easily set with the CML pin of the AD6655 (see Figure 47), and
the driver can be configured in a Sallen-Key filter topology to
provide band limiting of the input signal.
An alternative to using a transformer-coupled input at frequencies
in the second Nyquist zone is to use the AD8352 differential
driver is shown in Figure 50. See the AD8352 data sheet for
more information. In addition, if the application requires an
amplifier with variable gain, the AD8375 or AD8376 digital
variable gain amplifiers (DVGAs) provide good performance
driving the AD6655.
499Ω
R
AVDD
523Ω
AD6655
C
AD8138
0.1µF
VIN+
499Ω
R
CML
VIN–
In any configuration, the value of the shunt capacitor, C, is
dependent on the input frequency and source impedance and
may need to be reduced or removed. Table 14 displays recommended values to set the RC network. However, these values are
dependent on the input signal and should be used only as a
starting guide.
06709-049
49.9Ω
1V p-p
499Ω
Figure 47. Differential Input Configuration Using the AD8138
For baseband applications where SNR is a key parameter,
differential transformer coupling is the recommended input
configuration. An example is shown in Figure 48. To bias the
analog input, the CML voltage can be connected to the center
tap of the secondary winding of the transformer.
Table 14. Example RC Network
Frequency Range
(MHz)
0 to 70
70 to 200
200 to 300
>300
R
VIN+
2V p-p
49.9Ω
C
AD6655
R
CML
06709-050
VIN–
R Series
(Ω, Each)
33
33
15
15
0.1µF
Figure 48. Differential Transformer-Coupled Configuration
0.1µF
0.1µF
R
2V p-p
VIN+
25Ω
S
P
S
25Ω
0.1µF
AD6655
C
0.1µF
R
CML
VIN–
06709-051
PA
Figure 49. Differential Double Balun Input Configuration
VCC
0Ω
16
0.1µF
8, 13
1
11
0.1µF
2
CD
RD
RG
3
ANALOG INPUT
0.1µF 0Ω
VIN+
200Ω
AD8352
10
4
5
R
0.1µF
200Ω
C
R
AD6655
VIN–
CML
14
0.1µF
0.1µF
Figure 50. Differential Input Configuration Using the AD8352
Rev. B | Page 30 of 84
06709-052
0.1µF
ANALOG INPUT
C Differential
(pF)
15
5
5
Open
Data Sheet
AD6655
Single-Ended Input Configuration
A single-ended input can provide adequate performance in
cost-sensitive applications. In this configuration, SFDR and
distortion performance degrade due to the large input commonmode swing. If the source impedances on each input are matched,
there should be little effect on SNR performance. Figure 51 shows
a typical single-ended input configuration.
This puts the reference amplifier in a noninverting mode with
the VREF output defined as follows:
R2 
VREF = 0.5 × 1 +

R1 

The input range of the ADC always equals twice the voltage at
the reference pin for either an internal or an external reference.
AVDD
10µF
VIN+A/VIN+B
1kΩ
VIN–A/VIN–B
R
49.9Ω
0.1µF
1kΩ
AVDD
1kΩ
10µF
0.1µF
ADC
CORE
AD6655
C
R
VIN–
1kΩ
VREF
06709-053
2V p-p
VIN+
1.0µF
0.1µF
SELECT
LOGIC
Figure 51. Single-Ended Input Configuration
SENSE
VOLTAGE REFERENCE
06709-054
0.5V
A stable and accurate voltage reference is built into the AD6655.
The input range can be adjusted by varying the reference voltage
applied to the AD6655, using either the internal reference or an
externally applied reference voltage. The input span of the ADC
tracks reference voltage changes linearly. The various reference
modes are summarized in the sections that follow. The Reference
Decoupling section describes the best practices PCB layout of
the reference.
AD6655
Figure 52. Internal Reference Configuration
VIN+A/VIN+B
VIN–A/VIN–B
Internal Reference Connection
ADC
CORE
A comparator within the AD6655 detects the potential at the
SENSE pin and configures the reference into four possible modes,
which are summarized in Table 15. If SENSE is grounded, the
reference amplifier switch is connected to the internal resistor
divider (see Figure 52), setting VREF to 1.0 V. Connecting the
SENSE pin to VREF switches the reference amplifier output to
the SENSE pin, completing the loop and providing a 0.5 V
reference output. If a resistor divider is connected externally
to the chip, as shown in Figure 53, the switch again sets to the
SENSE pin.
VREF
1.0µF
0.1µF
R2
SELECT
LOGIC
SENSE
AD6655
Figure 53. Programmable Reference Configuration
Table 15. Reference Configuration Summary
Selected Mode
External Reference
SENSE Voltage
AVDD
Resulting VREF (V)
N/A
Resulting Differential
Span (V p-p)
2 × external reference
Internal Fixed Reference
VREF
0.5
1.0
Programmable Reference
0.2 V to VREF
R2  (see Figure 53)

0.5 × 1 +

R1 

2 × VREF
Internal Fixed Reference
AGND to 0.2 V
1.0
2.0
Rev. B | Page 31 of 84
06709-055
0.5V
R1
AD6655
Data Sheet
If the internal reference of the AD6655 is used to drive multiple
converters to improve gain matching, the loading of the reference
by the other converters must be considered. Figure 54 depicts
how the internal reference voltage is affected by loading.
AVDD
1.2V
CLK+
CLK–
0
2pF
06709-058
2pF
–0.25
VREF = 1.0V
Figure 56. Equivalent Clock Input Circuit
–0.50
Clock Input Options
The AD6655 has a very flexible clock input structure. Clock
input can be a CMOS, LVDS, LVPECL, or sine wave signal.
Regardless of the type of signal being used, clock source jitter
is of the most concern, as described in the Jitter Considerations
section.
–0.75
–1.00
–1.25
0
0.5
1.0
2.0
1.5
LOAD CURRENT (mA)
06709-056
Figure 54. VREF Accuracy vs. Load
External Reference Operation
The use of an external reference may be necessary to enhance
the gain accuracy of the ADC or improve thermal drift characteristics. Figure 55 shows the typical drift characteristics of the
internal reference in both 1.0 V and 0.5 V modes.
2.5
Figure 57 and Figure 58 show two preferred methods for clocking
the AD6655 (at clock rates to up to 625 MHz). A low jitter clock
source is converted from a single-ended signal to a differential
signal using an RF transformer. The back-to-back Schottky diodes
across the transformer secondary limit clock excursions into the
AD6655 to approximately 0.8 V p-p differential. This helps prevent
the large voltage swings of the clock from feeding through to other
portions of the AD6655, while preserving the fast rise and fall times
of the signal, which are critical to a low jitter performance.
REFERENCE VOLTAGE ERROR (mV)
2.0
1.5
Mini-Circuits®
ADT1–1WT, 1:1Z
0.1µF
XFMR
0.1µF
CLOCK
INPUT
1.0
CLK+
100Ω
50Ω
ADC
AD6655
0.1µF
0.5
CLK–
0
SCHOTTKY
DIODES:
HSMS2822
0.1µF
–0.5
06709-059
REFERENCE VOLTAGE ERROR (%)
VREF = 0.5V
Figure 57. Transformer Coupled Differential Clock (Up to 200 MHz)
–1.0
–1.5
–2.0
1nF
0
20
40
TEMPERATURE (°C)
60
80
CLOCK
INPUT
0.1µF
CLK+
50Ω
ADC
AD6655
0.1µF
1nF
Figure 55. Typical VREF Drift
When the SENSE pin is tied to AVDD, the internal reference is
disabled, allowing the use of an external reference. An internal
reference buffer loads the external reference with an equivalent
6 kΩ load (see Figure 18). The internal buffer generates the positive
and negative full-scale references for the ADC core. Therefore,
the external reference must be limited to a maximum of 1.0 V.
CLOCK INPUT CONSIDERATIONS
CLK–
SCHOTTKY
DIODES:
HSMS2822
06709-157
–20
06709-057
–2.5
–40
Figure 58. Balun-Coupled Differential Clock (Up to 625 MHz)
If a low jitter clock source is not available, another option is to
ac couple a differential PECL signal to the sample clock input
pins as shown in Figure 59. The AD9510/AD9511/AD9512/
AD9513/AD9514/AD9515/AD9516 clock drivers offer excellent
jitter performance.
For optimum performance, the AD6655 sample clock inputs,
CLK+ and CLK−, should be clocked with a differential signal.
The signal is typically ac-coupled into the CLK+ and CLK− pins
via a transformer or capacitors. These pins are biased internally
(see Figure 56) and require no external bias.
Rev. B | Page 32 of 84
Data Sheet
AD6655
Input Clock Divider
0.1µF
0.1µF
CLOCK
INPUT
CLK+
0.1µF
0.1µF
ADC
AD6655
CLK–
50kΩ
240Ω
50kΩ
240Ω
06709-060
CLOCK
INPUT
100Ω
AD951x
PECL DRIVER
Figure 59. Differential PECL Sample Clock (Up to 625 MHz)
A third option is to ac-couple a differential LVDS signal to the
sample clock input pins, as shown in Figure 60. The AD9510/
AD9511/AD9512/AD9513/AD9514/AD9515/AD9516 clock
drivers offer excellent jitter performance.
The AD6655 contains an input clock divider with the ability to
divide the input clock by integer values between 1 and 8. If a divide
ratio other than 1 is selected, the duty cycle stabilizer is automatically enabled.
The AD6655 clock divider can be synchronized using the external
SYNC input. Bit 1 and Bit 2 of Register 0x100 allow the clock
divider to be resynchronized on every SYNC signal or only on
the first SYNC signal after the register is written. A valid SYNC
causes the clock divider to reset to its initial state. This synchronization feature allows multiple parts to have their clock dividers
aligned to guarantee simultaneous input sampling.
Clock Duty Cycle
0.1µF
0.1µF
CLOCK
INPUT
CLK+
0.1µF
0.1µF
ADC
AD6655
CLK–
50kΩ
06709-061
CLOCK
INPUT
100Ω
AD951x
LVDS DRIVER
50kΩ
Figure 60. Differential LVDS Sample Clock (Up to 625 MHz)
In some applications, it may be acceptable to drive the sample
clock inputs with a single-ended CMOS signal. In such applications, the CLK+ pin should be driven directly from a CMOS
gate, and the CLK− pin should be bypassed to ground with a
0.1 µF capacitor in parallel with a 39 kΩ resistor (see Figure 61).
CLK+ can be driven directly from a CMOS gate. Although the
CLK+ input circuit supply is AVDD (1.8 V), this input is designed
to withstand input voltages of up to 3.6 V, making the selection
of the drive logic voltage very flexible.
VCC
0.1µF
CLOCK
INPUT
50Ω
1kΩ
OPTIONAL
0.1µF
100Ω
AD951x
CMOS DRIVER
1kΩ
CLK+
ADC
AD6655
CLK–
39kΩ
06709-062
0.1µF
Figure 61. Single-Ended 1.8 V CMOS Sample Clock (Up to 150 MSPS)
Typical high speed ADCs use both clock edges to generate a
variety of internal timing signals and, as a result, may be sensitive to
clock duty cycle. Commonly, a ±5% tolerance is required on the
clock duty cycle to maintain dynamic performance characteristics.
The AD6655 contains a duty cycle stabilizer (DCS) that retimes
the nonsampling (falling) edge, providing an internal clock
signal with a nominal 50% duty cycle. This allows the user to
provide a wide range of clock input duty cycles without affecting
the performance of the AD6655. Noise and distortion performance
are nearly flat for a wide range of duty cycles with the DCS on, as
shown in Figure 44.
Jitter on the rising edge of the input clock is still of paramount
concern and is not easily reduced by the internal stabilization
circuit. The duty cycle control loop does not function for clock
rates less than 20 MHz nominally. The loop has a time constant
associated with it that must be considered when the clock rate
can change dynamically. A wait time of 1.5 µs to 5 µs is required
after a dynamic clock frequency increase or decrease before the
DCS loop is relocked to the input signal. During the time
period that the loop is not locked, the DCS loop is bypassed,
and internal device timing is dependent on the duty cycle of the
input clock signal. In such applications, it may be appropriate to
disable the duty cycle stabilizer. In all other applications, enabling
the DCS circuit is recommended to maximize ac performance.
Jitter Considerations
High speed, high resolution ADCs are sensitive to the quality of
the clock input. The degradation in SNR at a given input
frequency (fIN) due to jitter (tJ) can be calculated by
VCC
50Ω
1kΩ
AD951x
CMOS DRIVER
OPTIONAL 0.1µF
100Ω
1kΩ
0.1µF
CLK+
ADC
AD6655
CLK–
Figure 62. Single-Ended 3.3 V CMOS Sample Clock (Up to 150 MSPS)
SNRHF = −10 log[(2π × fIN × tJRMS)2 + 10 ( / SNRLF /10) ]
06709-063
0.1µF
CLOCK
INPUT
In the equation, the rms aperture jitter represents the rootmean-square of all jitter sources, which include the clock input,
the analog input signal, and the ADC aperture jitter specification.
IF undersampling applications are particularly sensitive to jitter,
as shown in Figure 63.
Rev. B | Page 33 of 84
AD6655
Data Sheet
75
0.6
1.50
0.05ps
TOTAL POWER
MEASURED
70
1.25
0.5
0.50ps
1.00ps
1.50ps
100
0.1
IDRVDD
1000
INPUT FREQUENCY (MHz)
0
0
50
25
100
75
125
0
150
06709-065
10
0.2
0.25
2.00ps
2.50ps
3.00ps
1
IDVDD
0.50
SAMPLE RATE (MSPS)
Figure 64. AD6655-150 Power and Current vs. Sample Rate
Figure 63. SNR vs. Input Frequency and Jitter
Refer to Application Note AN-501 and Application Note AN-756
for more information about jitter performance as it relates to ADCs
(see www.analog.com).
0.6
1.50
0.5
1.25
TOTAL POWER
TOTAL POWER (W)
The clock input should be treated as an analog signal in cases
where aperture jitter may affect the dynamic range of the AD6655.
Power supplies for clock drivers should be separated from the
ADC output driver supplies to avoid modulating the clock signal
with digital noise. Low jitter, crystal-controlled oscillators make
the best clock sources. If the clock is generated from another type
of source (by gating, dividing, or another method), it should be
retimed by the original clock at the last step.
1.00
0.4
IAVDD
0.75
0.3
0.50
0.2
IDVDD
0.25
SUPPLY CURRENT (A)
45
0.3
0.1
POWER DISSIPATION AND STANDBY MODE
IDRVDD
0
As shown in Figure 64 through Figure 67, the power dissipated
by the AD6655 is proportional to its sample rate. In CMOS
output mode, the digital power dissipation is determined
primarily by the strength of the digital drivers and the load on
each output bit. The maximum DRVDD current (IDRVDD) can be
calculated by
0
25
75
50
100
0
125
06709-166
50
0.75
SAMPLE RATE (MSPS)
Figure 65. AD6655-125 Power and Current vs. Sample Rate
0.5
1.25
IDRVDD = VDRVDD × CLOAD × fCLK × N
0.4
1.00
This maximum current occurs when every output bit switches
on every clock cycle, that is, a full-scale square wave at the
Nyquist frequency of fCLK/2. In practice, the DRVDD current
is established by the average number of output bits switching,
which is determined by the sample rate and the characteristics
of the analog input signal. Reducing the capacitive load presented
to the output drivers can minimize digital power consumption.
The data in Figure 64 through Figure 67 was taken using the same
operating conditions as those used for the Typical Performance
Characteristics, with a 5 pF load on each output driver.
TOTAL POWER (W)
TOTAL POWER
where N is the number of output bits (30, in the case of the
AD6655, assuming the FD bits are inactive).
0.75
0.3
IAVDD
0.50
0.2
IDVDD
0.25
Rev. B | Page 34 of 84
SUPPLY CURRENT (A)
55
0.4
0.1
IDRVDD
0
0
25
50
75
100
0
SAMPLE RATE (MSPS)
Figure 66. AD6655-105 Power and Current vs. Sample Rate
06709-167
60
IAVDD
1.00
06709-064
SNR (dBc)
65
SUPPLY CURRENT (A)
TOTAL POWER (W)
0.20ps
Data Sheet
AD6655
1.00
0.4
0.3
TOTAL POWER
IAVDD
0.50
0.2
IDVDD
0.25
The output data format can be selected for either offset binary
or twos complement by setting the SCLK/DFS pin when operating
in the external pin mode (see Table 16). As detailed in Application
Note AN-877, Interfacing to High Speed ADCs via SPI, the data
format can be selected for offset binary, twos complement, or
gray code when using the SPI control.
SUPPLY CURRENT (A)
TOTAL POWER (W)
0.75
tions requiring the ADC to drive large capacitive loads or large
fanouts may require external buffers or latches.
0.1
Table 16. SCLK/DFS Mode Selection (External Pin Mode)
0
0
20
40
60
0
80
SAMPLE RATE (MSPS)
06709-168
IDRVDD
Figure 67. AD6655-80 Power and Current vs. Sample Rate
By asserting PDWN (either through the SPI port or by asserting
the PDWN pin high), the AD6655 is placed in power-down
mode. In this state, the ADC typically dissipates 2.5 mW.
During power-down, the output drivers are placed in a high
impedance state. Asserting the PDWN pin low returns the
AD6655 to its normal operating mode. Note that PDWN is
referenced to the digital output driver supply (DRVDD) and
should not exceed that supply voltage. PDWN can be driven
with 1.8 V logic, even when DRVDD is at 3.3 V.
Low power dissipation in power-down mode is achieved by
shutting down the reference, reference buffer, biasing networks,
and clock. Internal capacitors are discharged when entering
power-down mode and then must be recharged when returning
to normal operation. As a result, wake-up time is related to the
time spent in power-down mode, and shorter power-down
cycles result in proportionally shorter wake-up times.
When using the SPI port interface, the user can place the ADC
in power-down mode or standby mode. Standby mode allows
the user to keep the internal reference circuitry powered when
faster wake-up times are required. See the Memory Map Register
Description section and Application Note AN-877, Interfacing
to High Speed ADCs via SPI at www.analog.com for additional
details.
DIGITAL OUTPUTS
The AD6655 output drivers can be configured to interface with
1.8 V to 3.3 V CMOS logic families by matching DRVDD to the
digital supply of the interfaced logic. Alternatively, the AD6655
outputs can be configured for either ANSI LVDS or reduced
drive LVDS using a 1.8 V DRVDD supply.
In CMOS output mode, the output drivers are sized to provide
sufficient output current to drive a wide variety of logic families.
However, large drive currents tend to cause current glitches on
the supplies that may affect converter performance. Applica-
Voltage at Pin
AGND (default)
AVDD
SCLK/DFS
Offset binary
Twos complement
SDIO/DCS
DCS disabled
DCS enabled
Digital Output Enable Function (OEB)
The AD6655 has a flexible three-state ability for the digital output
pins. The three-state mode is enabled using the SMI SDO/OEB pin
or through the SPI interface. If the SMI SDO/OEB pin is low, the
output data drivers are enabled. If the SMI SDO/OEB pin is high,
the output data drivers are placed in a high impedance state.
This OEB function is not intended for rapid access to the data
bus. Note that OEB is referenced to the digital output driver
supply (DRVDD) and should not exceed that supply voltage.
OEB can be driven with 1.8 V logic even when DRVDD is at 3.3 V.
When using the SPI interface, the data and fast detect outputs of
each channel can be independently three-stated by using the
output enable bar bit (Bit 4) in Register 0x14.
Interleaved CMOS Mode
Setting Bit 5 in Register 0x14 enables interleaved CMOS output
mode. In this mode, output data is routed through Port A with
the ADC Channel A output data present on the rising edge of
DCO and the ADC Channel B output data present on the
falling edge of DCO.
Timing
The AD6655 provides latched data with a pipeline delay that is
dependent on which of the digital back end features are enabled.
Data outputs are available one propagation delay (tPD) after the
rising edge of the clock signal.
The length of the output data lines and loads placed on them
should be minimized to reduce transients within the AD6655.
These transients can degrade converter dynamic performance.
The lowest typical conversion rate of the AD6655 is 10 MSPS. At
clock rates below 10 MSPS, dynamic performance may degrade.
Data Clock Output (DCO)
The AD6655 also provides data clock output (DCO) intended for
capturing the data in an external register. Figure 2 through Figure 6
show a graphical timing description of the AD6655 output modes.
Rev. B | Page 35 of 84
AD6655
Data Sheet
Table 17. Output Data Format
Input (V)
VIN+ – VIN–
VIN+ – VIN–
VIN+ – VIN–
VIN+ – VIN–
VIN+ – VIN–
Condition (V)
< –VREF – 0.5 LSB
= –VREF
=0
= +VREF – 1.0 LSB
> +VREF – 0.5 LSB
Offset Binary Output Mode
00 0000 0000 0000
00 0000 0000 0000
10 0000 0000 0000
11 1111 1111 1111
11 1111 1111 1111
Rev. B | Page 36 of 84
Twos Complement Mode
10 0000 0000 0000
10 0000 0000 0000
00 0000 0000 0000
01 1111 1111 1111
01 1111 1111 1111
OR
1
0
0
0
1
Data Sheet
AD6655
DIGITAL DOWNCONVERTER
The AD6655 includes a digital processing section that provides
filtering and reduces the output data rate. This digital processing
section includes a numerically controlled oscillator (NCO),
a half-band decimating filter, an FIR filter, and a second coarse
NCO (fADC/8 fixed value) for output frequency translation. Each
of these processing blocks (except the decimating half-band
filter) has control lines that allow it to be independently enabled
and disabled to provide the desired processing function. The
digital downconverter can be configured to output either real data
or complex output data. These blocks can be configured in five
recommended combinations to implement different signal
processing functions.
DOWNCONVERTER MODES
a maximum usable bandwidth of 16.5 MHz when using the filter
in real mode (NCO bypassed) or a maximum usable bandwidth
of 33.0 MHz when using the filter in the complex mode (NCO
enabled).
The optional fixed-coefficient FIR filter provides additional
filtering capability to sharpen the half-band roll-off to enhance
the alias protection. It removes the negative frequency images
to avoid aliasing negative frequencies for real outputs.
fADC/8 FIXED-FREQUENCY NCO
A fixed fADC/8 NCO is provided to translate the filtered, decimated
signal from dc to fADC/8 to allow a real output. Figure 68 to
Figure 71 show an example of a 20 MHz input as it is processed
by the blocks of the AD6655.
Table 18. Downconverter Modes
Output Type
Real
Real
Complex
Complex
Real
–50
–24
–14
–4 0 4
14
24
50
Figure 68. Example AD6655 Real 20 MHz Bandwidth Input Signal Centered at
14 MHz (fADC = 100 MHz)
NUMERICALLY CONTROLLED OSCILLATOR (NCO)
Frequency translation is accomplished with an NCO. Each of
the two processing channels shares a common NCO. Amplitude
and phase dither can be enabled on chip to improve the noise and
spurious performance of the NCO. A phase offset word is available
to create a known phase relationship between multiple AD6655s.
Because the decimation filter prevents usage of half the Nyquist
spectrum, a means is needed to translate the sampled input
spectrum into the usable range of the decimation filter. To
achieve this, a 32-bit, fine tuning, complex NCO is provided.
This NCO/mixer allows the input spectrum to be tuned to dc,
where it can be effectively filtered by the subsequent filter
blocks to prevent aliasing.
–50
–38
–28
–18 –10
0
10
50
06709-067
NCO/Filter
Half-band filter only
Half-band filter and FIR filter
NCO and half-band filter
NCO, half-band filter, and FIR filter
NCO, half-band filter, FIR filter, and
fADC/8 NCO
Figure 69. Example AD6655 20 MHz Bandwidth Input Signal Tuned to DC
Using the NCO (NCO Frequency = 14 MHz)
–50
–38
–28
–18 –10
0
10
50
06709-068
Mode
1
2
3
4
5
06709-066
Table 18 details the recommended downconverter modes of
operation in the AD6655.
Figure 70. Example AD6655 20 MHz Bandwidth Input Signal with the
Negative Image Filtered by the Half-Band and FIR Filters
The goal of the AD6655 digital filter block is to allow the sample
rate to be reduced by a factor of 2 while rejecting aliases that fall
into the band of interest. The half-band filter is designed to operate
as either a low-pass or high-pass filter and to provide greater
than 100 dB of alias protection for 22% of the input rate of the
structure. For an ADC sample rate of 150 MSPS, this provides
–50
0.25
12.5
22.5
50
06709-069
HALF-BAND DECIMATING FILTER AND FIR FILTER
Figure 71. Example AD6655 20 MHz Bandwidth Input Signal Tuned to
fADC/8 for Real Output
Rev. B | Page 37 of 84
AD6655
Data Sheet
NUMERICALLY CONTROLLED OSCILLATOR (NCO)
FREQUENCY TRANSLATION
PHASE OFFSET
This processing stage comprises a digital tuner consisting of
a 32-bit complex numerically controlled oscillator (NCO). The
two channels of the AD6655 share a single NCO. The NCO is
optional and can be bypassed by clearing Bit 0 of Register 0x11D.
This NCO block accepts a real input from the ADC stage and
outputs a frequency translated complex (I and Q) output.
The NCO phase offset register at Address 0x122 and
Address 0x123 adds a programmable offset to the phase
accumulator of the NCO. This 16-bit register is interpreted as
a 16-bit unsigned integer. A 0x00 in this register corresponds
to no offset, and a 0xFFFF corresponds to an offset of 359.995°.
Each bit represents a phase change of 0.005°. This register allows
multiple NCOs to be synchronized to produce outputs with
predictable phase differences. Use the following equation to
calculate the NCO phase offset value:
The NCO frequency is programmed in Register 0x11E,
Register 0x11F, Register 0x120, and Register 0x121. These four
8-bit registers make up a 32-bit unsigned frequency programming
word. Frequencies between −CLK/2 and +CLK/2 are represented
using the following frequency words:



0x8000 0000 represents a frequency given by −CLK/2.
0x0000 0000 represents dc (frequency = 0 Hz).
0x7FFF FFFF represents CLK/2 − CLK/232.
NCO_FREQ  2

where:
NCO_PHASE is a decimal number equal to the 16-bit binary
number to be programmed at Register 0x122 and Register 0x123.
PHASE is the desired NCO phase in degrees.
NCO AMPLITUDE AND PHASE DITHER
Use the following equation to calculate the NCO frequency:
32
NCO_PHASE = 216 × PHASE/360
Mod( f , f CLK )
f CLK
where:
NCO_FREQ is a 32-bit twos complement number representing
the NCO frequency register.
f is the desired carrier frequency in hertz (Hz).
fCLK is the AD6655 ADC clock rate in hertz (Hz).
NCO SYNCHRONIZATION
The NCO block contains amplitude and phase dither to improve
the spurious performance. Amplitude dither improves performance by randomizing the amplitude quantization errors within
the angular-to-Cartesian conversion of the NCO. This option
reduces spurs at the expense of a slightly raised noise floor. With
amplitude dither enabled, the NCO has an SNR of >93 dB and
an SFDR of >115 dB. With amplitude dither disabled, the SNR
is increased to >96 dB at the cost of SFDR performance, which
is reduced to 100 dB. The NCO amplitude dither is recommended
and is enabled by setting Bit 1 of Register 0x11D.
The AD6655 NCOs within a single part or across multiple parts
can be synchronized using the external SYNC input. Bit 3 and
Bit 4 of Register 0x100 allow the NCO to be resynchronized on
every SYNC signal or only on the first SYNC signal after the
register is written. A valid SYNC causes the NCO to restart at
the programmed phase offset value.
Rev. B | Page 38 of 84
Data Sheet
AD6655
DECIMATING HALF-BAND FILTER AND FIR FILTER
0
–10
–20
–30
AMPLITUDE (dBc)
The goal of the AD6655 half-band digital filter is to allow the
sample rate to be reduced by a factor of 2 while rejecting aliases
that fall into the band of interest. This filter is designed to operate
as either a low-pass or a high-pass filter and to provide >100 dB
of alias protection for 11% of the input rate of the structure.
Used in conjunction with the NCO and the FIR filter, the halfband filter can provide an effective band-pass. For an ADC
sample rate of 150 MSPS, this provides a maximum usable
bandwidth of 33 MHz.
Table 19. Fixed Coefficients for Half-Band Filter
Decimal Coefficient
(20-Bit)
844
−6189
25080
−79170
321584
524287
HALF-BAND FILTER FEATURES
In the AD6655, the half-band filter cannot be disabled. The
filter can be set for a low-pass or high-pass response. For a highpass filter, Bit 1 of Register 0x103 should be set; for a low-pass
response, this bit should be cleared. The low-pass response of the
filter with respect to the normalized output rate is shown in
Figure 72, and the high-pass response is shown in Figure 73.
0
–10
–20
–110
0
0.1
0.2
0.3
0.4
FRACTION OF INPUT SAMPLE RATE
06709-071
–100
Figure 73. Half-Band Filter High-Pass Response
The half-band filter has a ripple of 0.000182 dB and a rejection
of 100 dB. For an alias rejection of 100 dB, the alias protected
bandwidth is 11% of the input sample rate. If both the I and the
Q paths are used, a complex bandwidth of 22% of the input rate
is available.
In the event of even Nyquist zone sampling, the half-band filter
can be configured to provide a spectral reversal. Setting Bit 2
high in Address 0x103 enables the spectral reversal feature.
The half-band decimation phase can be selected such that
the half-band filter starts on the first or second sample following
synchronization. This shifts the output from the half-band between
the two input sample clocks. The decimation phase can be set to
0 or 1, using Bit 3 of Register 0x103.
FIXED-COEFFICIENT FIR FILTER
Following the half-band filters is a 66-tap, fixed-coefficient FIR
filter. This filter is useful in providing extra alias protection for
the decimating half-band filter. It is a simple sum-of-products
FIR filter with 66 filter taps and 21-bit fixed coefficients. Note
that this filter does not decimate. The normalized coefficients
used in the implementation and the decimal equivalent value of
the coefficients are listed in Table 20.
The user can either select or bypass this filter, but the FIR filter
can be enabled only when the half-band filter is enabled. Writing
Logic 0 to the enable FIR filter bit (Bit 0) in Register 0x102
bypasses this fixed-coefficient filter. The filter is necessary when
using the final NCO with a real output; bypassing it when using
other configurations results in power savings.
–30
–40
–50
–60
–70
–80
–90
–100
–110
0
0.1
0.2
0.3
0.4
FRACTION OF INPUT SAMPLE RATE
06709-070
AMPLITUDE (dBc)
–70
–90
The 19-tap, symmetrical, fixed-coefficient half-band filter has low
power consumption due to its polyphase implementation. Table 19
lists the coefficients of the half-band filter. The normalized
coefficients used in the implementation and the decimal
equivalent value of the coefficients are also listed. Coefficients
not listed in Table 19 are 0s.
Normalized
Coefficient
0.0008049
−0.0059023
0.0239182
−0.0755024
0.3066864
0.5
–50
–60
–80
HALF-BAND FILTER COEFFICIENTS
Coefficient
Number
C0, C18
C2, C16
C4, C14
C6, C12
C8, C10
C9
–40
Figure 72. Half-Band Filter Low-Pass Response
Rev. B | Page 39 of 84
AD6655
Data Sheet
COMBINED FILTER PERFORMANCE
Normalized
Coefficient
0.0001826
0.0006824
0.0009298
0.0000458
−0.0012689
−0.0008345
0.0011806
0.0011387
−0.0018439
−0.0024557
0.0018063
0.0035825
−0.0021510
−0.0056810
0.0017405
0.0078602
−0.0013437
−0.0110626
−0.0000229
0.0146618
0.0018959
−0.0195594
−0.0053153
0.0255623
0.0104036
−0.0341468
−0.0192165
0.0471258
0.0354118
−0.0728111
−0.0768890
0.1607208
0.4396725
Decimal Coefficient
(21-Bit)
383
1431
1950
96
−2661
−1750
2476
2388
−3867
−5150
3788
7513
−4511
−11914
3650
16484
−2818
−23200
−48
30748
3976
−41019
−11147
53608
21818
−71611
−40300
98830
74264
−152696
−161248
337056
922060
SYNCHRONIZATION
The AD6655 half-band filters within a single part or across
multiple parts can be synchronized using the external SYNC
input. Bit 5 and Bit 6 of Register 0x100 allow the half-bands to
be resynchronized on every SYNC signal or only on the first
SYNC signal after the register is written. A valid SYNC causes
the half-band filter to restart at the programmed decimation
phase value.
The combined response of the half-band filter and the FIR filter
is shown in Figure 74. The act of bandlimiting the ADC data with
the half-band filter ideally provides a 3 dB improvement in the
SNR at the expense of the sample rate and available bandwidth
of the output data. As a consequence of finite math, additional
quantization noise is added to the system due to truncation in
the NCO and half-band. As a consequence of the digital filter
rejection of out-of-band noise (assuming no quantization in the
filters and with a white noise floor from the ADC), there should
be a 3.16 dB improvement in the ADC SNR. However, the added
quantization lessens improvement to about 2.66 dB.
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
–110
0
0.1
0.2
0.3
0.4
FRACTION OF INPUT SAMPLE RATE
06709-072
Coefficient
Number
C0, C65
C1, C64
C2, C63
C3, C62
C4, C61
C5, C60
C6, C59
C7, C58
C8, C57
C9, C56
C10, C55
C11, C54
C12, C53
C13, C52
C14, C51
C15, C50
C16, C49
C17, C48
C18, C47
C19, C46
C20, C45
C21, C44
C22, C43
C23, C42
C24, C41
C25, C40
C26, C39
C27, C38
C28, C37
C29, C36
C30, C35
C31, C34
C32, C33
AMPLITUDE (dBc)
Table 20. FIR Filter Coefficients
Figure 74. Half-Band Filter and FIR Filter Composite Response
FINAL NCO
The output of the 32-bit fine tuning NCO is complex and
typically centered in frequency around dc. This complex output
is carried through the stages of the half-band and FIR filters to
provide proper antialiasing filtering. The final NCO provides a
means to move this complex output signal away from dc so that
a real output can be provided from the AD6655. The final NCO,
if enabled, translates the output from dc to a frequency equal to
the ADC sampling frequency divided by 8 (fADC/8). This provides
the user a decimated output signal centered at fADC/8 in frequency.
Optionally, this final NCO can be bypassed, and the dc-centered
I and Q values can be output in an interleaved fashion.
Rev. B | Page 40 of 84
Data Sheet
AD6655
ADC OVERRANGE AND GAIN CONTROL
In receiver applications, it is desirable to have a mechanism
to reliably determine when the converter is about to be clipped.
The standard overflow indicator provides after-the-fact information on the state of the analog input that is of limited usefulness.
Therefore, it is helpful to have a programmable threshold below
full scale that allows time to reduce the gain before the clip actually
occurs. In addition, because input signals can have significant
slew rates, latency of this function is of major concern. Highly
pipelined converters can have significant latency. A good compromise is to use the output bits from the first stage of the ADC for
this function. Latency for these output bits is very low, and overall
resolution is not highly significant. Peak input signals are typically
between full scale and 6 dB to 10 dB below full scale. A 3-bit or
4-bit output provides adequate range and resolution for this
function.
Using the SPI port, the user can provide a threshold above which
an overrange output is active. As long as the signal is below that
threshold, the output should remain low. The fast detect outputs
can also be programmed via the SPI port so that one of the pins
functions as a traditional overrange pin for customers who
currently use this feature. In this mode, all 14 bits of the converter
are examined in the traditional manner, and the output is high
for the condition normally defined as overflow. In either mode,
the magnitude of the data is considered in the calculation of the
condition (but the sign of the data is not considered). The threshold
detection responds identically to positive and negative signals
outside the desired range (magnitude).
FAST DETECT OVERVIEW
The AD6655 contains circuitry to facilitate fast overrange
detection, allowing very flexible external gain control implementations. Each ADC has four fast detect (FD) output pins
that are used to output information about the current state of
the ADC input level. The function of these pins is programmable
via the fast detect mode select bits and the fast detect enable bit
in Register 0x104, allowing range information to be output from
several points in the internal data path. These output pins can
also be set up to indicate the presence of overrange or underrange
conditions, according to programmable threshold levels. Table 21
shows the six configurations available for the fast detect pins.
Table 21. Fast Detect Mode Select Bits Settings
Fast Detect
Mode Select bits
(Register 0x104[3:1])
000
001
010
011
100
101
Information Presented on
Fast Detect (FD) Pins of Each ADC1, 2
FD[3]
FD[2]
FD[1]
FD[0]
ADC fast magnitude (see Table 22)
OR
ADC fast magnitude
(see Table 23)
OR
F_LT
ADC fast
magnitude
(see Table 24)
C_UT
F_LT
ADC fast
magnitude
(see Table 24)
OR
C_UT
F_UT
F_LT
OR
F_UT
IG
DG
1
The fast detect pins are FD0A/FD0B to FD3A/FD3B for the CMOS mode
configuration and FD0+/FD0− to FD3+/FD3− for the LVDS mode
configuration.
2
See the ADC Overrange (OR) and Gain Switching sections for more
information about OR, C_UT, F_UT, F_LT, IG, and DG.
ADC FAST MAGNITUDE
When the fast detect output pins are configured to output the
ADC fast magnitude (that is, when the fast detect mode select
bits are set to 0b000), the information presented is the ADC
level from an early converter stage with a latency of only two
clock cycles in CMOS output modes. In LVDS output mode,
the fast detect bits have a latency of six cycles in all fast detect
modes. Using the fast detect output pins in this configuration
provides the earliest possible level indication information. Because
this information is provided early in the datapath, there is significant uncertainty in the level indicated. The nominal levels, along
with the uncertainty indicated by the ADC fast magnitude, are
shown in Table 22. Because the DCO is at one-half the sample
rate, the user can obtain all the fast detect information by sampling
the fast detect outputs on both the rising and falling edge of
DCO (see Figure 2 for timing information).
Table 22. ADC Fast Magnitude Nominal Levels
with Fast Detect Mode Select Bits = 000
ADC Fast
Magnitude on
FD[3:0] Pins
0000
0001
0010
0011
0100
0101
0110
0111
1000
Rev. B | Page 41 of 84
Nominal Input
Magnitude
Below FS (dB)
<−24
−24 to −14.5
−14.5 to −10
−10 to −7
−7 to −5
−5 to −3.25
−3.25 to −1.8
−1.8 to −0.56
−0.56 to 0
Nominal Input
Magnitude
Uncertainty (dB)
Minimum to −18.07
−30.14 to −12.04
−18.07 to −8.52
−12.04 to −6.02
−8.52 to −4.08
−6.02 to −2.5
−4.08 to −1.16
−2.5 to FS
−1.16 to 0
AD6655
Data Sheet
When the fast detect mode select bits are set to 0b001, 0b010,
or 0b011, a subset of the fast detect output pins are available.
In these modes, the fast detect output pins have a latency of six
clock cycles, and the greater of the two input samples is output
at the DCO rate. Table 23 shows the corresponding ADC input
levels when the fast detect mode select bits are set to 0b001 (that
is, when the ADC fast magnitude is presented on the FD[3:1] pins).
Table 23. ADC Fast Magnitude Nominal Levels
with Fast Detect Mode Select Bits = 001
ADC Fast
Magitude on
FD[2:0] Pins
000
001
010
011
100
101
110
111
Nominal Input
Magnitude
Below FS (dB)
<−24
−24 to −14.5
−14.5 to −10
−10 to −7
−7 to −5
−5 to −3.25
−3.25 to −1.8
−1.8 to 0
Nominal Input
Magnitude
Uncertainty (dB)
Minimum to −18.07
−30.14 to −12.04
−18.07 to −8.52
−12.04 to −6.02
−8.52 to −4.08
−6.02 to −2.5
−4.08 to −1.16
−2.5 to 0
Table 24. ADC Fast Magnitude Nominal Levels
with Fast Detect Mode Select Bits = 010 or 011
Nominal Input
Magnitude
Below FS (dB)
<−14.5
−14.5 to −7
−7 to −3.25
−3.25 to 0
The coarse upper threshold indicator is asserted if the ADC fast
magnitude input level is greater than the level programmed in
the coarse upper threshold register (Address 0x105[2:0]). This
value is compared with the ADC Fast Magnitude Bits[2:0]. The
coarse upper threshold output is output two clock cycles after
the level is exceeded at the input and, therefore, provides a fast
indication of the input signal level. The coarse upper threshold
levels are shown in Table 25. This indicator remains asserted for
a minimum of two ADC clock cycles or until the signal drops
below the threshold level.
Table 25. Coarse Upper Threshold Levels
When the fast detect mode select bits are set to 0b010 or 0b011
(that is, when ADC fast magnitude is presented on the FD[2:1]
pins), the LSB is not provided. The input ranges for this mode
are shown in Table 24.
ADC Fast
Magitude on
FD[2:1] Pins
00
01
10
11
Coarse Upper Threshold (C_UT)
Nominal Input
Magnitude
Uncertainty (dB)
Minimum to −12.04
−18.07 to −6.02
−8.52 to −2.5
−4.08 to 0
Coarse Upper Threshold
Register[2:0]
000
001
010
011
100
101
110
111
C_UT Is Active When Signal
Magnitude Below FS
Is Greater Than (dB)
<−24
−24
−14.5
−10
−7
−5
−3.25
−1.8
Fine Upper Threshold (F_UT)
The fine upper threshold indicator is asserted if the input
magnitude exceeds the value programmed in the fine upper
threshold register located in Register 0x106 and Register 0x107.
The 13-bit threshold register is compared with the signal magnitude at the output of the ADC. This comparison is subject to
the ADC clock latency but is accurate in terms of converter
resolution. The fine upper threshold magnitude is defined by
the following equation:
dBFS = 20 log(Threshold Magnitude/213)
ADC OVERRANGE (OR)
Fine Lower Threshold (F_LT)
The ADC overrange indicator is asserted when an overrange is
detected on the input of the ADC. The overrange condition is
determined at the output of the ADC pipeline and, therefore, is
subject to a latency of 12 ADC clock cycles. An overrange at the
input is indicated by this bit 12 clock cycles after it occurs.
The fine lower threshold indicator is asserted if the input magnitude is less than the value programmed in the fine lower threshold
register located at Register 0x108 and Register 0x109. The fine
lower threshold register is a 13-bit register that is compared with
the signal magnitude at the output of the ADC. This comparison
is subject to ADC clock latency but is accurate in terms of
converter resolution. The fine lower threshold magnitude
is defined by the following equation:
GAIN SWITCHING
The AD6655 includes circuitry that is useful in applications
either where large dynamic ranges exist or where gain ranging
converters are employed. This circuitry allows digital thresholds
to be set such that an upper threshold and a lower threshold can
be programmed. Fast detect mode select bits = 010 through fast
detect mode select bits = 101 support various combinations of
the gain switching options.
dBFS = 20 log(Threshold Magnitude/213)
The operation of the fine upper threshold and fine lower
threshold indicators is shown in Figure 75.
One such use is to detect when an ADC is about to reach full
scale with a particular input condition. The result is to provide
an indicator that can be used to quickly insert an attenuator that
prevents ADC overdrive.
Rev. B | Page 42 of 84
Data Sheet
AD6655
Increment Gain (IG) and Decrement Gain (DG)
with the magnitude at the output of the ADC. This comparison
is subject to the ADC clock latency but allows a finer, more
accurate comparison. The fine upper threshold magnitude is
defined by the following equation:
The increment gain and decrement gain indicators are intended
to be used together to provide information to enable external
gain control. The decrement gain indicator works in conjunction
with the coarse upper threshold bits, asserting when the input
magnitude is greater than the 3-bit value in the coarse upper
threshold register (Address 0x105). The increment gain indicator,
similarly, corresponds to the fine lower threshold bits except
that it is asserted only if the input magnitude is less than the
value programmed in the fine lower threshold register after the
dwell time elapses. The dwell time is set by the 16-bit dwell time
value located at Address 0x10A and Address 0x10B and is set in
units of ADC input clock cycles ranging from 1 to 65,535. The
fine lower threshold register is a 13-bit register that is compared
dBFS = 20 log(Threshold Magnitude/213)
The decrement gain output works from the ADC fast detect
output pins, providing a fast indication of potential overrange
conditions. The increment gain uses the comparison at the
output of the ADC, requiring the input magnitude to remain
below an accurate, programmable level for a predefined period
before signaling external circuitry to increase the gain.
The operation of the increment gain output and decrement gain
output is shown graphically in Figure 75.
UPPER THRESHOLD (COARSE OR FINE)
DWELL TIME
TIMER RESET BY
RISE ABOVE F_LT
FINE LOWER THRESHOLD
DWELL TIME
C_UT OR F_UT*
TIMER COMPLETES BEFORE
SIGNAL RISES ABOVE F_LT
F_LT
DG
*C_UT AND F_UT DIFFER ONLY IN ACCURACY AND LATENCY.
NOTE: OUTPUTS FOLLOW THE INSTANTEOUS SIGNAL LEVEL AND NOT THE ENVELOPE BUT ARE GUARANTEED ACTIVE FOR A MINIMUM OF 2 ADC CLOCK CYCLES.
Figure 75. Threshold Settings for C_UT, F_UT, F_LT, DG, and IG
Rev. B | Page 43 of 84
06709-073
IG
AD6655
Data Sheet
SIGNAL MONITOR
The signal monitor block provides additional information
about the signal being digitized by the ADC. The signal monitor
computes the rms input magnitude, the peak magnitude, and/or
the number of samples by which the magnitude exceeds a
particular threshold. Together, these functions can be used to
gain insight into the signal characteristics and to estimate the
peak/average ratio or even the shape of the complementary
cumulative distribution function (CCDF) curve of the input
signal. This information can be used to drive an AGC loop to
optimize the range of the ADC in the presence of real-world
signals.
current ADC input signal magnitude. This comparison continues
until the monitor period timer reaches a count of 1.
The signal monitor result values can be obtained from the part by
reading back internal registers at Address 0x116 to Address 0x11B,
using the SPI port or the signal monitor SPORT output. The output
contents of the SPI-accessible signal monitor registers are set via
the two signal monitor mode bits of the signal monitor control
register (Address 0x112). Both ADC channels must be configured
for the same signal monitor mode. Separate SPI-accessible, 20-bit
signal monitor result (SMR) registers are provided for each ADC
channel. Any combination of the signal monitor functions can
also be output to the user via the serial SPORT interface. These
outputs are enabled using the peak detector output enable, the
rms magnitude output enable, and the threshold crossing output
enable bits in the signal monitor SPORT control register
(Address 0x1111).
Figure 76 is a block diagram of the peak detector logic. The
SMR register contains the absolute magnitude of the peak
detected by the peak detector logic.
For each signal monitor measurement, a programmable signal
monitor period register (SMPR) controls the duration of the
measurement. This time period is programmed as the number
of input clock cycles in a 24-bit signal monitor period register
located at Address 0x113, Address 0x114, and Address 0x115.
This register can be programmed with a period from 128 samples
to 16.78 (224) million samples.
When the monitor period timer reaches a count of 1, the 13-bit
peak level value is transferred to the signal monitor holding
register (not accessible to the user), which can be read through
the SPI port or output through the SPORT serial interface. The
monitor period timer is reloaded with the value in the SMPR,
and the countdown is restarted. In addition, the magnitude of
the first input sample is updated in the peak level holding
register, and the comparison and update procedure, as
explained previously, continues.
FROM
MEMORY
MAP
POWER MONITOR
PERIOD REGISTER
TO
INTERRUPT
CONTROLLER
DOWN
COUNTER
IS COUNT = 1?
LOAD
CLEAR
MAGNITUDE
STORAGE
REGISTER
POWER MONITOR
HOLDING
REGISTER
LOAD
TO
MEMORY
MAP
LOAD
COMPARE
A>B
06709-074
FROM
INPUT
PORTS
Figure 76. ADC Input Peak Detector Block Diagram
RMS/MS MAGNITUDE MODE
PEAK DETECTOR MODE
In this mode, the root-mean-square (rms) or mean-square (ms)
magnitude of the input port signal is integrated (by adding an
accumulator) over a programmable time period (determined by
SMPR) to give the rms or ms magnitude of the input signal.
This mode is set by programming Logic 0 in the signal monitor
mode bits of the signal monitor control register or by setting the
rms magnitude output enable bit in the signal monitor SPORT
control register. The 24-bit SMPR, representing the period over
which integration is performed, must be programmed before
activating this mode.
The magnitude of the input port signal is monitored over a
programmable time period (determined by SMPR) to give the
peak value detected. This function is enabled by programming a
Logic 1 in the signal monitor mode bits of the signal monitor
control register or by setting the peak detector output enable bit
in the signal monitor SPORT control register. The 24-bit SMPR
must be programmed before activating this mode.
After enabling the rms/ms magnitude mode, the value in the
SMPR is loaded into a monitor period timer, and the countdown
is started immediately. Each input sample is converted to floatingpoint format and squared. It is then converted to 11-bit, fixedpoint format and added to the contents of the 24-bit accumulator.
The integration continues until the monitor period timer
reaches a count of 1.
After enabling this mode, the value in the SMPR is loaded into a
monitor period timer, and the countdown is started. The
magnitude of the input signal is compared with the value in the
internal peak level holding register (not accessible to the user),
and the greater of the two is updated as the current peak level.
The initial value of the peak level holding register is set to the
When the monitor period timer reaches a count of 1, the square
root of the value in the accumulator is taken and transferred
(after some formatting) to the signal monitor holding register,
which can be read through the SPI port or output through the
SPORT serial port. The monitor period timer is reloaded with
the value in the SMPR, and the countdown is restarted.
Because the dc offset of the ADC can be significantly larger
than the signal of interest (affecting the results from the signal
monitor), a dc correction circuit is included as part of the signal
monitor block to null the dc offset before measuring the power.
Rev. B | Page 44 of 84
Data Sheet
AD6655
In addition, the first input sample signal power is updated in
the accumulator, and the accumulation continues with the
subsequent input samples. Figure 77 illustrates the rms
magnitude monitoring logic.
When the monitor period timer reaches a count of 1, the value
in the internal count register is transferred to the signal monitor
holding register, which can be read through the SPI port or
output through the SPORT serial port.
FROM
MEMORY
MAP
The monitor period timer is reloaded with the value in the
SMPR register, and the countdown is restarted. The internal
count register is also cleared to a value of 0. Figure 78 illustrates
the threshold crossing logic. The value in the SMR register is
the number of samples that have a magnitude greater than the
threshold register.
DOWN
COUNTER
IS COUNT = 1?
LOAD
CLEAR
ACCUMULATOR
LOAD
POWER MONITOR
HOLDING
REGISTER
TO
MEMORY
MAP
06709-075
FROM
INPUT
PORTS
FROM
MEMORY
MAP
Figure 77. ADC Input RMS Magnitude Monitoring Block Diagram
For rms magnitude mode, the value in the signal monitor result
(SMR) register is a 20-bit fixed-point number. The following
equation can be used to determine the rms magnitude in dBFS
from the MAG value in the register. Note that if the signal
monitor period (SMP) is a power of 2, the second term in the
equation becomes 0.
MAG
SMP
RMS Magnitude = 20 log  20  / 10 log  ceil [log (SMP )] 
2
 2 
2

For ms magnitude mode, the value in the SMR is a 20-bit fixedpoint number. The following equation can be used to determine
the ms magnitude in dBFS from the MAG value in the register.
Note that if the SMP is a power of 2, the second term in the
equation becomes 0.
MAG
SMP
MS Magnitude = 10 log  20  / 10 log  ceil [log (SMP )] 
2
 2 
2

THRESHOLD CROSSING MODE
In the threshold crossing mode of operation, the magnitude of
the input port signal is monitored over a programmable time
period (given by SMPR) to count the number of times it crosses
a certain programmable threshold value. This mode is set by
programming Logic 1x (where x is a don’t care bit) in the signal
monitor mode bits of the signal monitor control register or by
setting the threshold crossing output enable bit in the signal
monitor SPORT control register. Before activating this mode,
the user needs to program the 24-bit SMPR and the 13-bit
upper threshold register for each individual input port. The
same upper threshold register is used for both signal monitoring
and gain control (see the ADC Overrange and Gain Control
section).
After entering this mode, the value in the SMPR is loaded into a
monitor period timer, and the countdown is started. The
magnitude of the input signal is compared with the upper
threshold register (programmed previously) on each input clock
cycle. If the input signal has a magnitude greater than the upper
threshold register, the internal count register is incremented by 1.
The initial value of the internal count register is set to 0. This
comparison and incrementing of the internal count register
continues until the monitor period timer reaches a count of 1.
POWER MONITOR
PERIOD REGISTER
DOWN
COUNTER
TO
INTERRUPT
CONTROLLER
IS COUNT = 1?
LOAD
FROM
INPUT
PORTS
FROM
MEMORY
MAP
CLEAR
A COMPARE
A>B
COMPARE
A>B
LOAD
POWER MONITOR
HOLDING
REGISTER
TO
MEMORY
MAP
B
UPPER
THRESHOLD
REGISTER
06709-076
POWER MONITOR
PERIOD REGISTER
TO
INTERRUPT
CONTROLLER
Figure 78. ADC Input Threshold Crossing Block Diagram
ADDITIONAL CONTROL BITS
For additional flexibility in the signal monitoring process, two
control bits are provided in the signal monitor control register.
They are the signal monitor enable bit and the complex power
calculation mode enable bit.
Signal Monitor Enable Bit
The signal monitor enable bit, located in Bit 0 of Register 0x112,
enables operation of the signal monitor block. If the signal
monitor function is not needed in a particular application, this
bit should be cleared to conserve power.
Complex Power Calculation Mode Enable Bit
When this bit is set, the part assumes that Channel A is digitizing
the I data and Channel B is digitizing the Q data for a complex
input signal (or vice versa). In this mode, the power reported is
equal to
I 2 + Q2
This result is presented in the Signal Monitor DC Value Channel A
register if the signal monitor mode bits are set to 00. The Signal
Monitor DC Value Channel B register continues to compute the
Channel B value.
DC CORRECTION
Because the dc offset of the ADC may be significantly larger
than the signal being measured, a dc correction circuit is included
to null the dc offset before measuring the power. The dc correction
circuit can also be switched into the main signal path, but this
may not be appropriate if the ADC is digitizing a time-varying
signal with significant dc content, such as GSM.
Rev. B | Page 45 of 84
AD6655
Data Sheet
DC Correction Bandwidth
SIGNAL MONITOR SPORT OUTPUT
The dc correction circuit is a high-pass filter with a programmable
bandwidth (ranging between 0.15 Hz and 1.2 kHz at 125 MSPS).
The bandwidth is controlled by writing the 4-bit dc correction
control register located at Register 0x10C, Bits[5:2]. The following
equation can be used to compute the bandwidth value for the dc
correction circuit:
The SPORT is a serial interface with three output pins: the SMI
SCLK (SPORT clock), SMI SDFS (SPORT frame sync), and SMI
SDO (SPORT data output). The SPORT is the master and drives
all three SPORT output pins on the chip.
DC _ Corr _ BW = 2 /k /14 ×
SMI SCLK
f CLK
2× π
where:
k is the 4-bit value programmed in Bits[5:2] of Register 0x10C
(values between 0 and 13 are valid for k; programming 14 or 15
provides the same result as programming 13).
fCLK is the AD6655 ADC sample rate in hertz (Hz).
DC Correction Readback
The current dc correction value can be read back in Register 0x10D
and Register 0x10E for Channel A and Register 0x10F and
Register 0x110 for Channel B. The dc correction value is a
14-bit value that can span the entire input range of the ADC.
DC Correction Freeze
Setting Bit 6 of Register 0x10C freezes the DC correction at its
current state and continues to use the last updated value as the
dc correction value. Clearing this bit restarts dc correction and
adds the currently calculated value to the data.
The data and frame sync are driven on the positive edge of the
SMI SCLK. The SMI SCLK has three possible baud rates: 1/2, 1/4,
or 1/8 the ADC clock rate, based on the SPORT controls. The
SMI SCLK can also be gated off when not sending any data, based
on the SPORT SMI SCLK sleep bit. Using this bit to disable the
SMI SCLK when it is not needed can reduce any coupling errors
back into the signal path, if these prove to be a problem in the
system. Doing so, however, has the disadvantage of spreading
the frequency content of the clock. If desired the SMI SCLK
can be left running to ease frequency planning.
SMI SDFS
The SMI SDFS is the serial data frame sync, and it defines the
start of a frame. One SPORT frame includes data from both
datapaths. The data from Datapath A is sent just after the frame
sync, followed by data from Datapath B.
SMI SDO
The SMI SDO is the serial data output of the block. The data is
sent MSB first on the next positive edge after the SMI SDFS.
Each data output block includes one or more of rms magnitude,
peak level, and threshold crossing values from each datapath in
the stated order. If enabled, the data is sent, rms first, followed
by peak and threshold, as shown in Figure 79.
DC Correction Enable Bits
Setting Bit 0 of Register 0x10C enables dc correction for use in
the signal monitor calculations. The calculated dc correction value
can be added to the output data signal path by setting Bit 1 of
Register 0x10C.
GATED, BASED ON CONTROL
SMI SCLK
SMI SDFS
MSB
PK CH A
RMS/MS CH A LSB
20 CYCLES
16 CYCLES
THR CH A
MSB
16 CYCLES
RMS/MS CH B LSB
20 CYCLES
PK CH B
16 CYCLES
THR CH B
RMS/MS CH A
06709-077
SMI SDO
16 CYCLES
Figure 79. Signal Monitor SPORT Output Timing (RMS, Peak, and Threshold Enabled)
GATED, BASED ON CONTROL
SMI SCLK
SMI SDFS
MSB
RMS/MS CH A LSB
20 CYCLES
THR CH A
16 CYCLES
MSB
RMS/MS CH B LSB
20 CYCLES
THR CH B
16 CYCLES
Figure 80. Signal Monitor SPORT Output Timing (RMS and Threshold Enabled)
Rev. B | Page 46 of 84
RMS/MS CH A
06709-078
SMI SDO
Data Sheet
AD6655
CHANNEL/CHIP SYNCHRONIZATION
The AD6655 has a SYNC input that allows the user flexible
synchronization options for synchronizing the internal blocks.
The sync feature is useful for guaranteeing synchronized operation
across multiple ADCs. The input clock divider, NCO, half-band
filters, and signal monitor block can be synchronized using the
SYNC input. Each of these blocks, except for the signal monitor,
can be enabled to synchronize on a single occurrence of the
SYNC signal or on every occurrence.
The SYNC input is internally synchronized to the sample clock.
However, to ensure that there is no timing uncertainty between
multiple parts, the SYNC input signal should be synchronized
to the input clock signal. The SYNC input should be driven
using a single-ended CMOS type signal.
Rev. B | Page 47 of 84
AD6655
Data Sheet
SERIAL PORT INTERFACE (SPI)
The AD6655 serial port interface (SPI) allows the user to configure
the converter for specific functions or operations through a
structured register space provided inside the ADC. The SPI
gives the user added flexibility and customization, depending on
the application. Addresses are accessed via the serial port and
can be written to or read from via the port. Memory is organized
into bytes that can be further divided into fields. These fields are
documented in the Memory Map section. For detailed operational
information, see Application Note AN-877, Interfacing to High
Speed ADCs via SPI.
CONFIGURATION USING THE SPI
Three pins define the SPI of this ADC: the SCLK/DFS pin, the
SDIO/DCS pin, and the CSB pin (see Table 26). The SCLK/DFS
(serial clock) pin is used to synchronize the read and write data
presented from/to the ADC. The SDIO/DCS (serial data input/
output) pin is a dual-purpose pin that allows data to be sent and
read from the internal ADC memory map registers. The CSB
(chip select bar) pin is an active-low control that enables or disables
the read and write cycles.
Table 26. Serial Port Interface Pins
Pin
SCLK
SDIO
CSB
Function
Serial Clock. The serial shift clock input, which is used to
synchronize serial interface reads and writes.
Serial Data Input/Output. A dual-purpose pin that
typically serves as an input or an output, depending on
the instruction being sent and the relative position in the
timing frame.
Chip Select Bar. An active-low control that gates the read
and write cycles.
The falling edge of the CSB, in conjunction with the rising edge
of the SCLK, determines the start of the framing. An example of
the serial timing and its definitions can be found in Figure 81
and Table 9.
Other modes involving the CSB are available. The CSB can be
held low indefinitely, which permanently enables the device;
this is called streaming. The CSB can stall high between bytes
to allow for additional external timing. When CSB is tied high,
SPI functions are placed in a high impedance mode. This mode
turns on any SPI pin secondary functions.
During an instruction phase, a 16-bit instruction is transmitted.
Data follows the instruction phase, and its length is determined
by the W0 bit and the W1 bit.
All data is composed of 8-bit words. The first bit of each individual
byte of serial data indicates whether a read command or a write
command is issued. This allows the serial data input/output
(SDIO) pin to change direction from an input to an output.
In addition to word length, the instruction phase determines
whether the serial frame is a read or write operation, allowing
the serial port to be used both to program the chip and to read
the contents of the on-chip memory. If the instruction is a readback
operation, performing a readback causes the serial data input/
output (SDIO) pin to change direction from an input to an output
at the appropriate point in the serial frame.
Data can be sent in MSB-first mode or in LSB-first mode. MSB
first is the default on power-up and can be changed via the SPI
port configuration register. For more information about this
and other features, see Application Note AN-877, Interfacing to
High Speed ADCs via SPI at www.analog.com.
HARDWARE INTERFACE
The pins described in Table 26 comprise the physical interface
between the user programming device and the serial port of the
AD6655. The SCLK pin and the CSB pin function as inputs
when using the SPI interface. The SDIO pin is bidirectional,
functioning as an input during write phases and as an output
during readback.
The SPI interface is flexible enough to be controlled by either
FPGAs or microcontrollers. One method for SPI configuration
is described in detail in Application Note AN-812, MicrocontrollerBased Serial Port Interface (SPI) Boot Circuit.
The SPI port should not be active during periods when the full
dynamic performance of the converter is required. Because the
SCLK signal, the CSB signal, and the SDIO signal are typically
asynchronous to the ADC clock, noise from these signals can
degrade converter performance. If the on-board SPI bus is used for
other devices, it may be necessary to provide buffers between
this bus and the AD6655 to prevent these signals from transitioning at the converter inputs during critical sampling periods.
Some pins serve a dual function when the SPI interface is not
being used. When the pins are strapped to AVDD or ground
during device power-on, they are associated with a specific
function. The Digital Outputs section describes the strappable
functions supported on the AD6655.
Rev. B | Page 48 of 84
Data Sheet
AD6655
CONFIGURATION WITHOUT THE SPI
SPI ACCESSIBLE FEATURES
In applications that do not interface to the SPI control registers,
the SDIO/DCS pin, the SCLK/DFS pin, the SMI SDO/OEB pin,
and the SMI SCLK/PDWN pin serve as standalone CMOScompatible control pins. When the device is powered up, it is
assumed that the user intends to use the pins as static control
lines for the duty cycle stabilizer, output data format, output
enable, and power-down feature control. In this mode, the CSB
chip select should be connected to AVDD, which disables the
serial port interface.
Table 28 provides a brief description of the general features that
are accessible via the SPI. These features are described in detail
in Application Note AN-877, Interfacing to High Speed ADCs via
SPI (see www.analog.com). The AD6655 part-specific features
are described in the Memory Map Register Description section.
Table 27. Mode Selection
Clock
Offset
External
Voltage
AVDD (default)
Pin
SDIO/DCS
AGND
SCLK/DFS
AVDD
SMI SDO/OEB
AGND (default)
AVDD
SMI SCLK/PDWN
AGND (default)
AVDD
AGND (default)
Configuration
Duty cycle stabilizer
enabled
Duty cycle stabilizer
disabled
Twos complement
enabled
Offset binary enabled
Outputs in high
impedance
Outputs enabled
Chip in power-down or
standby
Normal operation
tHIGH
tDS
tS
Table 28. Features Accessible Using the SPI
Feature Name
Mode
tDH
Test I/O
Output Mode
Output Phase
Output Delay
VREF
Description
Allows the user to set either power-down
mode or standby mode
Allows the user to access the DCS via the SPI
Allows the user to digitally adjust the
converter offset
Allows the user to set test modes to have
known data on output bits
Allows the user to set up outputs
Allows the user to set the output clock polarity
Allows the user to vary the DCO delay
Allows the user to set the reference voltage
tCLK
tH
tLOW
CSB
SCLK DON’T CARE
R/W
W1
W0
A12
A11
A10
A9
A8
A7
D5
D4
D3
D2
D1
D0
DON’T CARE
06709-079
SDIO DON’T CARE
DON’T CARE
Figure 81. Serial Port Interface Timing Diagram
Rev. B | Page 49 of 84
AD6655
Data Sheet
MEMORY MAP
READING THE MEMORY MAP REGISTER TABLE
Logic Levels
Each row in the memory map register table has eight bit locations.
The memory map is roughly divided into four sections: the chip
configuration registers (Address 0x00 to Address 0x02); the
channel index and transfer registers (Address 0x05 and
Address 0xFF); the ADC functions registers, including setup,
control, and test (Address 0x08 to Address 0x18); and the digital
feature control registers (Address 0x100 to Address 0x123).
An explanation of logic level terminology follows:
The memory map register table (see Table 29) documents the
default hexadecimal value for each hexadecimal address shown.
The column with the heading Bit 7 (MSB) is the start of the
default hexadecimal value given. For example, Address 0x18, the
VREF select register, has a hexadecimal default value of 0xC0. This
means that Bit 7 = 1, Bit 6 = 1, and the remaining bits are 0s. This
setting is the default reference selection setting. The default value
uses a 2.0 V p-p reference. For more information on this function
and others, see Application Note AN-877, Interfacing to High Speed
ADCs via SPI. This document details the functions controlled by
Register 0x00 to Register 0xFF. The remaining registers, from
Register 0x100 to Register 0x123, are documented in the Memory
Map Register Description section.
Address 0x08 to Address 0x18 and Address 0x11E to
Address 0x123 are shadowed. Writes to these addresses do
not affect part operation until a transfer command is issued by
writing 0x01 to Address 0xFF, setting the transfer bit. This allows
these registers to be updated internally and simultaneously when
the transfer bit is set. The internal update takes place when the
transfer bit is set, and the bit autoclears.
Open Locations
All address and bit locations that are not included in Table 29
are not currently supported for this device. Unused bits of a
valid address location should be written with 0s. Writing to these
locations is required only when part of an address location is
open (for example, Address 0x18). If the entire address location
is open (for example, Address 0x13), this address location should
not be written.
Default Values
•
•
“Bit is set” is synonymous with “bit is set to Logic 1” or
“writing Logic 1 for the bit.”
“Clear a bit” is synonymous with “bit is set to Logic 0” or
“writing Logic 0 for the bit.”
Transfer Register Map
Channel-Specific Registers
Some channel setup functions, such as the signal monitor
thresholds, can be programmed differently for each channel.
In these cases, channel address locations are internally duplicated
for each channel. These registers and bits are designated in Table 29
as local. These local registers and bits can be accessed by setting
the appropriate Channel A or Channel B bits in Register 0x05. If
both bits are set, the subsequent write affects the registers of both
channels. In a read cycle, only Channel A or Channel B should
be set to read one of the two registers. If both bits are set during
an SPI read cycle, the part returns the value for Channel A.
Registers and bits designated as global in Table 29 affect the entire
part or the channel features where independent settings are not
allowed between channels. The settings in Register 0x05 do not
affect the global registers and bits.
After the AD6655 is reset, critical registers are loaded with
default values. The default values for the registers are given in
the memory map register table, Table 29.
Rev. B | Page 50 of 84
Data Sheet
AD6655
MEMORY MAP REGISTER TABLE
All address and bit locations that are not included in Table 29 are not currently supported for this device.
Table 29. Memory Map Registers
Addr.
Register
Bit 7
(Hex)
Name
(MSB)
Chip Configuration Registers
0x00
0
SPI Port
Configuration
(Global)
0x01
Chip ID
(Global)
0x02
Chip Grade
(Global)
Bit 6
Bit 5
Bit 4
Bit 3
Bit 2
Bit 1
Bit 0
(LSB)
LSB first
Soft reset
1
1
Soft
reset
LSB first
0
Open
Speed Grade ID[4:3]
00 = 150 MSPS
01 = 125 MSPS
10 = 105 MSPS
11 = 80 MSPS
Default
Notes/
Comments
0x18
The nibbles
are mirrored
so that
LSB-first or
MSB-first
mode
registers
correctly,
regardless of
shift mode
Default is
unique chip
ID, different
for each
device; this is
a read-only
register
Speed grade
ID used to
differentiate
devices; this
is a read-only
register
0x0D
8-bit Chip ID[7:0]
(AD6655 = 0x0D)
(default)
Open
Default
Value
(Hex)
Open
Open
Open
Open
Channel Index and Transfer Registers
0x05
Open
Channel
Index
Open
Open
Open
Open
Open
Data
Channel B
(default)
Data
Channel A
(default)
0x03
0xFF
Open
Open
Open
Open
Open
Open
Open
Transfer
0x00
ADC Function Registers
0x08
Power Modes
Open
Open
Open
Open
Open
Internal power-down
mode (local)
00 = normal operation
01 = full power-down
10 = standby
11 = normal operation
0x00
0x09
Global Clock
(Global)
Open
Open
External
powerdown pin
function
(global)
0 = pdwn
1 = stndby
Open
Open
Open
0x0B
Clock Divide
(Global)
Open
Open
Open
Open
Open
Transfer
Rev. B | Page 51 of 84
Open
Open
Duty cycle
stabilize
(default)
Clock divide ratio
000 = divide by 1
001 = divide by 2
010 = divide by 3
011 = divide by 4
100 = divide by 5
101 = divide by 6
110 = divide by 7
111 = divide by 8
Bits are set to
determine
which device
on chip
receives the
next write
command;
applies to
local registers
Synchronously
transfers data
from the
master shift
register to
the slave
Determines
various
generic
modes of chip
operation
0x01
0x00
Clock divide
values other
than 000
automatically
activate
duty cycle
stabilization
AD6655
Data Sheet
Addr.
(Hex)
0x0D
Register
Name
Test Mode
(Local)
Bit 7
(MSB)
Open
Bit 6
Open
0x10
Offset Adjust
(Local)
Output Mode
Open
Open
Drive
strength
0 V to 3.3
V CMOS or
ANSI
LVDS;
1 V to 1.8
V CMOS or
reduced
LVDS
(global)
Invert
DCO clock
Output
type
0 = CMOS
1 = LVDS
(global)
Interleaved
CMOS
(global)
Output
enable
bar (local)
Open
Open
Open
Open
Open
Open
Open
Open
0x14
0x16
Clock Phase
Control
(Global)
0x17
DCO Output
Delay
(Global)
0x18
VREF Select
(Global)
Bit 5
Open
Bit 4
Open
Bit 3
Open
Bit 2
Bit 0
Bit 1
(LSB)
Output test mode
000 = off (default)
001 = midscale short
010 = positive FS
011 = negative FS
Others = not supported
Offset adjust in LSBs from +31 to -32 (twos complement format)
Output
invert
(local)
Configures
the outputs
and the
format of
the data
0x00
Allows
selection of
clock delays
into the input
divider
Open
Open
Half-band
sync
enable
NCO32
next sync
only
NCO32
sync
enable
0x101
0x102
0x103
0x104
0x105
0x00
0xC0
Clock
divider
sync
enable
Master sync
enable
0x00
fS/8 next
sync only
fS/8 sync
enable
0x00
FIR filter
enable
0x00
Open
0x01
Fast detect
enable
0x00
fS/8 Output
Mix Control
(Global)
FIR Filter and
Output Mode
Control
(Global)
Open
Open
fS/8 start state
Open
Open
Open
Open
Digital Filter
Control
(Global)
Fast Detect
Control
(Local)
Coarse Upper
Threshold
(Local)
Open
Open
Open
Open
Open
Open
Open
Open
FIR gain
fS/8
Complex
output
output
0 = gain of
mix
enable
2
disable
1 = gain of
1
Half-band
Spectral High-pass/
decimation reversal
low-pass
phase
select
Fast Detect Mode Select[2:0]
Open
Open
Open
Open
Open
Rev. B | Page 52 of 84
0x00
0x00
Reference voltage
selection
00 = 1.25 V p-p
01 = 1.5 V p-p
10 = 1.75 V p-p
11 = 2.0 V p-p
(default)
Digital Feature Control Registers
0x100
Sync Control
Signal
Half-band
monitor
next sync
(Global)
sync
only
enable
Open
Default
Notes/
Comments
When
enabled, the
test data is
placed on the
output pins
in place of
ADC output
data
00 = offset binary
01 = twos complement
01 = gray code
11 = offset binary
(local)
Input clock divider phase adjust
000 = no delay
001 = 1 input clock cycle
010 = 2 input clock cycles
011 = 3 input clock cycles
100 = 4 input clock cycles
101 = 5 input clock cycles
110 = 6 input clock cycles
111 = 7 input clock cycles
DCO clock delay
(delay = 2500 ps × register value/31)
00000 = 0 ps
00001 = 81 ps
00010 = 161 ps
…
11110 = 2419 ps
11111 = 2500 ps
Open
Open
Open
Open
Clock
divider
next
sync
only
Open
Default
Value
(Hex)
0x00
Coarse Upper Threshold[2:0]
0x00
Data Sheet
Addr.
(Hex)
0x106
0x107
0x108
0x109
0x10A
0x10B
0x10C
0x10D
0x10E
0x10F
0x110
0x111
0x112
Register
Name
Fine Upper
Threshold
Register 0
(Local)
Fine Upper
Threshold
Register 1
(Local)
Fine Lower
Threshold
Register 0
(Local)
Fine Lower
Threshold
Register 1
(Local)
Increase Gain
Dwell Time
Register 0
(Local)
Increase Gain
Dwell Time
Register 1
(Local)
Signal
Monitor
DC
Correction
Control
(Global)
Signal
Monitor
DC Value
Channel A
Register 0
(Global)
Signal
Monitor
DC Value
Channel A
Register 1
(Global)
Signal
Monitor
DC Value
Channel B
Register 0
(Global)
Signal
Monitor
DC Value
Channel B
Register 1
(Global)
Signal
Monitor
SPORT
Control
(Global)
Signal
Monitor
Control
(Global)
AD6655
Bit 7
(MSB)
Bit 6
Bit 5
Open
Open
Open
Bit 4
Bit 3
Fine Upper Threshold[7:0]
Bit 2
Bit 1
Bit 0
(LSB)
Fine Upper Threshold[12:8]
Open
Open
Open
0x00
Fine Lower Threshold[12:8]
0x00
Increase Gain Dwell Time[7:0]
0x00
In ADC clock
cycles
Increase Gain Dwell Time[15:8]
0x00
In ADC clock
cycles
DC Correction Bandwidth(k:[3:0])
DC
correction
freeze
DC
correction
for signal
path
enable
DC
correction
for signal
monitor
enable
0x00
DC Value Channel A[7:0]
Open
Default
Notes/
Comments
0x00
Fine Lower Threshold[7:0]
Open
Default
Value
(Hex)
0x00
Open
Read only
DC Value Channel A[13:8]
Read only
DC Value Channel B[7:0]
Open
Open
Open
RMS
magnitude
output
enable
Peak
detector
output
enable
Threshold
crossing
output
enable
Complex
power
calculation
mode
enable
Open
Open
Open
Read only
DC Value Channel B[13:8]
SPORT SMI SCLK
SPORT
divide
SMI SCLK
sleep
00 = Undefined
01 = divide by 2
10 = divide by 4
11 = divide by 8
Signal monitor mode
Signal
monitor
00 = rms/ms
rms/ms
magnitude
select
01 = peak detector
0 = rms
10 = threshold crossing
1 = ms
11 = threshold crossing
Rev. B | Page 53 of 84
Read only
Signal
monitor
SPORT
output
enable
0x04
Signal
monitor
enable
0x00
AD6655
Addr.
(Hex)
0x113
0x114
0x115
0x116
0x117
0x118
0x119
0x11A
0x11B
0x11D
0x11E
0x11F
0x120
0x121
0x122
0x123
Register
Name
Signal
Monitor
Period
Register 0
(Global)
Signal
Monitor
Period
Register 1
(Global)
Signal
Monitor
Period
Register 2
(Global)
Signal
Monitor
Result
Channel A
Register 0
(Global)
Signal
Monitor
Result
Channel A
Register 1
(Global)
Signal
Monitor
Result
Channel A
Register 2
(Global)
Signal
Monitor
Result
Channel B
Register 0
(Global)
Signal
Monitor
Result
Channel B
Register 1
(Global)
Signal
Monitor
Result
Channel B
Register 2
(Global)
NCO Control
(Global)
NCO
Frequency 0
NCO
Frequency 1
NCO
Frequency 2
NCO
Frequency 3
NCO Phase
Offset 0
NCO Phase
Offset 1
Data Sheet
Bit 7
(MSB)
Open
Bit 6
Open
Bit 5
Open
Default
Value
(Hex)
0x80
Default
Notes/
Comments
In ADC clock
cycles
Signal Monitor Period[15:8]
0x00
I In ADC clock
cycles
Signal Monitor Period[23:16]
0x00
In ADC clock
cycles
Bit 4
Bit 3
Signal Monitor Period[7:0]
Bit 2
Bit 1
Bit 0
(LSB)
Signal Monitor Result Channel A[7:0]
Read only
Signal Monitor Result Channel A[15:8]
Read only
Open
Signal Monitor Result Channel A[19:16]
Read only
Signal Monitor Result Channel B[7:0]
Read only
Signal Monitor Result Channel B[15:8]
Read only
Open
Open
Open
Open
Open
Open
Open
Open
Signal Monitor Result Channel B[19:16]
Open
NCO32
phase
dither
enable
NCO32
amplitude
dither
enable
NCO32
enable
Read only
0x00
NCO Frequency Value[7:0]
0x00
NCO Frequency Value[15:8]
0x00
NCO Frequency Value[23:16]
0x00
NCO Frequency Value[31:24]
0x00
NCO Phase Value[7:0]
0x00
NCO Phase Value[15:8]
0x00
Rev. B | Page 54 of 84
Data Sheet
AD6655
MEMORY MAP REGISTER DESCRIPTION
For more information on functions controlled in Register 0x00
to Register 0xFF, see Application Note AN-877, Interfacing to
High Speed ADCs via SPI, at www.analog.com. Note that not all
test modes are supported on the AD6655.
Bit 1—Clock Divider Sync Enable
Bit 1 gates the sync pulse to the clock divider. The sync signal
is passed when Bit 1 and Bit 0 are high. This is continuous
sync mode.
SYNC Control (Register 0x100)
Bit 7—Signal Monitor Sync Enable
Bit 0—Master Sync Enable
Bit 7 enables the sync pulse from the external sync input to the
signal monitor block. The sync signal is passed when Bit 7 and
Bit 0 are high. This is continuous sync mode.
fS/8 Output Mix Control (Register 0x101)
Bits[7:6]—Reserved
Bit 6—Half-Band Next Sync Only
Bit 5 and Bit 4 set the starting phase of the fS/8 output mix.
Bit 0 must be high to enable any of the sync functions.
If the master sync enable bit (Register 0x100, Bit 0) and the halfband sync enable bit (Register 0x100, Bit 5) are high, Bit 6 allows
the NCO32 to synchronize following the first sync pulse it
receives and ignore the rest. If Bit 6 is set, Bit 5 of Register
0x100 resets after this sync occurs.
Bit 5—Half-Band Sync Enable
Bit 5 gates the sync pulse to the half-band filter. When Bit 5
is set high, the sync signal causes the half-band to resynchronize, starting at the half-band decimation phase selected in
Register 0x103, Bit 3. This sync is active only when the master
sync enable bit (Register 0x100, Bit 0) is high. This is continuous
sync mode.
Bit 4—NCO32 Next Sync Only
If the master sync enable bit (Register 0x100, Bit 0) and the
NCO32 sync enable bit (Register 0x100, Bit 3) are high, Bit 4
allows the NCO32 to synchronize following the first sync pulse it
receives and ignore the rest. Bit 3 of Register 0x100 resets after
a sync occurs if Bit 4 is set.
Bit 3—NCO32 Sync Enable
Bit 3 gates the sync pulse to the 32-bit NCO. When this bit is set
high, the sync signal causes the NCO to resynchronize, starting
at the NCO phase offset value. This sync is active only when the
master sync enable bit (Register 0x100, Bit 0) is high. This is
continuous sync mode.
Bit 2—Clock Divider Next Sync Only
If the master sync enable bit (Register 0x100, Bit 0) and the
clock divider sync enable bit (Register 0x100, Bit 1) are high,
Bit 2 allows the clock divider to synchronize following the first
sync pulse it receives and ignores the rest. Bit 1 of Register 0x100
resets after it synchronizes.
Bits[5:4]—fS/8 Start State
Bits[3:2]—Reserved
Bit 1—fS/8 Next Sync Only
If the master sync enable bit (Register 0x100, Bit 0) and the fS/8
sync enable bit (Register 0x101, Bit 0) are high, Bit 1 allows the
fS/8 output mix to synchronize following the first sync pulse it
receives and ignore the rest. Bit 0 of Register 0x100 resets after it
synchronizes.
Bit 0—fS/8 Sync Enable
Bit 0 gates the sync pulse to the fS/8 output mix. This sync is
active only when the master sync enable bit (Register 0x100,
Bit 0) is high. This is continuous sync mode.
FIR Filter and Output Mode Control (Register 0x102)
Bits[7:4]—Reserved
Bit 3—FIR Gain
When Bit 3 is set high, the FIR filter path, if enabled, has a gain
of 1. When Bit 3 set low, the FIR filter path has a gain of 2.
Bit 2—fS/8 Output Mix Disable
Bit 2 disables the fS/8 output mix when enabled. Bit 2 should be
set along with Bit 1 to enable complex output mode.
Bit 1—Complex Output Mode Enable
Setting Bit 1 high enables complex output mode.
Bit 0—FIR Filter Enable
When set high, Bit 0 enables the FIR filter. When Bit 0 is
cleared, the FIR filter is bypassed and shut down for power
savings.
Rev. B | Page 55 of 84
AD6655
Data Sheet
Digital Filter Control (Register 0x103)
Bits[7:4]—Reserved
Bit 3—Half-Band Decimation Phase
When set high, Bit 3 uses the alternate phase of the decimating
half-band filter.
Bit 2—Spectral Reversal
Bit 2 enables the spectral reversal feature of the half-band filter.
Bit 1—High-Pass/Low-Pass Select
Bit 1 enables the high-pass mode of the half-band filter when
set high. Setting this bit low enables the low-pass mode (default).
Increase Gain Dwell Time (Register 0x10A and
Register 0x10B)
Register 0x10B, Bits[7:0]—Increase Gain Dwell Time
Bits[15:8]
Register 0x10A, Bits[7:0]—Increase Gain Dwell Time
Bits[7:0]
These register values set the minimum time in ADC sample
clock cycles (after clock divider) that a signal needs to stay below
the fine lower threshold limit before the F_LT and IG are
asserted high.
Bit 0—Reserved
Signal Monitor DC Correction Control (Register 0x10C)
Bit 7—Reserved
Bit 0 reads back as a 1.
Bit 6—DC Correction Freeze
Fast Detect Control (Register 0x104)
Bits[7:4]—Reserved
When Bit 6 is set high, the dc correction is no longer updated to
the signal monitor block, which holds the last dc value
calculated.
Bits[3:1]—Fast Detect Mode Select
Bits[3:1] set the mode of the fast detect output bits according to
Table 29.
Bit 0—Fast Detect Enable
Bit 0 is used to enable the fast detect output pins. When the FD
outputs are disabled, the outputs go into a high impedance state.
In LVDS mode when the outputs are interleaved, the outputs go
high-Z only if both channels are turned off (power-down/
standby/output disabled). If only one channel is turned off
(power-down/standby/output disabled), the fast detect outputs
repeat the data of the active channel.
Coarse Upper Threshold (Register 0x105)
Bits[7:3]—Reserved
Bits[5:2]—DC Correction Bandwidth
Bits[5:2] set the averaging time of the signal monitor dc
correction function. This 4-bit word sets the bandwidth of the
correction block, according to the following equation:
DC _ Corr _ BW = 2/ k / 14 ×
fCLK
2× π
where:
k is the 4-bit value programmed in Bits[5:2] of Register 0x10C
(values between 0 and 13 are valid for k; programming 14 or 15
provides the same result as programming 13).
fCLK is the AD6655 ADC sample rate in hertz (Hz).
Bit 1—DC Correction for Signal Path Enable
Setting this bit high causes the output of the dc measurement
block to be summed with the data in the signal path to remove
the dc offset from the signal path.
Bits[2:0]—Coarse Upper Threshold
These bits set the level required to assert the coarse upper
threshold indication (see Table 25).
Fine Upper Threshold (Register 0x106 and Register 0x107)
Register 0x107, Bits[7:5]—Reserved
Register 0x107, Bits[4:0]—Fine Upper Threshold Bits[12:8]
Register 0x106, Bits[7:0]—Fine Upper Threshold Bits[7:0]
These registers provide a fine upper limit threshold. The 13-bit
value is compared to the 13-bit magnitude from the ADC block.
If the ADC magnitude exceeds this threshold value, the F_UT
indicator is set.
Bit 0—DC Correction for Signal Monitor Enable
This bit enables the dc correction function in the signal monitor
block. The dc correction is an averaging function that can be
used by the signal monitor to remove dc offset in the signal.
Removing this dc from the measurement allows a more
accurate power reading.
Signal Monitor DC Value Channel A (Register 0x10D and
Register 0x10E)
Register 0x10E, Bits[7:6]—Reserved
Fine Lower Threshold (Register 0x108 and Register 0x109)
Register 0x109, Bits[7:5]—Reserved
Register 0x10E, Bits[5:0]—DC Value Channel A[13:8]
Register 0x109, Bits[4:0]—Fine Lower Threshold Bits[12:8]
These read-only registers hold the latest dc offset value computed
by the signal monitor for Channel A.
Register 0x108, Bits[7:0]—Fine Lower Threshold Bits[7:0]
Register 0x10D, Bits[7:0]—DC Value Channel A[7:0]
These registers provide a fine lower limit threshold. This 13-bit
value is compared with the 13-bit magnitude from the ADC
block. If the ADC magnitude is less than this threshold value,
the F_LT indicator is set.
Rev. B | Page 56 of 84
Data Sheet
AD6655
Signal Monitor DC Value Channel B (Register 0x10F and
Register 0x110)
Register 0x110, Bits[7:6]—Reserved
Register 0x110, Bits[5:0]—Channel B DC Value Bits[13:8]
Register 0x10F, Bits[7:0]—Channel B DC Value Bits [7:0]
These read-only registers hold the latest dc offset value computed
by the signal monitor for Channel B.
Signal Monitor SPORT Control (Register 0x111)
Bit 7—Reserved
Bit 6—RMS/MS Magnitude Output Enable
Bit 6 enables the 20-bit rms or ms magnitude measurement as
output on the SPORT.
Bit 5—Peak Detector Output Enable
Bit 5 enables the 13-bit peak measurement as output on the SPORT.
Bit 4—Threshold Crossing Output Enable
Bit 4 enables the 13-bit threshold measurement as output on the
SPORT.
Bit 0—Signal Monitor Enable
Setting Bit 0 high enables the signal monitor block.
Signal Monitor Period (Register 0x113 to Register 0x115)
Register 0x115 Bits[7:0]—Signal Monitor Period[23:16]
Register 0x114 Bits[7:0]—Signal Monitor Period[15:8]
Register 0x113 Bits[7:0]—Signal Monitor Period[7:0]
This 24-bit value sets the number of clock cycles over which the
signal monitor performs its operation. The minimum value for
this register is 128 cycles (programmed values less than 128
revert to 128).
Signal Monitor Result Channel A (Register 0x116 to
Register 0x118)
Register 0x118, Bits[7:4]—Reserved
Register 0x118, Bits[3:0]—Signal Monitor Result
Channel A[19:16]
Register 0x117, Bits[7:0]—Signal Monitor Result
Channel A[15:8]
Bits[3:2]—SPORT SMI SCLK Divide
Register 0x116, Bits[7:0]—Signal Monitor Result
Channel A[7:0]
The values of these bits set the SPORT SMI SCLK divide ratio from
the input clock. A value of 0x01 sets divide by 2 (default), a value
of 0x10 sets divide by 4, and a value of 0x11 sets divide by 8.
This 20-bit value contains the power value calculated by the
signal monitor block for Channel A. The content is dependent
on the settings in Register 0x112, Bits[2:1].
Bit 1—SPORT SMI SCLK Sleep
Setting Bit 1 high causes the SMI SCLK to remain low when the
signal monitor block has no data to transfer.
Bit 0—Signal Monitor SPORT Output Enable
When set, Bit 0 enables the signal monitor SPORT output to
begin shifting out the result data from the signal monitor block.
Signal Monitor Control (Register 0x112)
Bit 7—Complex Power Calculation Mode Enable
This mode assumes I data is present on one channel and Q data
is present on the alternate channel. The result reported is the
complex power measured as
I 2 + Q2
Signal Monitor Result Channel B (Register 0x119 to
Register 0x11B)
Register 0x11B, Bits[7:4]—Reserved
Register 0x11B, Bits[3:0]—Signal Monitor Result
Channel B[19:16]
Register 0x11A, Bits[7:0]—Signal Monitor Result
Channel B[15:8]
Register 0x119, Bits[7:0]—Signal Monitor Result
Channel B[7:0]
This 20-bit value contains the power value calculated by the
signal monitor block for Channel B. The content is dependent
on the settings in Register 0x112, Bits[2:1].
Bits[6:4]—Reserved
NCO Control (Register 0x11D)
Bits[7:3]—Reserved
Bit 3—Signal Monitor RMS/MS Select
Bit 2—NCO32 Phase Dither Enable
Setting Bit 3 low selects rms power measurement mode. Setting
Bit 3 high selects ms power measurement mode.
When Bit 2 is set, phase dither in the NCO is enabled. When
Bit 2 is cleared, phase dither is disabled.
Bits[2:1]—Signal Monitor Mode
Bit 1—NCO32 Amplitude Dither Enable
Bit 2 and Bit 1 set the mode of the signal monitor for data
output to registers at Address 0x116 through Address 0x11B.
Setting these bits to 0x00 selects rms/ms magnitude output,
setting these bits to 0x01 selects peak detector output, and
setting 0x10 or 0x11 selects threshold crossing output.
When Bit 1 is set, amplitude dither in the NCO is enabled.
When Bit 1 is cleared, amplitude dither is disabled.
Rev. B | Page 57 of 84
AD6655
Data Sheet
Bit 0—NCO32 Enable
When Bit 0 is set, this bit enables the 32-bit NCO operating at
the frequency programmed into the NCO frequency register.
When Bit 0 is cleared, the NCO is bypassed and shuts down for
power savings.
NCO Frequency (Register 0x11E to Register 0x121)
Register 0x11E, Bits[7:0]—NCO Frequency Value[7:0]
Register 0x11F, Bits[7:0]—NCO Frequency Value[15:8]
Register 0x120, Bits[7:0]—NCO Frequency Value[23:16]
NCO Phase Offset (Register 0x122 and Register 0x123)
Register 0x122, Bits[7:0]—NCO Phase Value[7:0]
Register 0x123, Bits[7:0]—NCO Phase Value[15:8]
The 16-bit value programmed into the NCO phase value register
is loaded into the NCO block each time the NCO is started or
when an NCO SYNC signal is received. This process allows the
NCO to be started with a known nonzero phase.
Use the following equation to calculate the NCO phase offset
value:
NCO_PHASE = 216 × PHASE/360
Register 0x121, Bits[7:0]—NCO Frequency Value[31:24]
This 32-bit value is used to program the NCO tuning frequency.
The frequency value to be programmed is given by the
following equation:
NCO_FREQ = 232 ×
Mod( f , fCLK )
fCLK
where:
NCO_PHASE is a decimal number equal to the 16-bit binary
number to be programmed at Register 0x122 and Register 0x123.
PHASE is the desired NCO phase in degrees.
where:
NCO_FREQ is a 32-bit twos complement number representing
the NCO frequency register.
f is the desired carrier frequency in hertz (Hz).
fCLK is the AD6655 ADC clock rate in hertz (Hz).
Rev. B | Page 58 of 84
Data Sheet
AD6655
APPLICATIONS INFORMATION
DESIGN GUIDELINES
Before starting system-level design and layout of the AD6655,
it is recommended that the designer become familiar with these
guidelines, which discuss the special circuit connections and
layout requirements needed for certain pins.
For the specifications provided in Table 2, the fS/2 spur, if in
band, is excluded from the SNR values. It is treated as a
harmonic, in terms of SNR. The fS/2 level is included in the
SFDR and worst other specifications.
–60
Power and Ground Recommendations
Depending on the relationship of the IF frequency to the center
of the Nyquist zone, this spurious tone may or may not exist in the
AD6655 output band. Some residual fS/2 energy is present in
the AD6655, and the level of this spur is typically below the
level of the harmonics at clock rates of 125 MSPS and below.
Figure 82 shows a plot of the fS/2 spur level vs. analog input
frequency for the AD6655-125. At sampling rates above
125 MSPS, the fS/2 spur level increases and is at a higher level
than the worst harmonic, as shown in Figure 83, which shows
the AD6655-150 fS/2 levels.
–SFDR
–90
–100
fS/2 SPUR
06709-083
–110
–120
0
50
100
150
200
250
300
350
400
450
500
INPUT FREQUENCY (MHz)
Figure 82. AD6655-125 SFDR and fS/2 Spurious Level vs. Input Frequency (fIN)
with DRVDD = 1.8 V Parallel CMOS Output Mode
–60
–70
SFDR AND fS/2 SPUR (dBFS)
fS/2 Spurious
Because the AD6655 output data rate is at one-half the sampling
frequency, there is significant fS/2 energy in the outputs of the
part. If this fS/2 spur falls in band, care must be taken to ensure
that this fS/2 energy does not couple into either the clock circuit
or the analog inputs of the AD6655. When fS/2 energy is coupled
in this fashion, it appears as a spurious tone reflected around fS/4,
3fS/4, 5fS/4, and so on. For example, in a 125 MSPS sampling
application with a 90 MHz single-tone analog input, this energy
generates a tone at 97.5 MHz. In this example, the center of the
Nyquist zone is 93.75 MHz; therefore, the 90 MHz input signal is
3.75 MHz from the center of the Nyquist zone. As a result, the fS/2
spurious tone appears at 97.5 MHz, or 3.75 MHz above the center
of the Nyquist zone. These frequencies are then tuned by the NCOs
before being output by the AD6655.
–80
fS/2 SPUR
–80
–90
–SFDR
–100
–110
06709-084
A single PCB ground plane should be sufficient when using the
AD6655. With proper decoupling and smart partitioning of the
PCB analog, digital, and clock sections, optimum performance
is easily achieved.
SFDR AND fS/2 SPUR (dBFS)
–70
When connecting power to the AD6655, it is recommended
that two separate 1.8 V supplies be used: one supply should be
used for analog (AVDD) and digital (DVDD), and a separate
supply should be used for the digital outputs (DRVDD). The
AVDD and DVDD supplies, while derived from the same source,
should be isolated with a ferrite bead or filter choke and separate
decoupling capacitors. The designer can employ several different
decoupling capacitors to cover both high and low frequencies.
These capacitors should be located close to the point of entry
at the PC board level and close to the pins of the part with
minimal trace length.
–120
0
50
100
150
200
250
300
350
400
450
500
ANALOG INPUT FREQUENCY (MHz)
Figure 83. AD6655-150 SFDR and fS/2 Spurious Level vs. Input Frequency (fIN)
with DRVDD = 1.8 V Parallel CMOS Output Mode
Operating the part with a 1.8 V DRVDD voltage rather than a 3.3 V
DRVDD lowers the fS/2 spur. In addition, using LVDS, CMOS
interleaved, or CMOS IQ output modes also reduces the fS/2
spurious level.
LVDS Operation
The AD6655 defaults to CMOS output mode on power-up.
If LVDS operation is desired, this mode must be programmed
using the SPI configuration registers after power-up. When the
AD6655 powers up in CMOS mode with LVDS termination
resistors (100 Ω) on the outputs, the DRVDD current can be
higher than the typical value until the part is placed in LVDS
mode. This additional DRVDD current does not cause damage
to the AD6655, but it should be taken into account when considering the maximum DRVDD current for the part.
Rev. B | Page 59 of 84
AD6655
Data Sheet
To avoid this additional DRVDD current, the AD6655 outputs
can be disabled at power-up by taking the OEB pin high. After
the part is placed into LVDS mode via the SPI port, the OEB
pin can be taken low to enable the outputs.
Exposed Paddle Thermal Heat Slug Recommendations
It is mandatory that the exposed paddle on the underside of the
ADC be connected to analog ground (AGND) to achieve the
best electrical and thermal performance. A continuous, exposed
(no solder mask) copper plane on the PCB should mate to the
AD6655 exposed paddle, Pin 0.
The copper plane should have several vias to achieve the lowest
possible resistive thermal path for heat dissipation to flow through
the bottom of the PCB. These vias should be filled or plugged with
nonconductive epoxy.
To maximize the coverage and adhesion between the ADC
and the PCB, a silkscreen should be overlaid to partition the
continuous plane on the PCB into several uniform sections.
This provides several tie points between the ADC and the PCB
during the reflow process. Using one continuous plane with no
partitions guarantees only one tie point between the ADC and
the PCB. See the evaluation board for a PCB layout example.
For detailed information about packaging and PCB layout of
chip scale packages, refer to Application Note AN-772, A Design
and Manufacturing Guide for the Lead Frame Chip Scale Package
(LFCSP) (see www.analog.com).
CML
The CML pin should be decoupled to ground with a 0.1 μF
capacitor, as shown in Figure 48.
RBIAS
The AD6655 requires that a 10 kΩ resistor be placed between
the RBIAS pin and ground. This resistor sets the master current
reference of the ADC core and should have at least a 1% tolerance.
Reference Decoupling
The VREF pin should be externally decoupled to ground with
a low ESR, 1.0 μF capacitor in parallel with a low ESR, 0.1 μF
ceramic capacitor.
SPI Port
The SPI port should not be active during periods when the full
dynamic performance of the converter is required. Because the
SCLK, CSB, and SDIO signals are typically asynchronous to the
ADC clock, noise from these signals can degrade converter
performance. If the on-board SPI bus is used for other devices,
it may be necessary to provide buffers between this bus and the
AD6655 to keep these signals from transitioning at the converter
inputs during critical sampling periods.
Rev. B | Page 60 of 84
Data Sheet
AD6655
EVALUATION BOARD
The AD6655 evaluation board provides all of the support circuitry
required to operate the ADC in its various modes and configurations. The converter can be driven differentially through a double
balun configuration (default) or optionally through the AD8352
differential driver. The ADC can also be driven in a single-ended
fashion. Separate power pins are provided to isolate the DUT
from the AD8352 drive circuitry. Each input configuration can
be selected by proper connection of various components (see
Figure 85 to Figure 94). Figure 84 shows the typical bench
characterization setup used to evaluate the ac performance of
the AD6655.
External supplies can be used to operate the evaluation board
by removing L1, L3, L4, and L13 to disconnect the voltage
regulators supplied from the switching power supply. This enables
the user to individually bias each section of the board. Use P3
and P4 to connect a different supply for each section. At least
one 1.8 V supply is needed with a 1 A current capability for
AVDD and DVDD; a separate 1.8 V to 3.3 V supply is recommended for DRVDD. To operate the evaluation board using the
AD8352 option, a separate 5.0 V supply (AMP VDD) with
a 1 A current capability is needed. To operate the evaluation board
using the alternate SPI options, a separate 3.3 V analog supply
(VS) is needed, in addition to the other supplies. The 3.3 V
supply (VS) should have a 1 A current capability, as well. Solder
Jumper SJ35 allows the user to separate AVDD and DVDD,
if desired.
It is critical that the signal sources used for the analog input and
clock have very low phase noise (<<1 ps rms jitter) to realize the
optimum performance of the converter. Proper filtering of the
analog input signal to remove harmonics and lower the integrated
or broadband noise at the input is also necessary to achieve the
specified noise performance.
INPUT SIGNALS
When connecting the clock and analog source, use clean signal
generators with low phase noise, such as the Rohde & Schwarz
SMA100A signal generators or the equivalent. Use 1 m long,
shielded, RG-58, 50 Ω coaxial cable for making connections to the
evaluation board. Enter the desired frequency and amplitude for
the ADC. The AD6655 evaluation board from Analog Devices,
Inc., can accept a ~2.8 V p-p or 13 dBm sine wave input for the
clock. When connecting the analog input source, it is recommended that a multipole, narrow-band, band-pass filter with 50 Ω
terminations be used. Band-pass filters of this type are available
from TTE, Allen Avionics, and K&L Microwave, Inc. Connect
the filter directly to the evaluation board, if possible.
See Figure 85 to Figure 102 for the complete schematics and
layout diagrams that demonstrate the routing and grounding
techniques that should be applied at the system level.
POWER SUPPLIES
This evaluation board comes with a wall-mountable switching
power supply that provides a 6 V, 2 A maximum output. Connect
the supply to the rated 100 V ac to 240 V ac wall outlet at 47 Hz
to 63 Hz. The output of the supply is a 2.1 mm inner diameter
circular jack that connects to the PCB at J16. Once on the PC
board, the 6 V supply is fused and conditioned before connection
to six low dropout linear regulators that supply the proper bias
to each of the various sections on the board.
OUTPUT SIGNALS
The parallel CMOS outputs interface directly with the Analog
Devices standard ADC data capture board (HSC-ADC-EVALCZ).
For more information on the ADC data capture boards and their
optional settings, visit www.analog.com/FIFO.
WALL OUTLET
100V TO 240V AC
47Hz TO 63Hz
–
+
GND
VCP
AD6655
EVALUATION BOARD
14-BIT
PARALLEL
CMOS
14-BIT
PARALLEL
CMOS
CLK
SPI
Figure 84. Evaluation Board Connection
Rev. B | Page 61 of 84
HSC-ADC-EVALCZ
FPGA BASED
DATA
CAPTURE BOARD
PC RUNNING
VISUAL ANALOG
AND SPI
CONTROLLER
SOFTWARE
USB
CONNECTION
SPI
06709-108
ROHDE & SCHWARZ,
SMA100A,
2V p-p SIGNAL
SYNTHESIZER
+
VS
AINB
3.3V
–
GND
BAND-PASS
FILTER
3.3V
+
DRVDD IN
ROHDE & SCHWARZ,
SMA100A,
2V p-p SIGNAL
SYNTHESIZER
3.3V
–
GND
AINA
–
GND
BAND-PASS
FILTER
+
AMP VDD
ROHDE & SCHWARZ,
SMA100A,
2V p-p SIGNAL
SYNTHESIZER
1.8V
+
–
GND
5.0V
SWITCHING
POWER
SUPPLY
AVDD IN
6V DC
2A MAX
AD6655
Data Sheet
DEFAULT OPERATION AND JUMPER SELECTION
SETTINGS
The following is a list of the default and optional settings or
modes allowed on the AD6655 evaluation board.
POWER
Connect the switching power supply that is provided in the
evaluation kit between a rated 100 V ac to 240 V ac wall outlet
at 47 Hz to 63 Hz and P500.
VIN
The evaluation board is set up for a double balun configuration
analog input with optimum 50 Ω impedance matching from
70 MHz to 200 MHz. For more bandwidth response, the differential capacitor across the analog inputs can be changed or
removed (see Table 14). The common mode of the analog inputs
is developed from the center tap of the transformer via the CML
pin of the ADC (see the Analog Input Considerations section).
VREF
VREF is set to 1.0 V by tying the SENSE pin to ground by adding
a jumper on Header J5 (Pin 1 to Pin 2). This causes the ADC to
operate in 2.0 V p-p full-scale range. To place the ADC in 1.0 V p-p
mode (VREF = 0.5 V), a jumper should be placed on Header J4.
A separate external reference option is also included on the evaluation board. To use an external reference, connect J6 (Pin 1 to Pin 2)
and provide an external reference at TP5. Proper use of the VREF
options is detailed in the Voltage Reference section.
RBIAS
RBIAS requires a 10 kΩ resistor (R503) to ground and is used to
set the ADC core bias current.
CLOCK
The default clock input circuitry is derived from a simple baluncoupled circuit using a high bandwidth 1:1 impedance ratio balun
(T5) that adds a very low amount of jitter to the clock path. The
clock input is 50 Ω terminated and ac-coupled to handle singleended sine wave inputs. The transformer converts the single-ended
input to a differential signal that is clipped before entering the
ADC clock inputs. When the AD6655 input clock divider is
utilized, clock frequencies up to 625 MHz can be input into the
evaluation board through Connector S5.
PDWN
To enable the power-down feature, connect J7, shorting the
PDWN pin to AVDD.
CSB
The CSB pin is internally pulled up, setting the chip into
external pin mode, to ignore the SDIO and SCLK information.
To connect the control of the CSB pin to the SPI circuitry on the
evaluation board, connect J21, Pin 1 to J21, Pin 2.
SCLK/DFS
If the SPI port is in external pin mode, the SCLK/DFS pin sets the
data format of the outputs. If the pin is left floating, the pin is internally pulled down, setting the default data format condition to
offset binary. Connecting J2, Pin 1 to J2, Pin 2 sets the format to
twos complement. If the SPI port is in serial pin mode, connecting
J2, Pin 2 to J2, Pin 3 connects the SCLK pin to the on-board SPI
circuitry (see the Serial Port Interface (SPI) section).
SDIO/DCS
If the SPI port is in external pin mode, the SDIO/DCS pin sets
the duty cycle stabilizer. If the pin is left floating, the pin is
internally pulled up, setting the default condition to DCS enabled.
To disable the DCS, connect J1, Pin 1 to J1, Pin 2. If the SPI port
is in serial pin mode, connecting J1, Pin 2 to J1, Pin 3 connects
the SDIO pin to the on-board SPI circuitry (see the Serial Port
Interface (SPI) section).
ALTERNATIVE CLOCK CONFIGURATIONS
Two alternate clocking options are provided on the AD6655
evaluation board. The first option is to use an on-board crystal
oscillator (Y1) to provide the clock input to the part. To enable
this crystal, Resistor R8 (0 Ω) and Resistor R85 (10 kΩ) should
be installed, and Resistor R82 and Resistor R30 should be removed.
A second clock option is to use a differential LVPECL clock to
drive the ADC input using the AD9516 (U2). When using this
drive option, the AD9516 charge pump filter components need
to be populated (see Figure 89). Consult the AD9516 data sheet
for more information.
To configure the clock input from S5 to drive the AD9516
reference input instead of directly driving the ADC, the
following components need to be added, removed, and/or
changed.
1.
Remove R32, R33, R99, and R101 in the default
clock path.
2.
Populate C78 and C79 with 0.001 μF capacitors and
R78 and R79 with 0 Ω resistors in the clock path.
In addition, unused AD9516 outputs (one LVDS and one LVPECL)
are routed to optional Connector S8 through Connector S11 on
the evaluation board.
Rev. B | Page 62 of 84
Data Sheet
AD6655
ALTERNATIVE ANALOG INPUT DRIVE
CONFIGURATION
1.
Remove C1, C17, C18, and C117 in the default analog
input path.
This section provides a brief description of the alternative
analog input drive configuration using the AD8352. When
using this particular drive option, some additional components
need to be populated. For more details on the AD8352 differential
driver, including how it works and its optional pin settings,
consult the AD8352 data sheet.
2.
Populate C8 and C9 with 0.1 µF capacitors in the analog
input path. To drive the AD8352 in the differential input
mode, populate the T10 transformer; the R1, R37, R39,
R126, and R127 resistors; and the C10, C11, and C125
capacitors.
3.
Populate the optional amplifier output path with the
desired components including an optional low-pass filter.
Install 0 Ω resistors, R44 and R48. R43 and R47 should be
increased (typically to 100 Ω) to increase to 200 Ω the
output impedance seen by the AD8352.
To configure the analog input to drive the AD8352 instead of
the default transformer option, the following components need
to be added, removed, and/or changed for Channel A. For
Channel B, the corresponding components should be changed.
Rev. B | Page 63 of 84
AIN+
1
S2
2
R28
1
R1
57.6 OHM
R121
0 OHM
RES0402
R120
0 OHM
INA+
0.1U
C117
0.1U
C1
0 OHM
R2
INA+
0.1U
C47
INA-
0.1U
C9
T10
R54
0 OHM
P
S
3
2
1
DNP
R36
5
4
P
5
4
S
ETC1-1-13
T1
1ADT1_1WT6
2
3
T7
0 OHM
R110
1
2
3
1
2
3
CML
P
ETC1-1-13
S
T2
5
4
0.1U
C18
0.1U
C17
DEFAULT AMPLIFIER INPUT PATH
4
5
ETC1-1-1 3
F
0.1U
R31
R29
R35
0 OHM
24.9 OHM
24.9 OHM
C8
4.12K
0.1U
C11
0.1U
C10
R126
INA-
0 OHM
R48
0 OHM
R44
C125
.3PF
R37
100 OHM
CML
RGP
RDP
VIN
4 RDN
0 OHM
R40
5
16
VIP
3 RGN
2
1
R42
AMP+A
AMP-A
0 OHM
R39
DNP
R38
0 OHM
R127
W1
Z1
GND
VOP
13
VCC
AMPVDD
VCC
8
C3
0.1U
12
9
10
11
C22
0.1U
GND
VON
AD8352
VCM
14
GND
6
7
15
ENB
A B
R41
R5
OPTIONAL AMPLIFIER INPUT PATH
F
10K OHM
57.6 OHM
10K OHM
R43
R47
S1
2
AIN-
57.6 OHM
Figure 85. Evaluation Board Schematic, Channel A Analog Inputs
R4
33 OHM
33 OHM
Rev. B | Page 64 of 84
0 OHM
F
33 OHM
R27
33 OHM
R26
C23
0.1U
C27
10U
C12
0.001U
C16
0.001U
AMPVDD
C2
0.1U
C5
4.7PF
L15 1
IND0603
L14 1
IND0603
DNP
120NH
DNP
120NH
2
2
0 OHM
R49
DNP
180NH
DNP
R50
0 OHM
2
2
AVDD
AVDD
AMP+A
C139
12PF
DNP
AMP-A
Transformer/amp channel A
VIN+A
TP15
1
L16
180NH
VIN-A
TP14
1
L17 1
IND0603
C4
18PF
DNP
1
IND0603
06709-200
AMPVDD
AD6655
Data Sheet
SCHEMATICS
Figure 86. Evaluation Board Schematic, Channel B Analog Inputs
AIN+
AIN-
S4
S3
1
1
57.6 OH M
R52
57.6 OH M
R51
RES0402
0 OH M
R123
RES0402
0 OH M
R122
INB0.1U
C31
INB+
0.1U
C6
0.1U
C28
0 OH M
R67
INB -
4
5
P
3
2
1
DN P
0.1U
0.1U
C39
.3PF
C128
C51
0.1U
4
5
T8
P
T3
4
5
6
S
ETC1-1-1 3
3
2
1
ADT1_1W T
0 OH M
R111
3
2
1
4
5
CML
P
T4
S
ETC1-1-1 3
3
2
1
DN P
R133
0 OH M
R132
R6
0 OH M
DEFAULT AMP LIFIER INPUT PATH
0 OH M
R55
T11
S
ETC1-1-1 3
0 OH M
F
0.1U
R134
R135
INB+
24.9 OH M
24.9 OH M
C38
R128
R66
R68
F
R129
C30
4.12K
R69
2
2
0.1U
C82
0.1U
C7
100 OH M
RGP
RDP
VIN
4 RDN
5
16
VIP
3 RGN
2
1
15
0 OH M
R95
CML
0 OH M
R94
6
ENB
Z2
7
AMP+B
0 OH M
R96
AMP-B
GND
VON
VCC
8
GND
R53
VOP
VCC
13
AMPVDD
AD8352
VCM
14
GND
10K OH M
A B
W2
C60
0.1U
9
10
11
12
R70
R71
R131
C61
0.1U
C24
0.1U
C62
10U
C83
0.1U
AMPVDD
0.001U
C140
0.001U
C46
57.6 OH M
OPTIONAL AMPLIFIER INPUT PATH
F
0 OH M
R72
10K OH M
33 OH M
33 OH M
Rev. B | Page 65 of 84
R73
33 OH M
R74
33 OH M
L19 1
IND0603
L18 1
IND0603
C84
4.7PF
DNP
180NH
DNP
120NH
DNP
L21 1
IND0603
180NH
C19
18PF
DNP
L20 1
IND0603
DNP
2
2
120NH
2
2
R80
R81
0 OH M
0 OH M
TP17
1
TP16
1
AMP-B
C29
12PF
DNP
VIN+B
VIN- B
AMP+B
AVDD
AVDD
06709-201
AMPVDD
Data Sheet
AD6655
S6
SMA200U P
ENC\
ENC
S5
SMA200U P
1
1
R30
R7
R8
57.6OHM
57.6OHM
0 OHM
R85
R82
R90
10KOHM
10KOHM
0 OHM
R3
0 OHM
Figure 87. Evaluation Board Schematic, DUT Clock Input
0.001U
C77
0.001U
C94
0.001U
C63
0.001U
4
5
0.1U
OPT_CLK-
3
S 2
T5
ETC1-1-13
P
1
6 T9
5
4
ADT1_1W
T
1
2
3
C56
OPT_CLK+
0.1U
R32
0.001U
C79
0 OHM
R33
0 OHM
0.001U
C78
OPT_CLK-
ALTCLK-
OPT_CLK+
ALTCLK+
R78
0 OHM
R79
0 OHM
R101
0 OHM
R99
0 OHM
R83
0.1U
C21
24.9OHM
R84
0.1U
C20
24.9OHM
C145
1
C64
F
2
2
Rev. B | Page 66 of 84
2
VS
CLK-
CLK+
AD6655
Data Sheet
06709-202
TP2
DNP
R34
S7
1
CLK IN
AD9516
VS_OUT_D R
VCXO_CLK-
RES0402
0 OHM
R125
RES0402
R89
0 OHM
C100
0.1U
0 OHM
R124
VCXO_CLK+
C104
0.1U
49.9 OHM
C101
0.1U
C98
0.1U
0.1U
C143
0.1U
C142
C80
18PF
VCP
C99
0.1U
VS
SCLK
VCP
BYPASS_LDO
9
BYPASS_LDO
CLK
C96
0.1U
SCLK
16
NC1
15
CLKB
14
13
C97
0.1U
VS_CLK_DIS
T
12
VS_VCO
11
10
LF
SYNCB
LF
REF_SEL
7
STATUS
6
CP
4
5
8
STATUS
REFMON
LD
2
3
SYNCB
CP
TP18
TEST
1
REF_SEL
VCP
REFMON
U2
AD9516_64LFCS
P
RESETB
TP19
TES1
T
RSET_CLOC
K58
23
R10
PDB
1
LD
OUT5B
VS_OUT45_DI
V
OUT153
28
OUT5
VS_OUT01_DR
V54
27
VS_OUT45_DR
V
SDO
CP_RSE
T62
19
NC3
NC4
CSB
R11
REFINB
18
63
NC2
5.1K
VS_PLL_
261
20
R12
SDIO
4.12K
VS_REF57
24
TEST
OUT4
VS_PLL_1
OUT4B
1
OUT056
25
TP20
2
OUT0B55
26
VS
VS_OUT01_DI
V51
30
OPT_CLK +
OUT1B52
29
VS
VS_OUT89_
1
AGND
VS_OUT89_
2
VS_OUT_DR
VS_OUT67_
250
31
VS
VS_OUT67_
149
32
OPT_CLK -
REFIN64
17
CSB_2
VS_PRESCALE
R60
21
SDO
GND_REF
59
22
SDI
Rev. B | Page 67 of 84
RESETB
Figure 88. Evaluation Board Schematic, Optional AD9516 Clock Circuit
PDB
VS
36
35
34
33
OUT9B
OUT9
OUT8B
OUT8
GND_OUT89_DI
V 37
39
38
40
OUT3B
OUT3
VS_OUT23_DI
V
VS
R88
200
R86
200
AGND
VS
R92
200
R91
200
ALTCLK+
VS_OUT_DR
42
OUT2B
VS_OUT23_DR
V 41
ALTCLK-
43
LVPECL
TO ADC
1TP8
0.001U
C141
OUT2
AGND
R9
44
45
46
47
48
100 OHM
GND_ESD
OUT7B
OUT7
OUT6B
OUT6
OUT6N
SYNC
0.1U
C86
0.1U
C85
0.1U
C87
0.1U
C88
1
1
1
1
S8
S9
S10
S11
2
VS_OUT_D
R
2
PAD
2
LVDS
LVPECL
OUTPUT
OUTPUT
06709-203
OUT6P
Data Sheet
AD6655
2
100 OHM
R75
CP
Figure 89. Evaluation Board Schematic, Optional AD9516 Loop Filter/VCO and SYNC Input
Rev. B | Page 68 of 84
BYPASS_LDO
VAL
R136
SYNC
S12
SMA200UP
2
R98
VAL
C90
SEL
RES0603
57.6 OH M
R45
C89
SEL
R93
VAL
VAL
R137
SEL
C91
C144
SEL
0.1U
Charge Pump Filter
1
C25
VAL
R97
3
2
A2
NL27WZ04
C92
SEL
GND
Y1
4
5
6
RES0402
0 OH M
R117
RES0402
0 OH M
R116
Y2
VCC
LF
RES0402
TP1
1
R87
OSCVECTRON_VS500
RES0402
0 OH M
R104
U25
4
OUT2
3
GND
6
VCC
5
OUT1
24.9 OH M
2
OUT_DISABLE
VS-500
1
FREQ_CTRL_V
33 OH M
R46
SYNC
R106
R108
U3
10K OH M
10K OH M
A1
R107
R109
1
10K OH M
10K OH M
VS
R114
RES0402
0 OH M
RES0402
R139
0 OH M
VCP
VCP
RES0402
10K OH M
R100
VCXO_CLK-
VCXO_CLK+
VS
REF_SE L
VS
PD B
VS
SYNC B
VS
RESET B
06709-204
C26
0.1U
RES0402
RES0402
R76
200
RES0402
RES0402
LD
VS
AD6655
Data Sheet
RES0402
10K OH M
R105
RES0402
10K OH M
R103
RES0402
10K OH M
R102
Rev. B | Page 69 of 84
Figure 90. Evaluation Board Schematic, DUT
D1B
D9A
DVDD1
D0B_LSB_
DVDD2
FD2B
FD3B
RPAK8
D13A_MSB_
FD0B
FD1B
FD0A
FD1A
SPI_CSB
SYNC
FD2A
CLK-
FD3A
CLK+
57
52
50
51
49
C137
0.001U
D2B
U1
SPI_SCLK/DFS
DRVDD1
C121
0.1U
C120
0.1U
AVDD3
D8A
C109
0.1U
C40
0.1U
48
RES040 2
SPI_SDIO/DCS
D4B
C122
0.001U
C126
0.001U
SPI_SCLK
47
46
AVDD2
VIN+B
0.001U
D3B
C127
0.001U
R64
AVDD
SPI_SDIO
45
44
43
VIN-B
AD6655
C36
0.1U
C35
DRVDD
D7A
D1A
D2A
D3A
D4A
63
61
62
59
60
58
55
56
53
54
R57
22ohm
9
10
11
12
13
14
15
16
RPAK8
DRVDD
D6B
D7B
D8B
D9B
D10B
D11B
D12B
D13B_MSB_
DCOB
DCOA
DOA_LSB_
64
AVDD
1
AVDD
VIN+B
VIN-B
42
RBIAS
CML
SENSE
TP3
DRGND1
DVDD
TP6
R63RES0402
0 OHM
10KOHM
0.1U
41
40
D12A
CML
VREF
VIN-A
D11A
AVDD
RES0402
C32
39
38
AVDD1
VIN+A
1
D10A
TP5
VIN-A
37
32
36
31
VIN+A
AVDD
28
PWR_SDFS
27
PWR_SCLK_PDWN
26
C14
0.1U
PWR_SDFS
R62
RES0402
25
35
R112
PWR_SCL
K
PWR_SDO
FD3A
FD2A
FD1A
FD0A
8
7
6
5
4
3
2
1
30
PWR_SDO/OEB
0 OHM
R115
0 OHM
24
34
33
RES0402
23
1
C15
1U
J4 - INSTALLFOR 0.5V VREF/IV INPUTSPAN
J5 - INSTALLFORIV VREF/2VINPUTSPAN
J6 - INSTALLFOR EXTERNALREFERENCEMODE
J7 - INSTALLFORPDWN
J8 - INSTALLFOROUTPUTDISABLE
R113
22ohm
29
9
10
11
12
13
14
15
16
0 OHM
21
D5B
CLK-
22
DRGND
CLK+
DVDD
20
D6A
SPI_CSB
19
D5A
SYNC
17
DVDD
18
8
7
6
5
4
3
2
1
1
2
3
4
5
6
7
8
9
DRVDD
10
11
12
13
14
15
16
D1B
D0B
FD3B
FD2B
FD1B
FD0B
R58 5
6
7
8
C34
R59
RPAK8
22ohm
R60
RPAK8
22ohm
RPAK4
22ohm
4
3
2
1
0.001U
9
10
11
12
13
14
15
16
9
10
11
12
13
14
15
16
8
7
6
5
4
3
2
1
0.1U
R61
RPAK8
22ohm
C33
9
10
11
12
13
14
15
16
8
7
6
5
4
3
2
1
8
7
6
5
4
3
2
1
D5B
D4B
D3B
D2B
D13A
D12A
D11A
D10A
D9A
D8A
D7A
D6A
D13B
D12B
D11B
D10B
D9B
D8B
D7B
D6B
D5A
D4A
D3A
D2A
D1A
D0A
DCOA
DCOB
06709-205
DRVDD
Data Sheet
AD6655
Figure 91. Evaluation Board Schematic, Digital Output Interface
FD1B
FD0B
V_DIG
D1B
D0B
FD3B
FD2B
D5B
D4B
V_DIG
D3B
D2B
D7B
D6B
D9B
D8B
D13B
D12B
V_DIG
D11B
D10B
D1A
D0A
DCOA
DCOB
D5A
D4A
V_DIG
D3A
D2A
D7A
D6A
D9A
D8A
D13A
D12A
V_DIG
D11A
D10A
FD3A
FD2A
FD1A
FD0A
V_DIG
PWR_SDO
PWR_SDFS
PWR_SCLK
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
U17
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
74VCX162244MTD
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
74VCX162244MTD
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
U16
74VCX162244MTD
V_DIG
V_DIG
V_DIG
V_DIG
SDO_OUT
SDFS_OUT
SCLK_OUT
OUT6P
OUT6N
J11
BG1
BG2
BG3
BG4
BG5
BG6
BG7
BG8
BG9
BG10
DG1
DG2
DG3
DG4
DG5
DG6
DG7
DG8
DG9
DG10
CSB
TYCO_HM-ZD
CHANNELB
B1
C10
D10
C9
D9
A9
B9
C8
D8
A8
B8
C7
D7
A7
B7
C6
D6
A6
B6
A10
B10
C5
D5
A5
B5
C4
D4
A4
B4
C3
D3
A3
B3
C2
D2
A2
B2
C1
D1
A1
CSB_2
SCLK
TYCO_HM-ZD
J10
BG1
BG2
BG3
BG4
BG5
BG6
BG7
BG8
BG9
BG10
DG1
DG2
DG3
DG4
DG5
DG6
DG7
DG8
DG9
DG10
R140
RES0402
0 OHM
R145
RES0402
0 OHM
R144
VS
0 OHM
SCLK_OUT
OUT6N
TP22
TEST
1
TP23 TEST
1
TP24 TEST
1
OUT6P
SYNC
SDI
RES0402
R143
0 OHM
SDO_OUT
SDFS_OU
T
RES0402
0 OHM
R142
RES0402
0 OHM
R119
R141
RES0402
SDO
RESETB
TP21
TEST
1
VS
R118
CHANNELA
B1
C10
D10
C9
D9
A9
B9
C8
D8
A8
B8
C7
D7
A7
B7
C6
D6
A6
B6
A10
B10
C5
D5
A5
B5
C4
D4
A4
B4
C3
D3
A3
B3
C2
D2
A2
B2
C1
D1
A1
A1
D1
C1
B2
A2
D2
C2
B3
A3
D3
C3
B4
A4
D4
C4
B5
A5
D5
C5
B10
A10
B6
A6
D6
C6
B7
A7
D7
C7
B8
A8
D8
C8
B9
A9
D9
C9
D10
C10
B1
J12
DG10
DG9
DG8
DG7
DG6
DG5
DG4
DG3
DG2
DG1
BG10
BG9
BG8
BG7
BG6
BG5
BG4
BG3
BG2
BG1
TYCO_HM-ZD
V_DIG
V_DIG
C65
0.1U
C66
0.1U
C72
0.1U
C67
0.1U
C73
0.1U
C68
0.1U
C74
0.1U
C69
0.1U
C75
0.1U
C70
0.1U
C76
0.1U
C71
0.1U
06709-206
DIGITAL/HSC-ADC-EVALCZ INTERFACE
10KOHM
24
23
22
21
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
RES040 2
U15
R130
VAL
R77
Rev. B | Page 70 of 84
100OHM
25
26
27
28
29
30
31
32
33
34
35
36
37
38
39
40
41
42
43
44
45
46
47
48
AD6655
Data Sheet
RES040 2
10KOHM
Rev. B | Page 71 of 84
CSB
SDO
SDI
SCLK
CSB_2
V_DIG
RES0402
CSB
SCLK
10KOHM
R18
C13
0.1U
R24
A2
GND
A1
3
2
1
RES0402
A2
Y1
U8
Y2
Y2
VCC
NC7WZ16P6X
GND
A1
Y1
VCC
NC7WZ07P6X
RES0603
10KOHM
3
2
1
RES040 2
R19
U7 1KOHM
4
5
6
4
5
6
SDO
V_DIG
C81
0.1U
V_DIG
V_DIG
V_DIG
RES0603
1KOHM
R21
RES0603
1KOHM
R20
R17
RES0603
100KOHM
VS
V_DIG
R22
3
RES0603
SPI_CSB
100KOHM
R23
RES0603
100KOHM
J2
3
1
J1
SPI_SCLK
SPI_SDIO
1
V_DIG
J1 - JUMPERPINS 2 TO 3 FOR SPI OPERATION
JUMPERPINS 1 TO 2 FOR DCS ENABLE
J2 - JUMPERPINS 2 TO 3 FOR SPI OPERATION
JUMPERPINS 1 TO 2 FOR TWOS COMPLIMENT OUTPUT
J21 - INSTALLJUMPERFOR SPI OPERATION
06709-207
SDI
Data Sheet
AD6655
Figure 92. Evaluation Board Schematic, SPI Circuitry
10KOHM
R65
Rev. B | Page 72 of 84
Figure 93. Evaluation Board Schematic, Power Supply
P4
P3
P2
P1
P4
P66
P55
P44
P33
P22
P3
P11
VCP
VS
DRVDDI N
1
1
AVDDI N
SMDC110F
C41
10U
F2
OPTIONAL POWER SUPPLY INPUT S
POWER_JACK
2
L6
IND1210
10U H
10u h
L10
IND1210
L9
IND1210
10U H
1
2
2
CR7
2
C53
10U
C102
10U
C52
10U
BNX-01 6
3 PSG
1 BIAS
C58
0.1U
C103
0.1U
C57
0.1U
CG6
CG5
CG 4
CB 2
1
10u h
IND1210
L11
1
2
DRVD D
2
PWR_I N
DVDD
AVD D
R16
1
3
2
1
CR8
C54
10U
RES0603
SHOT_RECT
261 OH M
TP25
1
C59
0.1U
1
2
V_DI G
CR10
S2A_REC T
1
CR11
S2A_REC T
1
C42
1U
SJ35
TP13
1
TP12
1
TP10
1
1TP9
1TP4
SD
6
8 IN
7 IN2
ADP3334
2
C44
1U
CR12
S2A_REC T
GND TEST POINT S
2
3
VR3
PAD
5
GND
OUT
VR1
OUT 1
OUT2 2
FB 3
IN
4
GND
1
F1
C43
1U
1
10u h
1.8
2.5
3.3
DRVD D
R1 3
76.8 K
107 K
140 K
2
147 K
94.0 K
78.7 K
R1 4
C93
0.001U
L3
IND1210
DRVDD SETTIN G
ADP3339
R13
R14
J16
76.8 KOH M
147K OH M
S2A_REC T
C45
1U
AVDDI N
1
10u h
L4
IND1210
2
DRVDDI N
06709-208
POWER INPU T
6V, 2A MAX
AD6655
Data Sheet
PWR_IN
PWR_IN
Rev. B | Page 73 of 84
Figure 94. Evaluation Board Schematic, Power Supply (Continued)
PAD
ADP333 9
VCP
5
C119
10U
GND
OUT 1
OUT2 2
FB 3
VR2
OUT
OUT
C124
10U
VS_OUT_D
R
Power Supply ByPass Capaci tors
VCP
SD
6
8 IN
7 IN2
ADP3334
C132
1U
IN
VR6
C135
1U
3
C133
1U
PAD
ADP333 9
4
GND
1
4
GND
1
IN
VS
C136
1U
C134
1U
C118
10U
R25
VR5
140KOHM
R15
0.001U
C95
SJ36
78.7KOHM
1
1
1
2
2
C131
1U
L13
IND1210
10uh
L12
IND1210
10uh
L8
IND1210
10UH
2
VS
VCP
VS
VS_OUT_DR
C110
0.1U
PWR_IN
C112
0.1U
C108
0.1U
C129
1U
3
IN
PAD
ADP333 9
C111
0.1U
VR4
4
GND
1
3
C115
0.1U
OUT
C114
0.1U
C113
0.1U
C130
1U
1
2
C107
0.1U
L1
IND1210
10UH
C116
0.1U
C105
0.1U
AMPVDD
06709-209
PWR_IN
Data Sheet
AD6655
SJ37
AD6655
Data Sheet
06709-100
EVALUATION BOARD LAYOUTS
Figure 95. Evaluation Board Layout, Primary Side
Rev. B | Page 74 of 84
AD6655
06709-101
Data Sheet
Figure 96. Evaluation Board Layout, Ground Plane
Rev. B | Page 75 of 84
Data Sheet
06709-102
AD6655
Figure 97. Evaluation Board Layout, Power Plane
Rev. B | Page 76 of 84
AD6655
06709-103
Data Sheet
Figure 98. Evaluation Board Layout, Power Plane
Rev. B | Page 77 of 84
Data Sheet
06709-104
AD6655
Figure 99. Evaluation Board Layout, Ground Plane
Rev. B | Page 78 of 84
AD6655
06709-105
Data Sheet
Figure 100. Evaluation Board Layout, Secondary Side (Mirrored Image)
Rev. B | Page 79 of 84
Data Sheet
06709-106
AD6655
Figure 101. Evaluation Board Layout, Silkscreen, Primary Side
Rev. B | Page 80 of 84
AD6655
06709-107
Data Sheet
Figure 102. Evaluation Board Layout, Silkscreen, Secondary Side
Rev. B | Page 81 of 84
AD6655
Data Sheet
BILL OF MATERIALS
Table 30. Evaluation Board Bill of Materials (BOM) 1, 2
Item
1
2
Qty
1
55
3
1
Reference
Designator
AD6655CE_REVB
C1 to C3, C6, C7,
C13, C14, C17, C18,
C20 to C26, C32,
C57 to C61, C65
to C76, C81 to
C83, C96 to C101,
C103, C105, C107,
C108, C110 to
C116, C145
C80
4
2
C5, C84
5
10
6
13
7
10
8
1
C33, C35, C63,
C93 to C95, C122,
C126, C127, C137
C15, C42 to C45,
C129 to C136
C27, C41, C52 to
C54, C62, C102,
C118, C119, C124
CR5
9
2
CR6, CR9
10
4
11
1
12
13
Description
PCB
0.1 µF, 16 V ceramic
capacitor, SMT 0402
Package
PCB
C0402SM
Manufacturer
Analog Devices
Murata
GRM155R71C104KA88D
18 pF, COG, 50 V, 5% ceramic
capacitor, SMT 0402
4.7 pF, COG, 50 V, 5% ceramic
capacitor, SMT 0402
0.001 µF, X7R, 25 V, 10%
ceramic capacitor, SMT 0402
C0402SM
Murata
GJM1555C1H180JB01J
C0402SM
Murata
GJM1555C1H4R7CB01J
C0402SM
Murata
GRM155R71H102KA01D
1 µF, X5R, 25 V, 10% ceramic
capacitor, SMT 0805
10 µF, X5R, 10 V, 10% ceramic
capacitor, SMT 1206
C0805
Murata
GR4M219R61A105KC01D
C1206
Murata
GRM31CR61C106KC31L
Schottky diode HSMS2822, SOT23
SOT23
Avago Technologies
HSMS-2822-BLKG
LED RED, SMT, 0603, SS-type
LED0603
Panasonic
LNJ208R8ARA
CR7, CR10 to CR12
50 V, 2 A diode
DO_214AA
Micro Commercial Components
S2A-TP
CR8
30 V, 3 A diode
DO_214AB
Micro Commercial Components
SK33-TP
1
F1
EMI filter
FLTHMURATABNX01
Murata
BNX016-01
1
F2
L1206
Tyco Raychem
NANOSMDC150F-2
14
2
J1, J2
HDR3
Samtec
TWS-1003-08-G-S
15
9
HDR2
Samtec
TWS-102-08-G-S
16
3
J4 to J9, J18, J19,
J21
J10 to J12
6.0 V, 3.0 A, trip current
resettable fuse
3-pin, male, single row,
straight header
2-pin, male, straight header
Interface connector
TYCO_HM_ZD
Tyco
6469169-1
17
1
J14
CNBERG2X4H350LD
Samtec
TSW-104-08-T-D
18
1
J16
8-pin, male, double row,
straight header
DC power jack connector
PWR_JACK1
Cui Stack
PJ-002A
19
10
10 µH, 2 A bead core, 1210
1210
Panasonic
EXC-CL3225U1
20
1
L1, L3, L4, L6, L8
to L13
P3
6-terminal connector
PTMICRO6
Weiland Electric, Inc.
Z5.531.3625.0
21
1
P4
4-terminal connector
PTMICRO4
Weiland Electric, Inc.
Z5.531.3425.0
22
3
R7, R30, R45
R0603
NIC Components
NRC06F57R6TRF
23
27
R0402SM
NIC Components
NRC04ZOTRF
24
1
R2, R3, R4, R32,
R33, R42, R64, R67,
R69, R90, R96, R99,
R101, R104, R110
to F113, R115,
R119, R121, R123,
R141 to R145
R13
57.6 Ω, 0603, 1/10 W,
1% resistor
0 Ω, 1/16 W, 5% resistor
76.8 kΩ, 0603, 1/10 W, 1% resistor
R0603
NIC Components
NRC06F7682TRF
25
1
R25
140 kΩ, 0603, 1/10 W, 1% resistor
R0603
NIC Components
NRC06F1403TRF
26
1
R14
147 kΩ, 0603, 1/10 W, 1% resistor
R0603
NIC Components
NRC06F1473TRF
27
1
R15
78.7 kΩ, 0603, 1/10 W, 1% resistor
R0603
NIC Components
NRC06F7872TRF
Rev. B | Page 82 of 84
Mfg. Part Number
Data Sheet
AD6655
Item
28
Qty
1
Reference
Designator
R16
Description
261 Ω, 0603, 1/10 W, 1% resistor
Package
R0603
Manufacturer
NIC Components
Mfg. Part Number
NRC06F2610TRF
29
3
R17, R22, R23
100 kΩ, 0603, 1/10 W, 1% resistor
R0603
NIC Components
NRC06F1003TRF
30
7
10 kΩ, 0402, 1/16 W, 1% resistor
R0402SM
NIC Components
NRC04F1002TRF
31
3
R18, R24, R63, R65,
R82, R118, R140
R19, R21
1 kΩ, 0603, 1/10 W, 1% resistor
R0603
NIC Components
NRC06F1001TRF
32
9
33 Ω, 0402, 1/16 W, 5% resistor
R0402SM
NIC Components
NRC04J330TRF
33
5
R26, R27, R43,
R46, R47, R70,
R71, R73, R74
R57, R59 to R62
22 Ω, 16-pin, 8-resistor,
resistor array
R_742
CTS Corporation
742C163220JPTR
34
1
R58
RES_ARRY
CTS Corporation
742C083220JPTR
35
1
R76
22 Ω, 8-pin, 4-resistor,
resistor array
200 Ω, 0402, 1/16 W, 1% resistor
R0402SM
NIC Components
NCR04F2000TRF
36
4
S2, S3, S5, S12
SMA_EDGE
1
SJ35
SLDR_PAD2MUYLAR
Emerson Network
Power
NIC Components
142-0701-201
37
SMA, inline, male,
coaxial connector
0 Ω, 1/8 W, 1% resistor
NRC10ZOTRF
38
5
T1 to T5
Balun
TRAN6B
M/A-COM
MABA-007159-000000
39
1
U1
IC, AD6655
LFCSP64-9X9-9E
Analog Devices
AD6655BCPZ
40
1
U2
Clock distribution, PLL IC
LFCSP64-9X9
Analog Devices
AD9516-4BCPZ
41
1
U3
Dual inverter IC
SC70_6
Fairchild Semiconductor
NC7WZ04P6X_NL
42
1
U7
SC70_6
Fairchild Semiconductor
NC7WZ07P6X_NL
43
1
U8
Dual buffer IC,
open-drain circuits
UHS dual buffer IC
SC70_6
Fairchild Semiconductor
NC7WZ16P6X_NL
44
3
U15 to U17
16-bit CMOS buffer IC
TSOP48_8_1MM
Fairchild Semiconductor
74VCX16244MTDX_NL
45
2
VR1, VR2
Adjustable regulator
LFCSP8-3X3
Analog Devices
ADP3334ACPZ
46
1
VR3
1.8 V high accuracy regulator
SOT223-HS
Analog Devices
ADP3339AKCZ-1.8
47
1
VR4
5.0 V high accuracy regulator
SOT223-HS
Analog Devices
ADP3339AKCZ-5.0
48
2
VR5, VR6
3.3 V high accuracy regulator
SOT223-HS
Analog Devices
ADP3339AKCZ-3.3
49
1
Y1
Oscillator clock, VFAC3
OSC-CTS-CB3
Valpey Fisher
VFAC3-BHL
50
2
Z1, Z2
High speed IC, op amp
LFCSP16-3X3-PAD
Analog Devices
AD8352ACPZ
1
2
This bill of materials is RoHS compliant.
The bill of materials lists only those items that are normally installed in the default condition. Items that are not installed are not included in the BOM.
Rev. B | Page 83 of 84
AD6655
Data Sheet
OUTLINE DIMENSIONS
9.10
9.00 SQ
8.90
0.30
0.25
0.18
0.60 MAX
0.60
MAX
64
48 49
1
PIN 1
INDICATOR
PIN 1
INDICATOR
8.85
8.75 SQ
8.65
0.50
BSC
0.50
0.40
0.30
33
32
0.22 MIN
7.50 REF
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
SEATING
PLANE
16
0.20 REF
FOR PROPER CONNECTION OF
THE EXPOSED PAD, REFER TO
THE PIN CONFIGURATION AND
FUNCTION DESCRIPTIONS
SECTION OF THIS DATA SHEET.
COMPLIANT TO JEDEC STANDARDS MO-220-VMMD-4
06-15-2012-B
12° MAX
17
BOTTOM VIEW
TOP VIEW
1.00
0.85
0.80
7.55
7.50 SQ
7.45
EXPOSED
PAD
Figure 103. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
9 mm × 9 mm Body, Very Thin Quad
(CP-64-6)
Dimensions shown in millimeters
ORDERING GUIDE
Model 1
AD6655ABCPZ-150
AD6655ABCPZ-125
AD6655ABCPZ-105
AD6655ABCPZ-80
AD6655ABCPZRL7-150
AD6655ABCPZRL7-125
AD6655-125EBZ
AD6655-150EBZ
1
Temperature Range
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
−40°C to +85°C
Package Description
64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
Evaluation Board with AD6655 and Software
Evaluation Board with AD6655 and Software
Z = RoHS Compliant Part.
©2007–2014 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06709-0-1/14(B)
Rev. B | Page 84 of 84
Package Option
CP-64-6
CP-64-6
CP-64-6
CP-64-6
CP-64-6
CP-64-6