Quad 8-Bit, 65 MSPS,
Serial LVDS 3 V A/D Converter
AD9289
FEATURES
FUNCTIONAL BLOCK DIAGRAM
Four ADCs in one package
Serial LVDS digital output data rates to 520 Mbps (ANSI-644)
Data and frame clock outputs
SNR = 48 dBc (to Nyquist)
Excellent linearity
DNL = ±0.2 LSB (typical)
INL = ±0.25 LSB (typical)
300 MHz full power analog bandwidth
Power dissipation = 112 mW/channel at 65 MSPS
1 Vp-p to 2 Vp-p input voltage range
3.0 V supply operation
Power-down mode
Digital test pattern enable for timing alignments
AVDD
DFS
PDWN
DTP
DRVDD
DRGND
AD9289
VIN+A
VIN–A
SHA
PIPELINE
ADC
SHA
PIPELINE
ADC
SHA
PIPELINE
ADC
SHA
PIPELINE
ADC
VIN+B
VIN–B
VIN+C
VIN–C
VIN+D
VIN–D
8
8
8
8
SERIAL
LVDS
D1+A
SERIAL
LVDS
D1+B
SERIAL
LVDS
D1+C
SERIAL
LVDS
D1+D
D1–D
FCO–
REF
SELECT
DATA RATE
MULTIPLIER
REFT_B
DCO+
DCO–
REFB_B
SHARED_REF
AGND
LVDSBIAS
CML
CLK+
CLK–
03682-001
Tape drives
Medical imaging
REFB_A
D1–C
FCO+
0.5V
APPLICATIONS
D1–B
LOCK
VREF
SENSE
REFT_A
D1–A
Figure 1.
PRODUCT DESCRIPTION
PRODUCT HIGHLIGHTS
The AD9289 is a quad 8-bit, 65 MSPS analog-to-digital converter (ADC) with an on-chip sample-and-hold circuit that is
designed for low cost, low power, small size, and ease of use.
The product operates at up to a 65 MSPS conversion rate and is
optimized for outstanding dynamic performance where a small
package size is critical.
1.
Four ADCs are contained in a small, space-saving package.
2.
A data clock out (DCO) is provided, which operates up to
260 MHz and supports double-data rate operation (DDR).
3.
The outputs of each ADC are serialized LVDS with data
rates up to 520 Mbps (8 bits × 65 MSPS).
4.
The AD9289 operates from a single 3.0 V power supply.
5.
The internal clock duty cycle stabilizer maintains
performance over a wide range of input clock duty cycles.
The ADC requires a single, 3 V power supply and an LVDScompatible sample rate clock for full performance operation.
No external reference or driver components are required for
many applications.
The ADC automatically multiplies the sample rate clock for the
appropriate LVDS serial data rate. A data clock (DCO) for
capturing data on the output and a frame clock (FCO) trigger
for signaling a new output byte are provided. Power-down is
supported. The ADC typically consumes 7 mW when enabled.
Fabricated on an advanced CMOS process, the AD9289 is
available in a 64-ball mini-BGA package (64-BGA). It is
specified over the industrial temperature range of –40°C
to +85°C.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable.
However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2004 Analog Devices, Inc. All rights reserved.
AD9289
TABLE OF CONTENTS
Specifications..................................................................................... 3
Typical Performance Characteristics ..............................................9
AC Specifications.......................................................................... 4
Terminology .................................................................................... 12
Digital Specifications ................................................................... 4
Theory of Operation ...................................................................... 14
Switching Specifications .............................................................. 5
Analog Input and Reference Overview ................................... 14
Timing Diagrams.......................................................................... 5
Clock Input and Considerations .............................................. 15
Absolute Maximum Ratings............................................................ 6
Evaluation Board ............................................................................ 20
Explanation of Test Levels........................................................... 6
Outline Dimensions ....................................................................... 30
ESD Caution.................................................................................. 6
Ordering Guide .......................................................................... 30
Pin Configuration and Function Descriptions............................. 7
Equivalent Circuits ........................................................................... 8
REVISION HISTORY
10/04—Initial Version: Revision 0
Rev. 0 | Page 2 of 32
AD9289
SPECIFICATIONS
AVDD = 3.0 V, DRVDD = 3.0 V, conversion rate = 65 MSPS, 2 V p-p differential input, 1.0 V internal reference, AIN = –0.5 dBFS, unless
otherwise noted.
Table 1.
Parameter
RESOLUTION
ACCURACY
No Missing Codes
Offset Error
Offset Matching
Gain Error 1
Gain Matching1
Differential Nonlinearity (DNL)
Integral Nonlinearity (INL)
TEMPERATURE DRIFT
Offset Error
Gain Error
Reference Voltage (VREF = 1 V)
REFERENCE
Output Voltage Error (VREF = 1 V)
Load Regulation @ 1.0 mA (VREF = 1 V)
Output Voltage Error (VREF = 0.5 V)
Load Regulation @ 0.5 mA (VREF = 0.5 V)
Input Resistance
COMMON MODE
Common-Mode Level Output
ANALOG INPUTS
Differential Input Voltage Range (VREF = 1 V)
Differential Input Voltage Range (VREF = 0.5 V)
Common-Mode Voltage
Input Capacitance
Analog Bandwidth, Full Power
POWER SUPPLY
AVDD
DRVDD
IAVDD
DRVDD
Power Dissipation2
Power-Down Dissipation
CROSSTALK
1
2
2
1
2
Temperature
Test Level
Min
8
Full
Full
Full
Full
Full
25°C
Full
25°C
Full
VI
VI
VI
VI
VI
V
VI
V
VI
Guaranteed
±5
±12
±0.5
±0.2
±0.2
±0.2
±0.25
±0.25
Full
Full
Full
V
V
V
±16
±40
±10
Full
Full
Full
Full
Full
VI
V
VI
V
V
±10
0.7
±8
0.2
7
±35
±26
mV
mV
mV
mV
kΩ
Full
VI
±1.5
±50
mV
Full
Full
Full
Full
Full
VI
VI
V
V
V
2
1
1.5
5
300
Full
Full
Full
Full
Full
Full
Full
IV
IV
VI
VI
VI
VI
V
2.7
2.7
Typ
3.0
3.0
150
33
550
7
–75
Max
Unit
Bits
±57
±68
±2.5
±0.9
mV
mV
% FS
% FS
LSB
LSB
LSB
LSB
±0.6
±0.6
ppm/°C
ppm/°C
ppm/°C
V p-p
V p-p
V
pF
MHz
3.3
3.3
168
40
625
12
Gain error and gain temperature coefficients are based on the ADC only (with a fixed 1.0 V external reference and a 2 V p-p differential analog input).
Power dissipation measured with rated encode and 2.4 MHz analog input at –0.5 dBFS.
Rev. 0 | Page 3 of 32
V
V
mA
mA
mW
mW
dB
AD9289
AC SPECIFICATIONS
AVDD = 3.0 V, DRVDD = 3.0 V, conversion rate = 65 MSPS, 2 V p-p differential input, 1.0 V internal reference, AIN = –0.5 dBFS, unless
otherwise noted.
Table 2.
Parameter
SIGNAL-TO-NOISE RATIO (SNR)
fIN = 2.4 MHz
fIN = 10.3 MHz
fIN = 35 MHz
fIN = 2.4 MHz
fIN = 10.3 MHz
fIN = 35 MHz
fIN = 2.4 MHz
fIN = 10.3 MHz
fIN = 35 MHz
fIN = 2.4 MHz
fIN = 10.3 MHz
fIN = 35 MHz
fIN = 2.4 MHz
fIN = 10.3 MHz
fIN = 35 MHz
fIN = 2.4 MHz
fIN = 10.3 MHz
fIN = 35 MHz
fIN1 = 15 MHz
fIN2 = 16 MHz
SIGNAL-TO-NOISE RATIO (SINAD)
EFFECTIVE NUMBER OF BITS (ENOB)
SPURIOUS-FREE DYNAMIC RANGE (SFDR)
WORST HARMONIC (Second or Third)
WORST OTHER (Excluding Second or Third)
TWO TONE INTERMOD DISTORTION (IMD)
AIN1 and AIN2 = –7.0 dBFS
Temperature
Full
25°C
Full
Full
25°C
Full
Full
25°C
Full
Full
25°C
Full
Full
25°C
Full
Full
25°C
Full
25°C
Test Level
IV
V
VI
IV
V
VI
IV
V
VI
IV
V
VI
IV
V
VI
IV
V
VI
V
Min
47.7
46.7
47.6
46.2
7.6
7.4
61.0
54.0
Typ
49.0
48.5
48.0
48.9
48.4
47.5
7.8
7.7
7.6
70.0
68.0
65.0
–75.0
–70.0
–65.0
–70.0
–68.0
–65.0
–72.0
Max
–61.0
–54.0
–61.0
–57.5
Unit
dB
dB
dB
dB
dB
dB
Bits
Bits
Bits
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
dBc
DIGITAL SPECIFICATIONS
AVDD = 3.0 V, DRVDD = 3.0 V, conversion rate = 65 MSPS, 2 V p-p differential input, 1.0 V internal reference, AIN = –0.5 dBFS, unless
otherwise noted.
Table 3.
Parameter
CLOCK INPUTS1 (CLK+, CLK–)
Logic Compliance
Differential Input Voltage
High Level Input Current
Low Level Input Current
Input Common-Mode Voltage
Input Resistance
Input Capacitance
LOGIC INPUTS (DFS, PDWN, SHARED_REF)
Logic 1 Voltage
Logic 0 Voltage
Input Resistance
Input Capacitance
LOGIC OUTPUTS (LOCK)
Logic 1 Voltage
Logic 0 Voltage
DIGITAL OUTPUTS (D1+, D1–)
Logic Compliance
Differential Output Voltage
Output Offset Voltage
Output Coding
1
Temperature
Test Level
Full
Full
Full
Full
25°C
25°C
IV
VI
VI
IV
V
V
Full
Full
25°C
25°C
IV
IV
V
V
2.0
Full
Full
IV
IV
2.45
Full
Full
Full
VI
VI
VI
Clock inputs are LVDS-compatible. They require external dc bias and cannot be ac-coupled.
Rev. 0 | Page 4 of 32
Min
LVDS
250
1.125
Typ
Max
Unit
350
30
30
1.25
100
2
450
75
75
1.375
mV p-p
µA
µA
V
kΩ
pF
0.8
30
4
0.05
LVDS
260
1.15
350
440
1.25
1.35
Twos complement or binary
V
V
kΩ
pF
V
V
mV
V
AD9289
SWITCHING SPECIFICATIONS
AVDD = 3.0 V, DRVDD = 3.0 V, conversion rate = 65 MSPS, 2 V p-p differential input, 1.0 V internal reference, AIN = –0.5 dBFS, unless
otherwise noted.
Table 4.
Parameter
CLOCK
Maximum Clock Rate
Minimum Clock Rate
Clock Pulse Width High (tEH)
Clock Pulse Width Low (tEL)
OUTPUT PARAMETERS
Valid Time (tV)1
Propagation Delay (tPD)
Rise Time (tR) (20% to 80%)
Fall Time (tF) (20% to 80%)
FCO Propagation Delay (tFCO)
DCO Propagation Delay (tCPD)
DCO-to-Data Delay (tDATA)
DCO-to-FCO Delay (tFRAME)
Data-to-Data Skew (tDATA-MAX – tDATA-MIN)
PLL Lock Time (tLOCK)
Wake-Up Time
Pipeline Latency
APERTURE
Aperture Delay (tA)
Aperture Uncertainty (Jitter)
OUT-OF-RANGE RECOVERY TIME
1
Temp
Test Level
Min
Full
Full
Full
Full
VI
IV
VI
VI
65
Full
Full
Full
Full
Full
Full
Full
Full
Full
25°C
25°C
Full
IV
VI
V
V
V
V
VI
VI
IV
V
V
IV
25°C
25°C
25°C
V
V
V
Typ
Max
12
6.9
6.9
0.5
6.9
7.7
7.7
9.0
250
250
9.0
9.0
±100
±100
±100
1.8
7
6
<1.5
11.6
±550
±500
±250
4.5
<1
1
TIMING DIAGRAMS
N-1
AIN
N
tEH
CLK–
tEL
CLK+
tCPD
DCO–
STATIC
STATIC
DCO+
tFRAME
tFCO
FCO–
STATIC
STATIC
FCO+
tPD
D1–
STATIC
INVALID
D1+
tDATA
MSB D6
D5
D4
D3
D2
D1 LSB MSB
(N-7) (N-7) (N-7) (N-7) (N-7) (N-7) (N-7) (N-7) (N-6)
STATIC
03682-003
LOCK
tV
Figure 2. Timing Diagram
Rev. 0 | Page 5 of 32
MSPS
MSPS
ns
ns
CLK cycles
ns
ps
ps
ns
ns
ps
ps
ps
µs
ms
CLK cycles
ns
ps rms
CLK cycles
Actual valid time is dependent on the moment when LOCK goes low.
tA
Unit
AD9289
ABSOLUTE MAXIMUM RATINGS
Table 5.
Parameter
ELECTRICAL
AVDD
DRVDD
AGND
AVDD
Digital Outputs (D1+,
D1–, DCO+, DCO–,
FCO+, FCO–)
LOCK, LVDSBIAS
CLK+, CLK–
VIN+, VIN–
PDWN, DFS, DTP
REFT, REFB,
SHARED_REF, CML
VREF, SENSE
ENVIRONMENTAL
Operating
Temperature Range
(Ambient)
Maximum Junction
Temperature
Lead Temperature
(Soldering, 10 sec)
Storage
Temperature Range
(Ambient)
Thermal
Impedance1
EXPLANATION OF TEST LEVELS
With
Respect
To
Min
Max
Unit
AGND
DRGND
DRGND
DRVDD
DRGND
–0.3
–0.3
–0.3
–3.9
–0.3
+3.9
+3.9
+0.3
+3.9
V
V
V
V
V
DRGND
AGND
AGND
AGND
AGND
–0.3
–0.3
–0.3
–0.3
–0.3
DRVDD
AVDD
AVDD
AVDD
AVDD
V
V
V
V
V
AGND
–0.3
AVDD
V
–40
+85
°C
150
°C
300
°C
+150
°C
40
°C/W
I. 100% production tested.
DRVDD
II. 100% production tested at 25°C and guaranteed by design
and characterization at specified temperatures.
III. Sample tested only.
IV. Parameter is guaranteed by design and characterization
testing.
V. Parameter is a typical value only.
–65
VI. 100% production tested at 25°C and guaranteed by design
and characterization for industrial temperature range.
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those listed in the operational sections
of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
1
θJA for a 4-layer PCB with solid ground plane in still air.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 6 of 32
AD9289
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
1 2 3 4 5 6 7 8
03682-005
A
B
C
D
E
F
G
H
Figure 3. BGA Top View (Looking Through)
Table 6. Pin Function Descriptions
Pin
No.
A1
B1
C1
Mnemonic
D1–A
D1+A
FCO+
D1
E1
F1
G1
H1
A2
B2
C2
DNC
AGND
VIN–A
VIN+A
LVDSBIAS1
DNC
DNC
FCO–
D2
E2
F2
G2
H2
A3
B3
C3
D3
E3
F3
G3
H3
A4
B4
C4
D4
DNC
AGND
AVDD
AGND
VIN+B
D1–B
D1+B
DRVDD
DRGND
AGND
CML
SHARED_REF3
VIN–B
DNC
DNC
DCO+
LOCK
Description
ADC A Complement Digital Output
ADC A True Digital Output
Frame Clock Output (MSB Indicator)
True Output
Do Not Connect
Analog Ground
ADC A Analog Input—Complement
ADC A Analog Input—True
LVDS Output Bias Pin
Do Not Connect
Do Not Connect
Frame Clock Output (MSB Indicator)
Complement Output
Do Not Connect
Analog Ground
Analog Supply
Analog Ground
ADC B Analog Input—True
ADC B Complement Digital Output
ADC B True Digital Output
Digital Supply
Digital Ground
Analog Ground
Common Mode Level Output ( = AVDD/2)
Shared Reference Control Bit
ADC B Analog Input—Complement
Do Not Connect
Do Not Connect
Data Clock Output—True
PLL Lock Output
E4
F4
G4
H4
A5
B5
C5
AVDD
REFT_A
REFB_A
SENSE
D1–C
D1+C
DCO–
Analog Supply
Reference Buffer Decoupling (Positive)
Reference Buffer Decoupling (Negative)
Reference Mode Selection
ADC C Complement Digital Output
ADC C True Digital Output
Data Clock Output—Complement
Pin
No.
D5
E5
F5
G5
H5
A6
B6
C6
D6
E6
F6
G6
H6
A7
B7
C7
D7
E7
F7
G7
H7
A8
B8
C8
D8
E8
F8
G8
H8
1
Mnemonic
AGND
AGND
REFT_B
REFB_B
VREF
DNC
DNC
DRVDD
DRGND
AVDD
AGND
AGND
VIN–C
D1–D
D1+D
DFS2
AGND
AGND
AVDD
AGND
VIN+C
DNC
DNC
CLK+
CLK–
PDWN3
VIN–D
VIN+D
DTP3, 4
Description
Analog Ground
Analog Ground
Reference Buffer Decoupling (Positive)
Reference Buffer Decoupling (Negative)
Voltage Reference Input/Output
Do Not Connect
Do Not Connect
Digital Supply
Digital Ground
Analog Supply
Analog Ground
Analog Ground
ADC C Analog Input—Complement
ADC D Complement Digital Output
ADC D True Digital Output
Data Format Select
Analog Ground
Analog Ground
Analog Supply
Analog Ground
ADC C Analog Input—True
Do Not Connect
Do Not Connect
Input Clock—True
Input Clock—Complement
Power Down Selection
ADC D Analog Input—Complement
ADC D Analog Input—True
Digital Test Pattern
LVDSBIAS use a 3.9 kΩ resistor-to-analog ground to set the LVDS output
differential swing of 350 mV p-p.
DFS has an internal on-chip pull-down resistor and defaults to offset binary
output coding if untied. If twos complement output coding is desired then
tie this pin to AVDD.
3
To enable, tie this pin to AVDD. To disable, tie this pin to AGND.
4
DTP has an internal on-chip pull-down resistor.
2
Rev. 0 | Page 7 of 32
AD9289
EQUIVALENT CIRCUITS
AVDD
DRVDD
VIN+, VIN–
V
V
AGND
D1+
V
V
DRGND
Figure 4. Equivalent Analog Input Circuit
Figure 7. Equivalent Digital Output Circuit
DRVDD
AVDD
LOCK
CLK+, CLK–
03682-007
AGND
DRGND
Figure 8. Equivalent LOCK Output Circuit
Figure 5. Equivalent Clock Input Circuit
DRVDD
03682-008
375Ω
DRGND
03682-010
25Ω
375Ω
DFS, PDWN,
SHARED_REF
03682-009
03682-006
D1–
Figure 6. Equivalent Digital Input Circuit
Rev. 0 | Page 8 of 32
AD9289
TYPICAL PERFORMANCE CHARACTERISTICS
0
75
AIN = –0.5dBFS
SNR = 49.08dB
ENOB = 7.86 BITS
SFDR = 70.55dBc
1V p-p, SFDR (dBc)
65
–40
60
–60
55
03682-014
–100
0.0
2V p-p, SNR (dB)
50
4.1
8.1
12.2
16.3
20.3
FREQUENCY (MHz)
24.4
28.4
03682-017
–80
1V p-p, SNR (dB)
45
10
32.5
Figure 9. Single-Tone 32k FFT With fIN = 2.4 MHz, fSAMPLE = 65 MSPS
20
30
40
50
ENCODE (MSPS)
60
70
Figure 12. SNR/SFDR vs. fSAMPLE, fIN = 2.4 MHz
0
75
AIN = –0.5dBFS
SNR = 48.93dB
ENOB = 7.83 BITS
SFDR = 71.45dBc
–20
70
2V p-p, SFDR (dBc)
1V p-p, SFDR (dBc)
AIN = –0.5dBFS
65
–40
dB
60
–60
55
–80
03682-015
–100
0.0
4.1
8.1
12.2
16.3
20.3
FREQUENCY (MHz)
24.4
28.4
1V p-p, SNR (dB)
45
10
32.5
Figure 10. Single-Tone 32k FFT With fIN = 10.3 MHz, fSAMPLE = 65 MSPS
20
30
40
50
ENCODE (MSPS)
60
70
Figure 13. SNR/SFDR vs. fSAMPLE, fIN = 10.3 MHz
0
–20
2V p-p, SNR (dB)
50
03682-018
AMPLITUDE (dBFS)
AIN = –0.5dBFS
70
dB
AMPLITUDE (dBFS)
–20
2V p-p, SFDR (dBc)
75
2V p-p, SFDR (dBc)
AIN = –0.5dBFS
SNR = 48.8dB
ENOB = 7.8 BITS
SFDR = 68.5dBc
AIN = –0.5dBFS
70
dB
–40
60
–60
55
–100
0.0
2V p-p, SNR (dB)
50
4.1
8.1
12.2
16.3
20.3
FREQUENCY (MHz)
24.4
28.4
03682-019
–80
03682-016
AMPLITUDE (dBFS)
1V p-p, SFDR (dBc)
65
1V p-p, SNR (dB)
45
10
32.5
Figure 11. Single-Tone 32k FFT With fIN = 35 MHz, fSAMPLE = 65 MSP
20
30
40
50
ENCODE (MSPS)
60
Figure 14. SNR/SFDR vs. fSAMPLE, fIN = 35 MHz
Rev. 0 | Page 9 of 32
70
AD9289
75
75
60
70dB REFERENCE LINE
70
SFDR (dBc)
50
2V p-p, SNR (dB)
1V p-p, SFDR (dBc)
65
dB
dB
40
30
60
1V p-p, SNR (dB)
55
20
–35
03682-020
0
–40
50
2V p-p, SFDR (dBc)
–30
–25
–20
–15
–10
ANALOG INPUT LEVEL (dBFS)
–5
SNR (dB)
45
0.1
0
Figure 15. SNR/SFDR vs. Analog Input Level, fSAMPLE = 65 MSPS,
fIN = 2.4 MHz
1
10
FREQUENCY (MHz)
100
Figure 18. SNR/SFDR vs. fIN, fSAMPLE = 65 MHz
0
75
60
AIN1 AND AIN2 = –7.0dBFS
SFDR = 69.9dBc
IMD2 = 74.9dBc
IMD3 = 72.9dBc
70dB REFERENCE LINE
–20
2V p-p, SNR (dB)
1V p-p, SFDR (dBc)
AMPLITUDE (dBFS)
50
03682-023
10
dB
40
30
1V p-p, SNR (dB)
–40
–60
20
–35
03682-021
0
–40
2V p-p, SFDR (dBc)
–30
–25
–20
–15
–10
ANALOG INPUT LEVEL (dBFS)
–5
–100
0.0
0
Figure 16. SNR/SFDR vs. Analog Input Level, fSAMPLE = 65 MSPS,
fIN = 10.3 MHz
4.1
8.1
12.2
16.3
20.3
FREQUENCY (MHz)
24.4
28.4
32.5
Figure 19. Two-Tone 32k FFT with fIN1 = 15 MHz and fIN2 = 16 MHz,
fSAMPLE = 65 MSPS
75
80
60
50
03682-024
–80
10
70
70dB REFERENCE LINE
1V p-p, SFDR (dBc)
60
2V p-p, SNR (dB)
70dB REFERENCE LINE
50
dB
dB
40
30
40
1V p-p, SNR (dB)
SFDR (dBc)
30
20
20
2V p-p, SFDR (dBc)
–35
–30
–25
–20
–15
–10
ANALOG INPUT LEVEL (dBFS)
–5
Figure 17. SNR/SFDR vs. Analog Input Level, fSAMPLE = 65 MSPS,
fIN = 35 MHz
0
–40
0
03682-025
0
–40
10
03682-022
10
–35
–30
–25
–20
–15
–10
ANALOG INPUT LEVEL (dBFS)
–5
0
Figure 20. Two-Tone SFDR vs. Analog Input Level with fIN1 = 15 MHz and
fIN2 = 16 MHz, fSAMPLE = 65 MSPS
Rev. 0 | Page 10 of 32
AD9289
0.5
75
2V p-p, SFDR (dBc)
0.4
70
0.3
1V p-p, SFDR (dBc)
0.2
DNL (LSB)
60
55
0
–0.1
–0.2
–0.3
03682-026
2V p-p, SINAD (dB)
50
1V p-p, SINAD (dB)
45
–40
0.1
–20
0
20
49
TEMPERATURE (°C)
60
03682-028
dB
65
–0.4
–0.5
0
80
Figure 21. SINAD/SFDR vs. Temperature, fSAMPLE = 65 MSPS, fIN 10.3 MHz
32
64
96
128
CODE
160
192
224
256
Figure 23. Typical DNL, fIN = 2.4 MHz, fSAMPLE = 65 MSPS
0.5
15
0.4
0.3
0.2
5
INL (LSB)
SHARED REF MODE
(PIN TIED HIGH)
0
–5
0.1
0
–0.1
–0.2
SHARED REF MODE
(PIN TIED LOW)
–0.3
–15
–40
–20
0
20
49
TEMPERATURE (°C)
60
03682-029
–10
03682-027
GAIN ERROR (ppm/°C)
10
–0.4
–0.5
0
80
32
64
96
128
CODE
160
192
224
Figure 24. Typical INL, fIN = 2.4 MHz, fSAMPLE = 65 MSPS
Figure 22. Gain vs. Temperature
Rev. 0 | Page 11 of 32
256
AD9289
TERMINOLOGY
Analog Bandwidth
Effective Number of Bits (ENOB)
Analog Bandwidth is the analog input frequency at which the
spectral power of the fundamental frequency (as determined by
the FFT analysis) is reduced by 3 dB from full scale.
For a sine wave, SINAD can be expressed in terms of the
number of bits. Using the following formula, it is possible to
obtain a measure of performance expressed as N, the effective
number of bits:
Aperture Delay
N = (SINAD – 1.76)/6.02
Aperture delay is a measure of the sample-and-hold amplifier
(SHA) performance and is measured from the 50% point rising
edge of the clock input to the time at which the input signal is
held for conversion.
Thus, the effective number of bits for a device for sine wave
inputs at a given input frequency can be calculated directly
from its measured SINAD.
Aperture Uncertainty (Jitter)
Gain Error
Aperture jitter is the variation in aperture delay for successive
samples and can be manifested as frequency-dependent noise
on the ADC input.
The largest gain error is specified and is considered the
difference between the measured and ideal full-scale input
voltage range.
Clock Pulse Width and Duty Cycle
Gain Matching
Pulse width high is the minimum amount of time that the clock
pulse should be left in the Logic 1 state to achieve a rated
performance. Pulse width low is the minimum time the clock
pulse should be left in the low state. At a given clock rate, these
specifications define an acceptable clock duty cycle.
Expressed in %FSR. Computed using the following equation:
Crosstalk
where FSRMAX is the most positive gain error of the ADCs, and
FSRMIN is the most negative gain error of the ADCs.
Crosstalk is defined as the coupling of a channel when all
channels are driven by a full-scale signal.
GainMatching =
FSR max − FSR min
× 100%
max + FSR min ⎞
FSR
⎛
⎜
⎟
2
⎝
⎠
Second and Third Harmonic Distortion
Differential Analog Input Capacitance
The complex impedance simulated at each analog input port.
The ratio of the rms signal amplitude to the rms value of the
second or third harmonic component, reported in dBc.
Integral Nonlinearity (INL)
Differential Analog Input Voltage Range
The peak-to-peak differential voltage that must be applied to
the converter to generate a full-scale response. Peak differential
voltage is computed by observing the voltage on a pin and
subtracting the voltage from a second pin that is 180° out of
phase. Peak-to-peak differential is computed by rotating the
input phase 180° and taking the peak measurement again. The
difference is computed between both peak measurements.
Differential Nonlinearity (DNL, No Missing Codes)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed no
missing codes to an 8-bit resolution indicates that all 256 codes,
respectively, must be present over all operating ranges.
INL refers to the deviation of each individual code from a line
drawn from negative full scale through positive full scale The
point used as negative full scale occurs 1/2 LSB before the first
code transition. Positive full scale is defined as a level 1 1/2 LSB
beyond the last code transition. The deviation is measured from
the middle of each particular code to the true straight line.
Offset Error
The largest offset error is specified and is considered the
difference between the measured and ideal voltage at the analog
input that produces the midscale code at the outputs.
Rev. 0 | Page 12 of 32
AD9289
Offset Matching
Signal-to Noise and Distortion (SINAD) Ratio
Expressed in mV. Computed using the following equation:
SINAD is the ratio of the rms value of the measured input
signal to the rms sum of all other spectral components below
the Nyquist frequency, including harmonics but excluding dc.
The value for SINAD is expressed in decibels.
OffsetMatching = OFFMAX − OFFMIN
where OFFMAX is the most positive offset error and OFFMIN is the
most negative offset error.
Out-of-Range Recovery Time
Out-of-range recovery time is the time it takes for the ADC to
reacquire the analog input after a transient from 10% above
positive full scale to 10% above negative full scale, or from 10%
below negative full scale to 10% below positive full scale.
Output Propagation Delay
The delay between the clock logic threshold and the time when
all bits are within valid logic levels.
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the measured input signal to
the rms sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc. The value
for SNR is expressed in decibels.
Spurious-Free Dynamic Range (SFDR)
SFDR is the difference in dB between the rms amplitude of the
input signal and the peak spurious signal.
Temperature Drift
The temperature drift for offset error and gain error specifies
the maximum change from the initial (25°C) value to the value
at TMIN or TMAX.
Two-Tone SFDR
The ratio of the rms value of either input tone to the rms value
of the peak spurious component. The peak spurious component
may or may not be an IMD product. It may be reported in dBc
(i.e., degrades as signal levels are lowered) or in dBFS (always
related back to converter full scale).
Rev. 0 | Page 13 of 32
AD9289
THEORY OF OPERATION
The input stage contains a differential SHA that can be configured as ac- or dc-coupled in differential or single-ended modes.
The output-staging block aligns the data and carries out the
error correction. The data is serialized and aligned to the frame,
output clock, and lock detection circuitry.
The analog inputs of the AD9289 are not internally dc biased. In
ac-coupled applications, the user must provide this bias externally. Setting the device so that VCM = AVDD/2 is recommended
for optimum performance, but the device functions over a
wider range with reasonable performance (see Figure 26 and
Figure 27).
75
2V p-p, SFDR (dBc)
70
60
55
2V p-p, SNR (dB)
50
1V p-p, SNR (dB)
45
40
35
ANALOG INPUT AND REFERENCE OVERVIEW
0
The analog input to the AD9289 is a differential-switched
capacitor SHA that has been designed for optimum performance while processing a differential input signal. The SHA
input can support a wide common-mode range and maintain
excellent performance, as shown in Figure 26 sand Figure 27.
An input common-mode voltage of midsupply minimizes
signal dependent errors and provides optimum performance.
0.5
1.0
1.5
2.0
2.5
ANALOG INPUT COMMON-MODE VOLTAGE (V)
3.0
Figure 26. SNR, SFDR vs. Common-Mode Voltage, fIN = 2.4 MHz,
fSAMPLE = 65 MSPS
75
2V p-p, SFDR (dBc)
70
65
60
dB
H
S
1V p-p, SFDR (dBc)
65
03682-030
Each stage of the pipeline, excluding the last, consists of a low
resolution flash ADC connected to a switched capacitor digitalto-analog converter (DAC) and interstage residue amplifier
(MDAC). The MDAC magnifies the difference between the
reconstructed DAC output and the flash input for the next stage
in the pipeline. One bit of redundancy is used in each of the
stages to facilitate digital correction of flash errors. The last
stage simply consists of a flash ADC.
of a clock cycle. A small resistor in series with each input can
help reduce the peak transient current required from the output
stage of the driving source. Also, a small shunt capacitor can
be placed across the inputs to provide dynamic charging
currents. This passive network creates a low-pass filter at the
ADC’s input; therefore, the precise values are dependent on
the application.
dB
Each A/D converter in the AD9289 architecture consists of a
front send sample-and-hold amplifier (SHA) followed by a
pipe-lined, switched capacitor ADC. The pipelined ADC is
divided into two sections, consisting of six 1.5-bit stages and a
final 2-bit flash. Each stage provides sufficient overlap to correct
for flash errors in the preceding stages. The quantized outputs
from each stage are combined into a final 8-bit result in the
digital correc-tion logic. The pipelined architecture permits the
first stage to operate on a new input sample, while the
remaining stages operate on preceding samples. Sampling
occurs on the rising edge of the clock.
1V p-p, SFDR (dBc)
55
S
2V p-p, SNR (dB)
50
VIN+
CPAR
1V p-p, SNR (dB)
45
03682-031
40
S
35
VIN–
03682-051
0
CPAR
S
The clock signal alternately switches the SHA between sample
mode and hold mode (see Figure 25). When the SHA is
switched into sample mode, the signal source must be capable
of charging the sample capacitors and settling within one-half
3.0
Figure 27. SNR, SFDR vs. Common-Mode Voltage, fIN = 35 MHz,
fSAMPLE = 65 MSPS
H
Figure 25. Switched-Capacitor SHA Input UPDATE
0.5
1.0
1.5
2.0
2.5
ANALOG INPUT COMMON-MODE VOLTAGE (V)
For best dynamic performance, the source impedances driving
VIN+ and VIN− should be matched such that common-mode
settling errors are symmetrical. These errors are reduced by the
common-mode rejection of the ADC.
Rev. 0 | Page 14 of 32
AD9289
AVDD
R
2V p-p
VIN+
49.9Ω
C
R
AVDD
VIN–
AGND
1kΩ
REFT = 1/2 (AVDD + VREF)
REFB = 1/2 (AVDD − VREF)
Span = 2 × (REFT − REFB) = 2 × VREF
AD9289
03682-053
An internal reference buffer creates the positive and negative
reference voltages, REFT and REFB, respectively, that defines
the span of the ADC core. The output common-mode of the
reference buffer is set to midsupply, and the REFT and REFB
voltages and span are defined as
1kΩ
0.1µF
It can be seen from the equations above that the REFT and
REFB voltages are symmetrical about the midsupply voltage
and, by definition, the input span is twice the value of the VREF
voltage.
The internal voltage reference can be pin-strapped to fixed
values of 0.5 V or 1.0 V or adjusted within the same range, as
discussed in the Internal Reference Connection section.
Maximum SNR performance is achieved by setting the AD9289
to the largest input span of 2 V p-p.
Figure 29. Differential Transformer-Coupled Configuration
Single-Ended Input Configuration
A single-ended input may provide adequate performance in
cost-sensitive applications. In this configuration, there is a
degradation in SFDR and distortion performance due to the
large input common-mode swing. However, if the source
impedances on each input are matched, there should be little
effect on SNR performance. Figure 30 details a typical singleended input configuration.
The SHA should be driven from a source that keeps the signal
peaks within the allowable range for the selected reference
voltage. The minimum and maximum common-mode input
levels are defined in Figure 26 and Figure 27.
10µF
1kΩ
2V p-p
49.9Ω
R
0.1µF 1kΩ
Differential Input Configurations
C
AVDD
VIN+
AD9289
Optimum performance is achieved by driving the AD9289 in a
differential input configuration. For baseband applications, the
AD8351 differential driver provides excellent performance and
a flexible interface to the ADC (see Figure 28).
1kΩ
25Ω
1kΩ
GP1
VIN–
PWUP
AD8351
25Ω
AVDD
0.1µF R
VCM
GP2
1V p-p
50Ω
0.1µF
VIN–
AGND
1kΩ
Figure 30. Single-Ended Input Configuration
R
0.1µF 10Ω
1.2kΩ
AD9289
C
0.1µF
CLOCK INPUT AND CONSIDERATIONS
1kΩ
1kΩ
VIN+
AGND
1kΩ
AVDD
03682-054
0.1µF 10Ω
10µF
10kΩ
1kΩ
0.1µF
R
1kΩ
03682-052
AVDD
Figure 28. Differential Input Configuration Using the AD8351
However, the noise performance of most amplifiers is not
adequate to achieve the true performance of the AD9289. For
applications where SNR is a key parameter, differential transformer coupling is the recommended input configuration. An
example of this is shown in Figure 29.
Typical high speed ADCs use both clock edges to generate a
variety of internal timing signals, and as a result may be
sensitive to clock duty cycle. Typically, a 5% tolerance is
required on the clock duty cycle to maintain dynamic performance characteristics. The AD9289 has a self-contained clock
duty cycle stabilizer that retimes the nonsampling edge,
providing an internal clock signal with a nominal 50% duty
cycle. This allows a wide range of clock input duty cycles
without affecting the performance of the AD9289.
An on-board phase-locked loop (PLL) multiplies the input
clock rate for the purpose of shifting the serial data out. As a
result, any change to the sampling frequency requires a
minimum of 100 clock periods to allow the PLL to reacquire
and lock to the new rate.
In any configuration, the value of the shunt capacitor, C, is
dependent on the input frequency and may need to be reduced
or removed.
Rev. 0 | Page 15 of 32
AD9289
High speed, high resolution ADCs are sensitive to the quality of
the clock input. The degradation in SNR at a given full-scale
input frequency (fA) due only to aperture jitter (tA) can be
calculated with the following equation:
SNR degradation = 20 × log10 [1/2 × π × fA × tA]
In the equation, the rms aperture jitter, tA, represents the root
sum square of all jitter sources, which include the clock input,
analog input signal, and ADC aperture jitter specification.
Applications that require undersampling are particularly
sensitive to jitter.
The LVDS clock input should be treated as an analog signal in
cases where aperture jitter may affect the dynamic range of the
AD9289. Power supplies for clock drivers should be separated
from the ADC output driver supplies to avoid modulating the
clock signal with digital noise. Low jitter, crystal-controlled
oscillators make the best clock sources. If the clock is generated
from another type of source (by gating, dividing, or other
methods), it should be retimed by the original clock at the
last step.
The AD9289 can also support a single-ended CMOS clock.
Refer to the evaluation board schematics to enable this feature.
Power Dissipation and Standby Mode
As shown in Figure 31, the power dissipated by the AD9289 is
proportional to its sample rate. The digital power dissipation
does not vary because it is determined primarily by the strength
of the digital drivers and the load on each output bit.
Digital power consumption can be minimized by reducing the
capacitive load presented to the output drivers. The data in
Figure 31 was collected while a 5 pF load was placed on each
output driver.
The analog circuitry of the AD9289 is optimally biased to
achieve excellent performance while affording reduced
power consumption.
600
180
160
550
500
100
POWER
80
450
CURRENT (mA)
120
60
Digital Outputs
The AD9289’s differential outputs conform to the ANSI-644
LVDS standard. To set the LVDS bias current place a resistor
(RSET is nominally equal to 3.9 kΩ) to ground at the
LVDSBIAS pin. The RSET resistor current is derived on-chip
and sets the output current at each output equal to a nominal
3.5 mA. A 100 Ω differential termination resistor placed at the
LVDS receiver inputs results in a nominal ±350 mV swing at
the receiver. To adjust the differential signal swing, simply
change the resistor to a different value, as shown in Table 7.
Table 7. LVDSBIAS Pin Configuration
RSET
3.6k
3.9k (Default)
4.3k
IDRVDD
20
0
20
30
40
50
ENCODE (MSPS)
Differential Output Swing
375 mV p-p
350 mV p-p
325mV p-p
The AD9289’s LVDS outputs facilitate interfacing with LVDS
receivers in custom ASICs and FPGAs that have LVDS capability for superior switching performance in noisy environments. Single point-to-point net topologies are recommended
with a 100 Ω termination resistor placed as close to the receiver
as possible. It is recommended to keep the trace length no
longer than 12 inches and to keep differential output traces
close together and at equal lengths.
The format of the output data can be selected as offset binary or
twos complement. A quick example of each output coding
format can be found in Table 8. The DFS pin is used to set the
format (see Table 9).
Table 8. Digital Output Coding
40
400
60
70
Figure 31. Supply Current vs. fSAMPLE for fIN = 10.3 MHz
03682-032
POWER (mW)
In standby mode, low power dissipation is achieved by shutting
down the reference, reference buffer, and biasing networks. The
decoupling capacitors on REFT and REFB are discharged when
entering standby mode and then must be recharged when
returning to normal operation. As a result, the wake-up time is
related to the time spent in standby mode, and shorter standby
cycles result in proportionally shorter wake-up times. With the
recommended 0.1 µF and 10 µF decoupling capacitors on REFT
and REFB, it takes approximately 1 s to fully discharge the
reference buffer decoupling capacitors and 7 ms to restore full
operation.
140
IAVDD
350
10
By asserting the PDWN pin high, the AD9289 is placed in
standby mode. In this state, the ADC typically dissipates 7 mW.
During standby the LVDS output drivers are placed in a high
impedance state. Reasserting the PDWN pin low returns the
AD9289 into its normal operational mode.
Code
255
128
127
0
Rev. 0 | Page 16 of 32
VIN+ −
VIN− Input
Span = 2 V
p-p (V)
1.000
0
−0.00781
−1.00
VIN+ −
VIN− Input
Span = 1 V
p-p (V)
0.500
0
−0.00391
−0.5000
Digital
Output Offset
Binary
(D7...D0)
1111 1111
1000 0000
0111 1111
0000 0000
Digital
Output Twos
Complement
(D7...D0)
0111 1111
0000 0000
1111 1111
1000 0000
AD9289
Table 9. Data Format Configuration
Voltage Reference
DFS Mode
AVDD
AGND
A stable and accurate 0.5 V voltage reference is built into the
AD9289. The input range can be adjusted by varying the reference voltage applied to the AD9289, using either the internal
reference or an externally applied reference voltage. The input
span of the ADC tracks reference voltage changes linearly.
Data Format
Twos complement
Offset binary
Timing
Data from each ADC is serialized and provided on a separate
channel. The data rate for each serial stream is equal to eight
bits times the sample clock rate, with a maximum of 520 MHz
(8 bits x 65 MSPS = 520 MHz). The lowest typical conversion
rate is 12 MSPS.
Two output clocks are provided to assist in capturing data from
the AD9289. The DCO is used to clock the output data and is
equal to four times the sampling clock (CLK) rate. Data is
clocked out of the AD9289 and can be captured on the rising
and falling edges of the DCO that supports double-data rate
operation (DDR). The frame clock out (FCO) signals the start
of a new output byte and is equal to the sampling clock rate. See
the timing diagram shown in Figure 2 for more information.
The shared reference mode (see Figure 32) allows the user to
externally connect the reference buffers from the quad ADC for
better gain and offset matching performance. If the ADCs are to
function independently, the reference decoupling can be treated
independently and can provide better isolation between the four
channels. To enable shared reference mode, the SHARED_REF
pin must be tied high and external reference buffer decoupling
pins must be externally shorted. (REFT_A must be externally
shorted to REFT_B and REFB_A must be shorted to REFB_B.)
Note that Channels A and B are referenced to REFT_A and
REFB_A and Channels C and D are referenced to REFT_B
and REFB_B.
Table 10. Reference Settings
LOCK Pin
The AD9289 contains an internal PLL that is used to generate
the DCO. When the PLL is locked, the LOCK signal will be low,
indicating valid data on the outputs.
If for any reason the PLL loses lock, the LOCK signal goes high
as soon as the lock circuitry detects an unlocked condition.
While the PLL is unlocked, the data outputs and DCO remains
in the last known state. If the LOCK signal goes high in the
middle of a byte, no data or DCO signals will be available for
the rest of the byte. It takes at least 1.8 µs at 65 MSPS to regain
lock once it is lost. Note that regaining lock is sample ratedependent and takes at least 100 input periods after the PLL
acquires the input clock.
Once the PLL regains lock the DCO starts. The first valid data
byte is indicated by the FCO signal. The FCO rising edge occurs
0.5 to <1.5 input clock periods after LOCK goes low.
CML Pin
A common-mode level output is available at Pin F3. This output
self biases to AVDD/2. This is a relatively high impedance
output (2.5k nominal), which may need to be considered when
used as a reference.
Selected Mode
External
Reference
Internal,
1 V p-p FSR
Programmable
Internal,
2 V p-p FSR
Resulting
VREF (V)
N/A
VREF
0.5
0.2 V to
VREF
AGND to
0.2 V
0.5 ×
(1 + R2/R1)
1.0
Resulting
Differential Span
(V p-p)
2 × External
Reference
1.0
2 × VREF
2.0
Internal Reference Connection
A comparator within the AD9289 detects the potential at the
SENSE pin and configures the reference into four possible
states, which are summarized in Table 10. If SENSE is grounded,
the reference amplifier switch is connected to the internal resistor divider (see Figure 33), setting VREF to 1 V. Connecting the
SENSE pin to the VREF pin switches the amplifier output to the
SENSE pin, configuring the internal op amp circuit as a voltage
follower and providing a 0.5 V reference output. If an external
resistor divider is connected as shown in Figure 34 the switch is
again set to the SENSE pin. This puts the reference amplifier in
a noninverting mode with the VREF output defined as
R2 ⎞
VREF = 0.5 × ⎛⎜1 +
⎟
R1 ⎠
⎝
DTP Pin
When the digital test pattern (DTP) pin is enabled (pulled to
AVDD), all of the ADC channel outputs shift out the following
pattern: 11000000. The FCO and DCO outputs still work as
usual while all channels shift out the test pattern. This pattern
allows the user to perform timing alignment adjustments
between the DCO and the output data.
SENSE
Voltage
AVDD
In all reference configurations, REFT_A and REFT_B and
REFB_A and REFB_B establish their input span of the ADC
core. The input range of the ADC always equals twice the
voltage at the reference pin for either an internal or an external
reference.
Rev. 0 | Page 17 of 32
AD9289
VIN+A (+B)
VIN+A (+B)
VIN–A (–B)
VIN–A (–B)
REFT_A
A CORE
B CORE
0.1µF
+
VREF
0.1µF
A CORE
B CORE
10µF
REFB_A
0.1µF
10µF
0.1µF
R2
SELECT
LOGIC
10µF
0.1µF
VREF
0.1µF
+
REFB_A
VREF
0.1µF
VREF
10µF
0.1µF
REFT_A
SELECT
LOGIC
SENSE
SENSE
R1
0.5V
0.5V
VIN+C (+D)
VIN+C (+D)
VIN–C (–D)
VIN–C (–D)
REFT_B
VREF
C CORE
D CORE
0.1µF
0.1µF
+
0.1µF
C CORE
D CORE
10µF
0.1µF
03682-011
CONTROL
SHARED_REF
Figure 32. Shared Reference Mode Enabled
REFT_A
0.1µF
0.1µF
CONTROL
Figure 34. Programmable Reference Configuration
VIN+A (+B)
A CORE
B CORE
10µF
REFB_B
SHARED_REF
VIN–A (–B)
+
03682-013
REFB_B
AVDD
+
10µF
If the internal reference of the AD9289 is used to drive multiple
converters to improve gain matching, the loading of the reference by the other converters must be considered. Figure 35
depicts how the internal reference voltage is affected by loading.
REFB_A
VREF
0.05
0.1µF
VREF
0
0.1µF
–0.05
VREF ERROR (%)
SELECT
LOGIC
SENSE
0.5V
VREF = 0.5V
–0.10
–0.15
–0.20
VREF = 1.0V
–0.25
VIN+C (+D)
–0.30
VIN–C (–D)
–0.35
C CORE
D CORE
0.1µF
0.1µF
+
10µF
REFB_B
SHARED_REF
03682-033
REFT_B
VREF
CONTROL
03682-012
10µF
REFT_B
VREF
Figure 33. Internal Reference Configuration
Rev. 0 | Page 18 of 32
–0.40
0
0.5
1.0
1.5
ILOAD (mA)
Figure 35. VREF Accuracy vs. Load
2.0
2.5
AD9289
External Reference Operation
The use of an external reference may be necessary to enhance
the gain accuracy of the ADC or improve thermal drift
characteristics. Figure 36 shows the typical drift characteristics
of the internal shared reference in both 1 V and 0.5 V modes.
0.15
VREF = 0.5V
0.10
Power and Ground Recommendations
VREF = 1.0V
0
–0.05
–0.10
–0.15
–0.20
03682-034
VREF ERROR (%)
0.05
–0.25
–40
When the SENSE pin is tied to AVDD, the internal reference is
disabled, allowing the use of an external reference. An internal
reference buffer loads the external reference with an equivalent
7 kΩ load. The internal buffer still generates the positive and
negative full-scale references, REFT_A and REFT_B and
REFB_A and REFB_B , for the ADC core. The input span is
always twice the value of the reference voltage; therefore, the
external reference must be limited to a maximum of 1 V.
–20
0
20
40
TEMPERATURE (°C)
Figure 36. Typical VREF Drift
60
80
When connecting power to the AD9289, it is recommended that
two separate 3.0 V supplies be used. One for analog (AVDD)
and one for digital (DRVDD). If only one supply is available
then it should be routed to the AVDD first and tapped off and
isolated with a ferrite bead or filter choke with decoupling
capacitors proceeding. One may want to use several different
decoupling capacitors to cover both high and low frequencies.
These should be located close the point of entry at the pc board
level as well as close to the parts with minimal trace length.
A single pc board ground plane should be sufficient when using
the AD9289. With proper decoupling and smart partitioning of
the pc board’s analog, digital, and clock sections, optimum
performance is easily achieved.
Rev. 0 | Page 19 of 32
AD9289
EVALUATION BOARD
the signal sources that are used have very low phase noise
(< 1 ps rms jitter) to realize the ultimate performance of the
converter. Proper filtering of the analog input signal to
remove harmonics and lower the integrated or broadband
noise at the input is also necessary to achieve the specified
noise performance.
The AD9289 evaluation board provides all of the support circuitry required to operate the ADC in its various modes and
configurations. The converter can be driven differentially
through a transformer (default) or the AD8351 driver. Provisions have also been made to drive the ADC single-ended.
Separate power pins are provided to isolate the DUT from the
support circuitry. Each input configuration can be selected by
proper connection of various jumpers (refer to the schematics).
Figure 37 shows the typical bench characterization setup used
to evaluate the ac performance of the AD9289. It is critical that
XFMR
INPUT
–
+
DUT_DRVDD
GND
BRD_AVDD
3.3V
+
AD9289
EVALUATION BOARD
ROHDE AND SCHWARZ,
SMHU,
2V p-p SIGNAL
SYNTHESIZER
CLK
HSC-ADC-FPGA
HIGH-SPEED
DE-SERIALIZATION
BOARD
HSC-ADC-EVAL-DC
FIFO DATA
CAPTURE
BOARD
2 CH
8-BIT
PARALLEL
CMOS
USB
CONNECTION
CHA–CHD
8-BIT
SERIAL
LVDS
Figure 37. Evaluation Board Connections
Rev. 0 | Page 20 of 32
PC
RUNNING
ADC
ANALYZER
03682-002
BAND-PASS
FILTER
3.0V
–
GND
ROHDE AND SCHWARZ,
SMHU,
2V p-p SIGNAL
SYNTHESIZER
3.0V
+
DUT_AVDD
–
See Figure 37 to Figure 47 for complete schematics and layout
plots, which demonstrate the routing and grounding techniques
that should be applied at the system level.
03682-048
CLOCK
INPUT
NC
R41
50 Ω
2
V–
1
C7
0.1 µF
TP4
R109
1kΩ
R110
1kΩ
BRD_AVDD
R43
1k Ω
R40
1k Ω
1
4
3
2
1
RB
DNP
U1
13
U1
JP35
R70
1kΩ
R29
1kΩ
JP9
R73
100 Ω
5 V REF = 1V
TP2
JP1
R37
10k Ω
DUT_AVDD
TP1
TP3
DUT_AVDD
DUT_AVDD
VREF
SHARED_REF
H8
E6
F6
G6
E3
F5
G5
F4
G4
G7
F7
H5
H4
G3
F3
R76
3.9k Ω H1
6 V REF = 0.5V (1 + RA/RB)
4
8 V REF = 0.5V
7 V REF = EXTERNAL
3
2
1
U17
BRD_AVDD
R71
0Ω
R102
22Ω
5
6
7
JP5
R66
0Ω
SINGLE-ENDED
CLOCK DRIVER OPTION
2
12
DNC
DNC
DOUT+
8
C600
0.1 µF
JP3
C35
0.1 µF
C36
0.1 µF
VREF
BRD_AVDD
DOUT –
C80
0.1 µF
C100
10 µF
C44
0.1 µF
C120
10µF
FIN1017M
GND
DNC
DIN
VCC
U2
BRD_AVDD
C250
0.1 µF
CX
10 µF
RA
DNP
REMOVE CX
WHEN USING
EXTERNAL V REF U6
C47
0.1 µF
JP2
C1
0.1 µF
C53
0.1 µF
R1
1k Ω
BRD_AVDD
V+
C6
0.1 µF
ADR510
TRIM/NC
P5
3
U6
DTP
AVDD
AGND
AGND
AGND
REFT_B
REFB_B
REFT_A
REFB_A
AGND
AVDD
VREF
SENSE
SHARED_REF
CML
LVDSBIAS
AGND
E5
BRD_AVDD
F2
E4
VIN_B
H2
AD9289 EXTERNAL
REFERENCE OPTION
VIN_B
H6
AVDD
E2
SHARED_REF
DUT_AVDD
AD9289
VIN_A
R75
10k Ω
R72
10k Ω
FCO
C1
VOLTAGE REFERENCE CIRCUITRY
G2
AGND
DUT_AVDD
AVDD
H3
VIN –B
VIN –C
VIN_C
VIN+B
VIN+C
H7
VIN_C
AGND
E1
D7
AGND
E7
AGND
AGND
G1
G8
VIN_A
F1
F8
VIN+A
D1
DNC
DFS
C7
VIN –D
VIN_D
VIN+D
VIN_D
VIN –A
D2
DNC
CLK –
D8
FCO+
C2
E8
CLK+
C8
FCO
R32
10k Ω
FCO –
CHA
A8
PDWN
B1
D1+A
DNC
CHA
A1
D1 –A
B2
DRGND
DRVDD
DNC
DNC
D1 –C
D1+C
AGND
DCO –
DCO+
LOCK
DNC
DNC
D1 –B
D1+B
DRGND
DRVDD
D1 –D
A7
DNC
B8
DNC
A2
DNC
5
4
6
2
3
D3
C6
A6
B6
A5
B5
D5
C5
C4
D4
A4
B4
A3
B3
D6
C3
CHD
CHC
CHB
CHA
FCO
DCO
DUT_DRVDD
CHC
CHC
DCO
DCO
TP9
CHB
CHB
DUT_DRVDD
46
45
6
25
24
23
22
21
26
27
28
29
30
31
32
20
33
34
35
36
37
38
39
40
41
19
R4
DNP
R3
DNP
R2
DNP
R67
DNP
42
43
18
17
16
15
14
13
12
11
10
9
8
44
47
5
R25
DNP
48
4
7
49
3
R26
DNP
50
2
U5
1
CHD
CHC
CHB
CHA
FCO
DCO
** PLACE ONLY WHEN USING THE AD8351
AS INPUT. SEE DATASHEET FOR DETAILS.
DNP = DO NOT POPULATE
= GND
BRD_AVDD
1
J6
D1+D
Rev. 0 | Page 21 of 32
B7
Figure 38. Evaluation Board Schematic, DUT, VREF, and Clock Inputs
CHD
U7
CHD
DIGITAL LVDS OUTPUTS
AD9289
Figure 39. Evaluation Board Schematic, DUT Analog Input
03682-047
ANALOG
INPUT
CHANNEL D
ANALOG
INPUT
CHANNELC
P4
P3
ANALOG P2
INPUT
CHANNEL B
ANALOG P1
INPUT
CHANNEL A
JP20
JP26
JP18
JP21
JP24
JP25
JP27
Rev. 0 | Page 22 of 32
JP28
JP19
JP29
R42
50Ω
R21
1kΩ
CM4
C5
0.1µF
AVDD
R65**
DNP
AMP_IN4
R83
50Ω
R13
1kΩ
CM2
C3
0.1µF
AVDD
R64**
DNP
AMP_IN3
R30
50Ω
R63**
DNP
AMP_IN2
R15
50Ω
R62**
DNP
AMP_IN1
C20
0.1µF
4
3
R24
1kΩ
5
T4
6
2
1
C28
DNP
C15
0.1µF
4
3
R16
1kΩ
5
T2
6
2
1
C24
DNP
R84**
DNP
CM4
R82**
DNP
R74**
DNP
CM2
R69**
DNP
CH_D
CH_D
CH_B
CH_B
L13
120nH
L28
120nH
L27
120nH
L10
120nH
L9
120nH
L20
120nH
L19
120nH
L6
120nH
R11
33Ω
C4
0.1µF
AVDD
C9
DNP
R14
33Ω
R23
33Ω
C17
DNP
R27
33Ω
AVDD
CM1
VIN_D
C13
DNP
C10
20pF
R77**
DNP
CM3
R80**
DNP
R59**
DNP
CM1
R68**
DNP
CH_C
CH_C
CH_A
CH_A
L11
120nH
L22
120nH
L21
120nH
L12
120nH
L4
120nH
L1
120nH
L2
120nH
L3
120nH
R44
33Ω
C42
DNP
R47
33Ω
R34
33Ω
C31
DNP
R31
33Ω
VIN_C
C43
DNP
C41
20pF
VIN_C
VIN_A
C23
DNP
C30
20pF
VIN_A
** PLACE ONLY WHEN USING THE AD8351 AS INPUT. SEE
DATASHEET FOR DETAILS.
DNP = DO NOT POPULATE
= GND
C18
0.1µF
6
1
4
5
T3
C25
DNP
2
3
R19
1kΩ
VIN_D
R18
1kΩ
CM3
VIN_B
C22
DNP
C21
20pF
C16
0.1µF
6
1
4
5
T1
C19
DNP
2
3
R20
1kΩ
VIN_B
R12
1kΩ
C2
0.1µF
AD9289
Figure 40. Evaluation Board Schematic, Optional DUT Analog Input Drive
R8
10Ω
C64 R9
0.1µF 10Ω
R92
R79
25Ω
25Ω
C65
0.1µF
R90
10kΩ
C50 R7
0.1µF 10Ω
R49
R22
25Ω
25Ω
R5
10Ω
5
4
3
2
1
RGP2
INLO
INHI
RGP1
PWUP
U9
R78*
DNP
R50
1kΩ
R35
1.2kΩ
R93
1kΩ
R81
1.2kΩ
OPLO
03682-049
OPHI
VPOS
R94
1kΩ
6
7
8
9
10
COMM
OPLO
OPHI
VPOS
VOCM
6
7
8
9
10
C63
0.1µF
COMM
C62
0.1µF
R45
1kΩ
C49
0.1µF
VOCM
C48
0.1µF
BRD_AVDD
RGP2
INLO
INHI
RGP1
PWUP
U7
FOR PWDN
PLACE R78
5
4
3
2
1
R6*
DNP
FOR PWDN
PLACE R6
* TO ENABLE THE POWER DOWN OPTION ON THE
AD8351, PLACE A 0Ω RESISTOR IN THIS PLACE HOLDER.
R91
50Ω
AMP_IN3
R38
50Ω
AMP_IN1
C54
0.1µF
R36
10kΩ
BRD_AVDD
CH_A
CH_A
CH_C
CH_C
R106
25Ω
AMP_IN4
R99 C67
0Ω 0.1µF
R98 C66
0Ω 0.1µF
R55
25Ω
AMP_IN2
R48 C51
0Ω 0.1µF
R48 C52
0Ω 0.1µF
R10
10Ω
R33
10Ω
C70 R39
0.1µF 10Ω
R107
R101
50Ω
25Ω
C71
0.1µF
R104
10kΩ
C58 R17
0.1µF 10Ω
R56
R52
50Ω
25Ω
C59
0.1µF
R54
10kΩ
5
4
3
2
RGP2
INLO
INHI
RGP1
PWUP
U10
R100*
DNP
1
R57
1kΩ
R53
1.2kΩ
R58
1kΩ
R103
1.2kΩ
COMM
OPLO
OPHI
VPOS
6
7
8
9
10
C69
0.1µF
R108
1kΩ
VOCM
R111
1kΩ
6
7
8
9
C68
0.1µF
COMM
OPLO
OPHI
VPOS
10
C57
0.1µF
VOCM
C55
0.1µF
BRD_AVDD
RGP2
INLO
INHI
RGP1
PWUP
U8
FOR PWDN
PLACE R100
5
4
3
2
1
R51*
DNP
FOR PWDN
PLACE R51
BRD_AVDD
AD8351
AD8351
AD8351
Rev. 0 | Page 23 of 32
AD8351
AD8351 ANALOG INPUT DRIVER OPTION
R114 C73
0Ω 0.1µF
R113 C72
0Ω 0.1µF
R61 C61
0Ω 0.1µF
R60 C60
0Ω 0.1µF
CH_D
CH_D
CH_B
CH_B
AD9289
P6
P5
P4
P3
P2
P1
P6
Figure 41. Evaluation Board Schematic, Power, and Decoupling
03682-050
C14
0.1µF
DUT_AVDD
+3.0V
+3.0V
+3.3V
6
5
4
3
2
1
C12
0.1µF
C37
0.1µF
C171
10µF
L8
10µH
C176
10µF
L7
10µH
C170
10µF
C29
0.1µF
BRD_AVDD
DECOUPLING CAPS
+
+
+
C25
0.1µF
JP2
C11
0.1µF
11
9
5
3
10
8
6
4
GROUND TEST POINTS
U1
U1
U1
U1
** PLACE ONLY WHEN USING THE AD8351 AS INPUT. SEE
DATASHEET FOR DETAILS.
DNP = DO NOT POPULATE
= GND
C40
0.1µF
DUT_DRVDD
DUT_DRVDD
C110
0.1µF
DUT_AVDD
C183
0.1µF
BRD_AVDD
C108
0.1µF
TP14
JP1
JP4
UNUSED GATES
TP15
L5
10µH
TP16
Rev. 0 | Page 24 of 32
TP18
POWER CONNECTIONS
AD9289
03682-036
AD9289
03682-038
Figure 42. Evaluation Board Layout, Primary Side
Figure 43. Evaluation Board Layout, Primary Side (With Ground Copper Pour)
Rev. 0 | Page 25 of 32
03682-039
AD9289
03682-040
Figure 44. Evaluation Board Layout, Ground Plane
Figure 45. Evaluation Board Layout, Power Plane
Rev. 0 | Page 26 of 32
03682-045
AD9289
03682-041
Figure 46. Evaluation Board Layout, Secondary Side
Figure 47. Evaluation Board Layout, Secondary Side (With Ground Copper Pour)
Rev. 0 | Page 27 of 32
AD9289
Table 11. Evaluation Board Bill of Materials (BOM)
Item
1
Qnty.
per
Board
1
2
3
1
8
4
8
5
8
6
8
7
8
9
4
1
8
10
11
4
13
12
6
13
14
1
5
15
23
16
17
18
2
4
36
19
17
20
21
22
3
3
16
23
24
3
1
REFDES
AD9289 BGA
REVA/PCB
Assembly
R46, R48, R60, R61,
R98, R99, R113, R114
R5, R7, R8, R9, R10,
R17, R33, R39
R22,R49, R52, R55,
R79, R92, R101, R106
R11,R14, R23, R27,
R31, R34, R44, R47
R38, R56, R91, R107
R73
R45, R50, R57, R58,
R93, R94, R108, R111
R35, R53, R81, R103
R6, R32, R36, R51,
R54, R72, R75, R78,
R90, R100, R104, R37,
R76
R62, R63, R64, R65,
R66, R71
R102
R15, R30, R41, R42,
R83
R1, R12, R13, R16,
R18, R19, R20, R21,
R24, R29, R40, R43,
R59, R68, R69, R70,
R74, R77, R80, R82,
R84, R109, R110
R96, R97
C10, C21, C30, C41
C1, C35, C44, C47,
C80, C250, C600,
C11, C12, C14, C37,
C40, C48, C63, C64,
C65, C66, C67, C68,
C69, C70, C71, C72,
C73
C2, C3, C4,
C5, C6, C7, C15, C16,
C18, C20, C25, C29,
C36, C53, C108,
C110, C183
C100, C120, C163
C170, C171, C176
L1, L2 ,L3, L4, L6, L9,
L10, L11, L12, L13,
L19, L20, L21, L22,
L27, L28
L5,L7,L8
P6
Device
PCB
Package
PCB
Value
PCB
Manufacturing
PCSM
Mfg. Part Number
PCSM
RES_402
402
0
Protronics
Yageo America
Protronics
9C04021A0R00JLHF3
RES_402
402
10
Susumu Co Ltd
RR0510R-100-D
RES_402
402
25
Susumu Co Ltd
RR0510R-240-D
RES_402
402
33
Susumu Co Ltd
RR0510R-330-D
RES_402
RES_402
RES_402
402
402
402
50
100
1K
Panasonic-ECG
Yageo America
Panasonic-ECG
ERJ-L14KF50MU
9C04021A1000FLHF3
ERJ-2GEJ102X
RES_402
RES_402
402
402
1.2K
10K
Panasonic-ECG
Susumu Co Ltd
ERJ-2GEJ122X
RR0510P-103-D
BRES603
603
0
Panasonic-ECG
ERJ-3GEY0R00V
BRES603
BRES603
603
603
22
50
Susumu Co Ltd
Susumu Co Ltd
RR0816Q-220-D
RR0816Q-49R9-D-68R
BRES603
603
1K
Susumu Co Ltd
RR0816P-102-D
BRES603
CAP402
CAP402
603
402
402
XXX
20PF
1UF
Kemet
Panasonic-ECG
C0402C220J5GACTU
ECJ-0EF1C104Z
BYPASSCAP
603
0.1UF
Kemet
C0603C104Z3VACTU
TANTALUMB
TANTALUMB
INDUCTOR_6
805
T491B06K01
603
10UF
10UF
120NH
Panasonic-ECG
Kemet
Murata
ECJ-2FB0J106M
T491B106K016AS
BLM18BB750SN1D
IND1210
PTMICRO6
1210
PTMICRO6
10UH
6-Pole PCB
Header
Panasonic-ECG
Wieland
ELJ-SA100KF
Z5.531.3625.0
Rev. 0 | Page 28 of 32
AD9289
Item
Qnty.
per
Board
1
REFDES
Device
Package
25
5
P1, P2, P3, P4, P5
SMBMST
SMB
26
27
4
1
T1, T2, T3, T4
U17
ADT1-1WT
HEADER
CD542_X65
2MMSMT-872
28
29
1
1
U5
J6
DIFF_CONN
MINIJMPR3
FCN_268M01
2MMSMT-872
ADT1-1WT
WM18158ND
DIFF_CONN
MINIJMPR3
30
31
2
2
JP1, JP2
SGLJMPR
NYLON
87267-0850
1/4" 6-32
32
33
2
1
U1
SGLJMPR
11/4"
STANDOFF
6-32NUTS
74VHC04MTC
NYLON
TSSOP-14
6-32
74VHC04
34
1
U2
FIN1017M
MO8A_(SOIC)
FIN1017M
35
1
U4
9289BGA
36
37
1
4
U6
U7, U8, U9, U10
AD9289BBC65
ADR510
AD8351ARM
SOT23
MSOP010
Value
6-Pole PCB
Connector
SMBMST
Manufacturing
Wieland
Mfg. Part Number
25.600.5653.0
Amphenol-RF
Division
Minicircuits
Molex/Waldom
Electronics Corp
Fujitsu
Molex/Waldom
Electronics Corp
Samtec
RAF
901-144-8RFX
ADT1-1WT
87267-0850
FCN-268M012-G/1D
87267-0850
TSW-120-07-G-S
4040-632-N
9289BGA
RAF
Fairchild
Semiconductor
Fairchild
Semiconductor
ADI
AD9289BBC-65
ADR510
AD8351
ADI
ADI
ADR510
AD8351ARM
Rev. 0 | Page 29 of 32
3058-N
74VHC04MTC
FIN1017M
AD9289
OUTLINE DIMENSIONS
A1 CORNER
INDEX AREA
8.00
BSC SQ
8
7
6
5
4
3
2
1
A
BALL A1
INDICATOR
TOP VIEW
B
C
5.60
BSC SQ
D
0.80
BSC
E
F
G
H
BOTTOM VIEW
1.70
1.55
1.35
DETAIL A
DETAIL A
0.34 NOM
0.25 MIN
1.31
1.21
1.10
COPLANARITY
0.55
0.12
SEATING
0.50
PLANE
0.45
BALL DIAMETER
COMPLIANT TO JEDEC STANDARDS MO-205-BA
Figure 48. 64-Lead Chip Scale Package Ball Grid Array [CSP_BGA]
(BC-64-1)
Dimensions shown in millimeters
ORDERING GUIDE
Model
AD9289BBC
AD9289-65EB
Temperature Range
–40°C to +85°C
Package Description
64-Lead Chip Scale Package Ball Grid Array [CSP_BGA]
Evaluation Board
Rev. 0 | Page 30 of 32
Package Option
BC-64-1
AD9289
NOTES
Rev. 0 | Page 31 of 32
AD9289
NOTES
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D03682-0-10/04(0)
Rev. 0 | Page 32 of 32