IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 59, NO. 4, APRIL 2011
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Dual Grid Array Antennas in a Thin-Profile
Package for Flip-Chip Interconnection to
Highly Integrated 60-GHz Radios
Y. P. Zhang, Fellow, IEEE, M. Sun, Member, IEEE, Duixian Liu, Fellow, IEEE, and Yilong Lu, Member, IEEE
Abstract—We examine the current development of highly
integrated 60-GHz radios with an interest in antenna-circuit interfaces. We design and analyze grid array antennas with special
attention to the differential feeding and the patterned ground
plane. More importantly, we integrate two grid array antennas in
a package; propose the way of assembling it to the system printed
circuit board; and demonstrate a total solution of low cost and
thin profile to highly integrated 60-GHz radios. We show that the
package in low temperature cofired ceramic (LTCC) technology
measures only 13 2 13 2 0:575 mm3 ; can carry a 60-GHz radio die
of current and future sizes with flip-chip bonding; and achieves
good antenna performance in the 60-GHz band with maximum
gain of 13.5 and 14.5 dBi for the single-ended and differential
antennas, respectively.
Index Terms—Ball grid array package, grid array antenna, low
temperature cofired ceramic (LTCC), 60-GHz radio.
Fig. 1. Illustration of the (a-b) single-end and (c-d) differential antenna-circuit
interface in current highly integrated 60-GHz radios.
I. INTRODUCTION
RADITIONAL commercialized 60-GHz radios have
been designed as an assembly of several microwave
monolithic integrated circuits (MMICs) in gallium arsenide
(GaAs) semiconductor technology. They have been used for Gigabit Ethernet (1.25 Gb/s) bridges between local area networks
[1], [2]. Recently, integrated transmitter (Tx) and receiver (Rx)
GaAs pHEMT and mHEMT processes
MMICs in 0.15have been realized to support data rates of several Gb/s for
60-GHz short-range applications [3], [4]. However, the 60-GHz
radios in GaAs MMICs are expensive and bulky. In order for
60-GHz radios to have mass deployment and meet consumer
marketplace requirements, the cost and size of any solution
must be low and compact. That implies silicon, not GaAs as
the better technology choice. In fact, designs towards low-cost
highly integrated 60-GHz radios have been realized in silicon
technologies. For example, Floyd, et al. have demonstrated a
T
Manuscript received March 26, 2010; revised August 05, 2010; accepted August 30, 2010. Date of publication January 28, 2011; date of current version
April 06, 2011.
Y. P. Zhang and Y. L. Lu are with the School of Electrical and Electronic
Engineering, Nanyang Technological University, Singapore 639798, Singapore
(e-mail: eypzhang@ntu.edu.sg; eylu@ntu.edu.sg).
M. Sun is with the Institute for Infocomm Research, Singapore 138623
(e-mail: msun@i2r.a-star.edu.sg).
D. Liu is with the IBM T. J. Watson Research Center, Yorktown Heights, NY
10598 USA (e-mail: duixian@us.ibm.com).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TAP.2011.2109358
60-GHz Tx and Rx chipset in a 0.13silicon-germanium
(SiGe) technology [5] and Tanomura, et al. in a 90-nm complementary metal oxide semiconductor (CMOS) technology [6].
An examination of the above works and many other reported
60-GHz highly integrated radios in SiGe and CMOS reveals
that two types of antenna-circuit interfaces as shown in Fig. 1
can be identified in the current two-chip solutions. The first
type features the 50- single-end and the second type the
100- differential antenna-circuit interfaces. For the first type,
the 60-GHz on-chip input/output pads are designed as the
ground-signal-ground (GSG) pads; while for the second type as
the ground-signal-ground-signal-ground (GSGSG) pads. The
GSG pads are bonded to an off-chip but in-package single-end
antenna; while the GSGSG pads a differential antenna with
either flip-chip or wire-bonding techniques [7]–[10].
A single-chip solution of a 60-GHz radio transceiver (TRX)
in CMOS has been attempted [11], where differential Tx and Rx
are integrated on the same die. It is known that CMOS scaling
improves amplifier noise performance and gain but exacerbates
the difficulty of generating sufficient output power by the power
amplifier (PA) at 60 GHz [12]. Theoretically, a differential PA
yields 3 dB more output power than a single-end one does.
Hence, the differential antenna-circuit interface in Fig. 1(c) is
preferred to the single-end antenna-circuit interface in Fig. 1(a)
for the Tx integration of the TRX. Furthermore, the differential
antenna-circuit interface in Fig. 1(d) is the better choice than the
single-end antenna-circuit interface in Fig. 1(b) for the Rx integration of the TRX because the differential low noise amplifier
0018-926X/$26.00 © 2011 IEEE
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IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 59, NO. 4, APRIL 2011
(LNA) can achieve higher linearity, lower offset and better immunity to common-mode noise due to power supply variations
or substrate coupling than the single-end LNA does [13]. The
advantage of the fully differential architecture from the antenna
to the circuit has been well understood for modern single-chip
solutions of radio transceivers [14].
Regardless of the antenna-circuit interfaces; two antennas,
one for transmission and the other for reception, are required for
current highly integrated 60-GHz radios. Use of two antennas
is not a problem as the antenna form factor at 60 GHz is on the
order of millimeters. However, it may become a problem for the
highly integrated 60-GHz radio that employs multiple antennas
for beam steering to search the available path to enhance the link
quality [15]. This is because multiple antennas not only require
corresponding multiple electrostatic discharge circuits that consume substantial die area but also makes the whole radio bulky.
It is known that the number of multiple antennas can be reduced
to half by using transmit/receive (T/R) switches or circulators.
Unfortunately, the T/R switches in CMOS in the 60-GHz band
are still too lossy to be used [11], [16] and information on circulators for highly integrated 60-GHz radios is unknown.
The remainder of the paper is organized as follows: Section II
presents the design and analysis of grid array antennas with
special attention to the differential feeding and the patterned
ground plane. Section III describes the integration of the grid
array antennas into a chip package in LTCC for highly integrated 60-GHz radio chipsets. As an example, a dual-feed grid
array antenna is integrated for the differential Tx antenna-circuit interface and another single-feed grid array antenna for the
single-end Rx antenna-circuit interface. Finally, Section IV concludes the paper.
II. DESIGN AND ANALYSIS OF GRID ARRAY ANTENNAS
The grid array antenna was first proposed by Kraus in 1964
[17]. Since then, there have been some studies but all conducted
at lower microwave frequencies [18]–[23]. Fig. 2 shows the
basic grid arrangement and its variations in microstrip technology. The basic structure shown in Fig. 2(a) consists of rectangular meshes of microstrip lines on a dielectric substrate backed
by a metallic ground plane and fed by a metal via through an
aperture on the ground plane. Depending on the electrical length
of the sides of the meshes, the grid array antenna may be resonant or nonresonant. For a resonant grid array antenna, the sides
of the meshes should be one wavelength by a half-wavelength
in the dielectric and the instantaneous currents would be out of
phase on the long sides of the meshes and in phase on the short
sides of the meshes, respectively. As a result, the long sides of
the meshes behave essentially as microstrip line elements and
the short sides act as both radiating and microstrip line elements
producing the main lobe of radiation in the boresight direction.
While for a nonresonant grid array antenna, the length of the
short side of the meshes can be slightly more than one-third
wavelength and the length of the long side of the meshes should
be two times longer but three times shorter than the length of
the short side of the meshes in the dielectric. Assuming that it is
fed from one end, the currents in the short sides of the meshes
Fig. 2. The basic grid array antenna (a) and its variations (b-g).
follow a phase progression producing the maximum radiation in
a backward angle-fire direction [17].
The grid array antenna was temporarily revived by Conti,
Dowling, and Weiss in 1981 [18]. Fig. 2(b) shows their methods
of amplitude control through control of microstrip line impedances to lower the first sidelobe. The grid array antenna has
caught considerable attention of Nakano and his associates
[20]–[23]. Since the middle of 1990s, they have reported the
design and analysis of various grid array antennas. Fig. 2(c)–(e)
show their miniaturized grid array antenna by meandering the
long sides of the meshes, dual-linearly polarized grid array
antenna by crossing the meshes, circularly polarized grid array
antenna by modifying the short sides of the meshes, respectively. Fig. 2(f) and (g) shows our new 45 linearly polarized
grid array antenna by adjusting the angle between the long and
short sides of the meshes and miniaturized grid array antenna
by meandering the long sides and bending the short sides of the
meshes in a multi-layer metal structure, respectively, [24]. The
bending makes the large part of the short sides of the meshes
further away from the ground plane, which may improve the
radiation.
Although both resonant and nonresonant grid array antennas
are useful for many applications, we only focus on the design
and analysis of the resonant grid array antenna in this work at
millimeter-wave frequencies. The design determines the dielectric substrate dimensions, the number of meshes, the microstrip
ZHANG et al.: DUAL GRID ARRAY ANTENNAS IN A THIN-PROFILE PACKAGE FOR FLIP-CHIP INTERCONNECTION
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Fig. 3. The dual-feed grid array antenna: (a) top and (b) bottom views.
line impedances, and the excitation location with the associated
diameters of the metal via and the aperture through the analysis
of the HFSS simulations. We demonstrate that the grid array antenna is easy to design, more importantly; the grid array antenna
is particularly suitable for fabrication in LTCC as its meshed
structure avoids micro fractures or warpage. Based on the A6
LTCC from Ferro, we design grid array antennas to operate in
the 60-GHz band. The A6 ceramic type has the dielectric constant 5.9 and loss tangent 0.0015 and after firing and metallic
paste is either silver or gold with good conductivity.
A. Basic Dual-Feed Structure
A basic dual-feed grid array antenna is designed, which targets the specifications at 61.5 GHz with the maximum gain of
15 dBi, the impedance and radiation bandwidth of 7 GHz, and
the efficiency of 80%. Considering the specified gain value of
15 dBi and the various losses, one can find the required number
of meshes to be at least 14, which leads to an estimation of the
length and width of the substrate as 11.5 mm by 5 mm, respectively. The thickness of the substrate should be chosen to avoid
mode surface wave. Finally, a body
the excitation of the
is determined by also taking the
size of
LTCC layout rule into account. For low cost and easy fabrication, the width and thickness of the microstrip lines are kept uniform as 0.15 and 0.01 mm, respectively. The optimized mesh dimensions and the location of dual feeds as well as the associated
diameters of the metal vias and the apertures on the ground plane
,
,
,
are
,
,
,
,
.
and
The grid array antenna has dual feeds as shown in Fig. 3. It is
excited for differential and single-end operations, respectively.
The differential feeding scheme here is different from those in
[17], [23] where a gap is made on the short side of a mesh
to connect to the differential source. Fig. 4(a) and (b) shows
the simulated current distributions on the grid at 61.5 GHz for
both excitations. Note that the instantaneous currents do not distribute as shown in [17]–[19], that is, they are out of phase on
the long sides of the meshes and in phase on the short sides of
the meshes, respectively. Rather, they are only truly out of phase
on the long sides of the meshes and in phase on the short sides
of the meshes near the feeding points. This is because at such
a high frequency a slight mesh dimension change will cause a
big change in signal phase over transmission, for example at 60
variation in will cause
signal phase difGHz a 70ference, thus making the control over phase synchronization of
Fig. 4. The simulated results of the dual-feed grid array antenna: current distributions at 61.5 GHz for (a) differential and (b) single-end operations, (c) return
loss, (d) E or xz - and (e) H or yz -plane patterns at 61.5 GHz, and (f) peak realized gain for both single-end and differential operations.
the far meshes from the source more difficult. The grid array antenna excited for differential operation has two source points, so
it has more meshes of desirable current distributions than that
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Fig. 5. The ground planes: (a) meshed and (b) patterned.
excited for single-end operation. Fig. 4(c) compares the simulated antenna return losses. The impedance bandwidth is 10
GHz (16.7%) from 53 to 63 GHz for the single-end and 8 GHz
(13%) from 57 to 65 GHz for the differential operations, respectively. The wider impedance bandwidth for the single-ended antenna is the result of less number of radiating elements that have
in-phase currents. Fig. 4(d) and (e) compares the simulated antenna radiation patterns in the E-and H-planes at 61.5 GHz. As
expected, the main lobe and deep null of radiation appear in
the boresight direction for the co-and cross-polarization fields,
respectively. The differential excitation yields a sharper main
beam in the E-plane and a similar main beam in the H-plane
and much weaker cross-polarization field in both planes as compared with the single-end operation, due to the current distributions shown in Fig. 4(a) and (b). The front-to-back ratio is
21 dB. Fig. 4(f) shows the simulated antenna peak gain. They
are 16 and 13.5 dBi at 61.5 GHz for differential and single-end
operations, respectively. The 3-dB gain bandwidths are enough.
The simulated efficiency is better than 90% for both cases.
B. Patterned Ground Plane
The meshed ground plane is required in LTCC from the mechanical perspective. A patterned ground plane is created in this
work which not only meets the requirement of mechanical reliability but also reduces the gain penalty. Fig. 5 shows the patterned ground plane of the grid array antenna. Note that the large
metal patches are formed on the meshed ground plane below the
radiating elements, which reduce radiation through the ground
plane meshes and therefore help to reduce the gain penalty.
Fig. 6 compares the simulated results of the grid array antenna
between the conventional meshed and patterned ground planes
for the differential operation. It is evident that both meshed and
patterned ground planes shift down the resonant frequencies.
They perturb the current return path and lead to the excitation of
an electric field across the rectangular openings, and the reactive energy stored near the discontinuities is responsible for the
downward frequency shift. Also, it can be observed that both
meshed and patterned ground planes enhance the impedance
bandwidth but degrade the radiation characteristics such as gain,
side-lobe level, and front-to-back ratio. Nevertheless, as confirmed in Fig. 6(d), the patterned ground plane reduces the gain
penalty by 0.7 dBi at 60 GHz compared with the case of the
meshed ground plane.
C. Dual Grid Array Antennas
As previously discussed, current highly integrated 60-GHz
radios require dual antennas. Nakano et al. designed dual grid
Fig. 6. The simulated results of the dual-feed grid array antenna for differential
operation with the meshed and patterned ground planes: (a) return loss, (b) E or
xz - and (c) H or yz -plane patterns at 61.5 GHz, and (d) peak realized gain.
array antennas in a double-layer structure [20]. A perpendicular orientation was arranged for the upper and lower grid array
antennas. In this way, the upper grid array antenna radiates a
horizontally polarized wave; while the lower grid array antenna
does a vertically polarized one. A high isolation between both
centrally located feeding terminals can be guaranteed. Fig. 7
shows our dual grid array antennas in a single-layer structure.
It is formed by two grid array antennas of the basic structure
studied in earlier in this section and has a body size of
. A parallel orientation is arranged for the dual
grid array antennas. Both grid array antennas radiate the wave of
the same polarization. The single-layer structure simplifies the
fabrication process. The parallel orientation reduces the outrage
probability of 60-GHz radio links, which are usually deployed
in line-of-sight environments, due to the polarization loss.
Fig. 8 compares the simulated results of the grid array antenna with the dual feeds for differential operation and the grid
array antenna with a single feed for single-end operation. As
this is a transitional step in our development, no performance
enhancement is made and no patterned ground plane is used.
When one grid array antenna is excited, the other grid array antenna acts as a parasitic element, and vice versa. Fig. 8(a) shows
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Fig. 7. The dual grid array antennas: (a) top and (b) bottom views.
that the impedance bandwidth is 5.3 GHz (8.6%) from 59 to
64.3 GHz for the differential antenna and 9.7 GHz (15.8%) from
52.5 to 62.2 GHz for the single-end antenna. With reference to
Fig. 4(d)–(f), one can find from Fig. 8(b)–(d) that the radiation
patterns and the maximum gain of the single-end antenna are
improved more noticeably than those of the differential antenna
by the parasitic element. For example, the beamwidth in the
E-plane is narrowed and the radiation of cross-polarization in
the H-plane is suppressed, so the gain is improved to 15 dBi for
the single-end antenna. Fig. 8(e) shows the simulated isolation
level between the dual grid array antennas. For the sake of simulation simplicity, both grid array antennas are fed for single-end
operation. Note the isolation is high because of the large physical separation between the two feeding terminals.
III. INTEGRATION OF ARRAY ANTENNAS IN PACKAGE
The integration of the grid array antenna in a wirebond
package has been realized [25], [26]. In this Section, the integration of dual grid array antennas in a flip-chip package
is described. Fig. 9 illustrates the integration. Note that the
package features standard flip-chip bonding and there are three
cofired laminated ceramic layers for the package. The 1st ceramic layer is 0.385 mm thick, the second to the third ceramic
layers are both 0.095 mm thick. There are four metallic layers
for the package. The top layer provides the metallization for the
dual grid array antennas, the 1st buried layer metallization for
the patterned ground plane, the second buried layer the metallization for the antenna feeding traces, and the bottom exposed
layer the metallization for the signal traces. The package has
48 input/outputs with a JEDEC standard pitch of 0.25 mm. The
.
size of the whole package is
Fig. 9 also illustrates the zoom-in view of the feeding networks of the dual grid array antennas. For the dual-feed one,
it consists of such packaging elements as two quasi-coaxial cables cascaded first with two striplines, then another two quasicoaxial cables, and finally vias through two apertures on the
ground plane in a GSGSG arrangement. It is interesting to note
that the differential feeding ports in Fig. 9 are brought closer to
each other as compared with those in Fig. 3, due to the requirement of flip-chip bonding to the on-die GSGSG pads. The radiating element between the differential feeding ports is removed
to enhance their isolation. For the single-feed one, it consists
of a quasi-coaxial cable cascaded with via through one aperture
on the ground plane in a GSG arrangement. It is known that
Fig. 8. The simulated results of the dual grid array antennas with one for differential and the other for single-end operations: (a) return loss, (b) E or xz - and
(c) H or yz -plane patterns at 61.5 GHz, (d) peak realized gain, and (e) isolation.
the GSG and GSGSG arrangements not only minimize potential
electromagnetic interference but also improve the feeding performance. The GSG and GSGSG feeding networks are designed
together with the grid array antennas. Both GSG and GSGSG
pads have a pitch of 0.25 mm.
Fig. 10 shows the bottom view of the package without the
signal traces but with the integrated balun for testing the differential grid array antenna with the single-ended equipment.
The package with the dual microstrip grid array antennas
were fabricated in FERRO A6 LTCC in Singapore Institute of
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Fig. 10. Illustration of the integrated balun for testing the differential grid array
antenna with the single-ended equipment.
Fig. 11. Photos of the dual grid array antennas in the package on the fixture for
testing: (a) top view. (b) Bottom view with signal traces. (c) Bottom view with
the integrated balun.
Fig. 9. Illustration of integration of dual grid array antennas in a package:
(a) top view. (b) Explored view. (c) Bottom view, as well as the zoom-in view
of the feeding networks of the dual grid array antennas.
Manufacturing Technology. Fig. 11 illustrates the test fixture to
hold the package for testing.
Fig. 12 compares the simulated and measured results of the
single-end antenna for sample A. The return losses agree very
well from 56 to 58.5 GHz. The agreement becomes poor for
higher frequencies, due to the following reasons: dimension tolerance control, material property variation, and the difference
between the wave-port excitation in simulation and the probe excitation in measurement. The measured return loss is higher than
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Fig. 12. The simulated and measured results of the single-end antenna in the
package: (a) return loss. (b) E or xz - and (c) H or yz -plane patterns at 61.5 GHz.
(d) Peak realized gain.
Fig. 13. The simulated and measured results of the differential antenna in the
package: (a) return loss. (b) E or xz - and (c) H or yz -plane patterns at 61.5 GHz.
(d) Peak realized gain.
10 dB from 56.4 to 61.7 GHz and 8 dB from 55 to 63.4 GHz indicating acceptable matching to 50- sources at these frequencies. The simulated and measured radiation patterns are in close
agreement at 61.5 GHz. The measured maximum gain is 13.5
dBi at 59 GHz with 3-dB gain bandwidth of 4.5 GHz. The simulated radiation efficiency is better than 85%.
Fig. 13 compares the simulated and measured results of the
differential antenna for sample B. The discrepancies between
the simulated and measured return losses are due to the same
reasons explained above for single-end antenna. The measured
return loss is higher than 8 dB from 56.2 to 63.2 GHz indicating acceptable matching to 50- sources at these frequencies. Again, the simulated and measured radiation patterns at
61.5 GHz are in close agreement. The larger side lobes in the
back side are caused by the balun. The measured maximum gain
is 13.5 dBi at 57.5 GHz with 3-dB gain bandwidth of 5.3 GHz.
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Fig. 14. The concept of assembling the highly integrated 60-GHz radio in the
package to the system PCB.
If the additional loss of 1 dB from the integrated balun is de-embedded, the maximum gain of the differential antenna becomes
14.5 dBi. The simulated radiation efficiency is also better than
85%.
Generally, the measured results confirm the simulated ones.
For instance, the differential grid array antenna exhibits a
sharper radiation beam in the E-plane, a similar radiation pattern in the H-plane, and a higher gain as compared with those
of the single-end grid array antenna.
Fig. 14 illustrates our concept of assembling the highly integrated 60-GHz radio in the chip-scale package to the system
printed-circuit board (PCB). A cavity or even an opening needs
to be created in the PCB to house and protect the radio die. The
lands on the chip package are soldered to the PCB to finish interconnect from the chip to the PCB through the package. This is
believed to be a very cheap solution to highly integrated 60-GHz
radios with a rather thin profile.
IV. CONCLUSION
The examination of the reported highly integrated 60-GHz
radios was made. It was found that both 50- single-end
and 100- differential antenna-circuit interfaces were used
in current two-and single-chip solutions. A dual-feed grid
array antenna was designed in LTCC and analyzed for both
single-end and differential feedings. It was shown that the
differential feeding resulted in higher gain but narrower
impedance bandwidth and the patterned ground plane reduced
the gain penalty. Dual grid array antennas in a single-layer
structure were also designed and integrated in a package with
one antenna for single-end driving and the other for differential
feeding. The results showed that the package achieved good
antenna performance in the 60-GHz band with maximum gain
of 13.5 and 14.5 dBi for the single-ended and differential
antennas, respectively. Furthermore, the novel concept of
assembling a highly integrated 60-GHz radio in the package to
the system PCB was disclosed, which was believed to be a very
cheap system solution to highly integrated 60-GHz radios with
a rather thin profile.
ACKNOWLEDGMENT
The authors would like to thank K. M. Chua and L. L. Wai
of Singapore Institute of Manufacturing Technology for their
support in fabrication of the package in LTCC.
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ZHANG et al.: DUAL GRID ARRAY ANTENNAS IN A THIN-PROFILE PACKAGE FOR FLIP-CHIP INTERCONNECTION
Y. P. Zhang (M’03–SM’07–F’10) received the B.E.
and M.E. degrees from Taiyuan Polytechnic Institute
and Shanxi Mining Institute of Taiyuan University
of Technology, Shanxi, China, in 1982 and 1987, respectively, and the Ph.D. degree from the Chinese
University of Hong Kong, Hong Kong, in 1995, all
in electronic engineering.
From 1982 to 1984, he was with the Shanxi Electronic Industry Bureau; from 1990 to 1992, the University of Liverpool, Liverpool, U. K.; and from 1996
to 1997, City University of Hong Kong. From 1987
to 1990, he taught at the Shanxi Mining Institute and from 1997 to 1998, the
University of Hong Kong. He was promoted to a Full Professor at Taiyuan University of Technology in 1996. He is now an Associate Professor and the Deputy
Supervisor of Integrated Circuits and Systems Laboratories with the School of
Electrical and Electronic Engineering, Nanyang Technological University, Singapore. He has broad interests in radio science and technology and published
widely across seven IEEE societies.
Dr. Zhang received the Sino-British Technical Collaboration Award in 1990
for his contribution to the advancement of subsurface radio science and technology. He received the Best Paper Award from the Second International Symposium on Communication Systems, Networks and Digital Signal Processing,
July 18–20, 2000, Bournemouth, U.K., and the Best Paper Prize from the Third
IEEE International Workshop on Antenna Technology, March 21–23, 2007,
Cambridge, U.K. He has organized/chaired dozens of technical sessions of international symposia. He was awarded a William Mong Visiting Fellowship from
the University of Hong Kong in 2005. He has delivered scores of invited papers/keynote addresses at international scientific conferences. He was a Guest
Editor of the International Journal of RF and Microwave Computer-Aided Engineering and an Associate Editor of the International Journal of Microwave
Science and Technology. He serves as an Editor of ETRI Journal, an Associate
Editor of the IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, and an
Associate Editor of the International Journal of Electromagnetic Waves and Applications. He also serves on the Editorial Boards of a large number of Journals
including the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES
and IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS.
M. Sun (M’09) received the B.E. and M.E. degrees
from the Hunan University and Beijing Institute of
Technology, China, in 2000 and 2003, respectively,
and the Ph.D. degree from the Nanyang Technological University (NTU), Singapore, in 2007, all in electronic engineering.
She became a Research Associate with NTU
in 2006 and subsequently converted to Research
Fellowship in 2007. In 2009, she joined the Institute for Infocomm Research, Singapore, as a
Research Fellow. Her research interests include
millimeter-wave and Terahertz antenna design.
Dr. Sun was a recipient of the Best Paper Prize from the Third IEEE International Workshop on Antenna Technology, March 21–23, 2007, Cambridge,
U.K.
1199
Duixian Liu (S’85–M’90–SM’98–F’10) received
the B.S. degree in electrical engineering from Xidian
University, Xi’an, China, in 1982, and the M.S. and
Ph.D. degrees in electrical engineering from The
Ohio State University, Columbus, in 1986 and 1990,
respectively.
From 1990 to 1996, he was with Valor Enterprises,
Inc., Piqua, OH, initially as an Electrical Engineer
and then as the Chief Engineer, during which time
he designed an antenna product line ranging from 3
MHz to 2.4 GHz for the company, a very important
factor for the prestigious Presidential “E” Award for Excellence in Exporting in
1994. Since April 1996, he has been with the IBM T. J. Watson Research Center,
Yorktown Heights, NY, as a Research Staff Member. He has 28 patents issued
and 15 patents pending. His research interests are antenna design, EM modeling, digital signal processing, and communications technology. He has served
as external Ph.D. examiner for several universities and external examiner for
some government organizations on research grants. His research interests are
antenna design, chip packaging, electromagnetic modeling, digital signal processing, and communications technology. He has authored or coauthored approximately 60 journal and conference papers.
Dr. Liu is an Associate Editor for the IEEE TRANSACTIONS ON ANTENNAS
AND PROPAGATION and was a Guest Editor for the IEEE TRANSACTIONS ON
ANTENNAS AND PROPAGATION Special Issue on “Antennas and Propagation Aspects of 60–90 GHz Wireless Communications” (July 2009). He has received
three IBM’s Outstanding Technical Achievement Awards and one Corporate
Award, the IBM’s highest technical award. He was named Master Inventor in
2007. He has been the Organizer or Chair for numerous international conference
sessions or special sessions and also a Technical Program Committee member.
He was the General Chair of the 2006 IEEE International Workshop on Antenna
Technology: Small Antennas and Novel Metamaterials, White Plains, NY.
Yilong Lu (S’90–M’92) received the B.Eng. degree
from Harbin Institute of Technology, China, in January 1982, the M. Eng. degree from Tsinghua University, China, in November 1984, and the Ph.D. degree
from University College London, U.K., in November
1991, all in electronic engineering.
From November 1984 to September 1988, he was
with the Department of Electromagnetic Fields Engineering, University of Electronic Science and Technology of China, Chengdu, China, as a lecturer in the
Antenna Division. In December 1991, he joined the
School of Electrical and Electronic Engineering, Nanyang Technological University (NTU), where he is currently a full Professor in the Communication
Engineering Division and is also the leader of Radar Research Group, the Coordinator of the Microwave Circuits, Antennas and Propagation Research Group,
and Deputy Director of Centre for Modeling and Control of Complex Systems,
in NTU. He was a Visiting Academic with the University of California—Los
Angeles from October 1998 to June 1999. His research interests include antennas, array based signal processing, radar systems, computational electromagnetics, and evolutionary computation for optimization of complex problems.
He is a member of Editorial Board for IET Radar, Sonar and Navigation.