IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
1423
Effect of Adjustable-Speed Drives on the Operation
of Low-Voltage Ground-Fault Indicators
Gary L. Skibinski, Barry M. Wood, Senior Member, IEEE, John J. Nichols, Member, IEEE, and
Louis A. Barrios, Senior Member, IEEE
Abstract—Over the years, the petroleum and chemical industry
has found increasing favor with 60-Hz low-voltage ( 600 Vac)
power systems that utilize a high-resistance grounded (HRG)
neutral philosophy. Historically, the older generation of adjustable-speed drives (ASDs) had little or no effect on the normal
operation of ground-fault indicators (GFIs) used with the installed
HRG systems. This paper focuses its investigation into nuisance
GFI alarms that may occur when present-generation ASDs are
retrofitted into the existing plant. The paper first reviews possible
neutral grounding systems, with emphasis on the types of HRG
systems possible and GFI alarm philosophy. The paper then discusses how ASDs may generate zero-sequence high-frequency noise
currents in the HRG neutral circuit, which may cause nuisance
ground-fault alarms and potentially mask a legitimate ground
fault. GFI noise current magnitude is defined for both present
and older ASD technologies (e.g., insulated gate bipolar transistor
versus bipolar junction transistor drives). The effect this transient
zero-sequence noise current magnitude has on GFI operation is
described. Mitigation methods used at the drive to reduce ASD noise
current magnitude to acceptable nonalarm levels is investigated.
Filter solutions located at the HRG/GFI meter that reduce nuisance
alarms are also investigated. The pros and cons of at the drive or at
the meter filter solutions are supported with laboratory and field
test data. Application guidelines are given to help avoid nuisance
problems with a plant ground-fault protection scheme, which
needs to successfully operate in the presence of multiple ASDs.
Index Terms—Adjustable-speed drives, common-mode noise,
electromagnetic interference filters, ground-fault indicators, highresistance neutral ground, pulsewidth modulation, zero-sequence
current.
I. INTRODUCTION
A. System Grounding Philosophy
S
YSTEM grounding philosophy for process industry applications is based on providing safe and reliable power
distribution, while insuring maximum protection against
Paper PID 00–2, presented at the 1999 IEEE Petroleum and Chemical Industry Technical Conference, San Diego, CA, September 13–15, and approved
for publication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS by the
Petroleum and Chemical Industry Committee of the IEEE Industry Applications
Society. Manuscript submitted for review September 15, 1999 and released for
publication June 5, 2001.
G. L. Skibinski is with Standard Drives, Rockwell Automation–AllenBradley Company, Inc., Mequon, WI 53092 USA (e-mail: glskibinski@ra.rockwell.com).
B. M. Wood is with Chevron Research and Technology Company, Richmond,
CA 94802 USA (e-mail: bamw@chevron.com).
J. J. Nichols is with the Wood River Refinery, Tosco Refining Company,
Roxana, IL 62084 USA (e-mail: jnichols@tosco.com).
L. A. Barrios is with Equilon Technology, Houston, TX 77082-3101 USA
(e-mail: labarrios@equilontech.com).
Publisher Item Identifier S 0093-9994(01)08316-5.
Fig. 1.
Typical neutral grounding systems.
transient voltages and maintaining uptime availability during
ground faults [1].
An advantage of the ungrounded system of Fig. 1 is that a
single line-to-ground fault does not require immediate interruption of power flow. A disadvantage is that primary line-toground voltage transients are passed to the secondary without attenuation. Impulse testing of a transformer with an ungrounded
neutral shows a 2000-V-peak 1.2 s/50 s impulse test applied
line to ground on the transformer primary is virtually unattenuated line to ground on the secondary side. Another disadvanneutral point is capacitively coupled to ground,
tage is that
voltage to float toward the line voltage during tranallowing
sients and overstress the line-to-neutral insulation system. The
principal problem with an ungrounded neutral system is that an
arcing ground fault can cause escalation of the system line-toground voltages to several times normal line-to-ground voltage.
This occurs as the arcing fault alternatively extinguishes and
reestablishes itself, successively trapping a higher charge on the
system shunt capacitance each time. This causes displacement
of the system neutral, which is only connected to ground through
the shunt capacitance, to a voltage that is several times normal
line-to-ground voltage. Voltages to ground which are sufficient
to cause insulation failures, especially in motors, are very possible [2].
An advantage of grounded wye systems in Fig. 1 is a
5 : 1– 10 : 1 attenuation of transformer primary line-to-ground
voltage transients. Impulse testing of a transformer with a
grounded neutral shows a 2000-V-peak 1.2 s/50 s impulse
0093–9994/01$10.00 © 2001 IEEE
1424
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
test applied line-to-ground on the transformer primary had only
a 200-V-peak line-to-ground voltage on the secondary side.
neutral solidly connected to earth ground potential
Also, an
reduces the possibility of overstressing line to neutral insulation. A disadvantage is that a line-to-ground fault will interrupt
power to the process line. With a solidly grounded neutral
system, the problem with line-to-ground voltage escalation due
to an arcing ground fault is eliminated, since the neutral is held
to ground potential. However, the disadvantage is that a ground
fault must now be immediately interrupted due to the higher
ground fault current. The load being supplied by this circuit is
immediately shut down without warning. The higher ground
fault current also has the possibility to inflict more damage due
to the higher energy to be dissipated. Other circuits may also be
disrupted due to the severe voltage drop caused by the higher
fault current.
The high-resistance grounded (HRG) system of Fig. 1 adds
sufficient neutral-to-ground resistance to limit the ground-fault
current to a value greater than or equal to the system capacitive
charging current, which is normally in the range of 1–5 A.
Although the neutral is not held at earth ground potential as
with the solidly grounded system, this value of resistance will
limit the voltages to ground during an arcing ground fault to
levels sufficiently low to avoid insulation failures. Secondary
attenuation of primary line-to-ground voltage transients depends on the resistor value chosen, which is outside the scope
of this paper. The largest advantage of the high-resistance
neutral grounding method is that a line-to-ground fault does
not require immediate shutdown of the affected equipment.
When the ground-fault alarm is initiated, an orderly shutdown
of the affected equipment can be organized within a reasonable
amount of time period, usually 24 h. The ground fault still must
be given attention, to avoid escalation into a higher magnitude
phase-to-phase fault if another ground fault should occur on a
different phase.
A detriment of HRG systems is additional equipment cost
over a solidly grounded or ungrounded system. The fault on an
HRG system may be more difficult to locate than on a solidly
grounded system because there is usually minimal damage and
the individual affected circuit is not automatically isolated by
circuit breaker or fuse action. The HRG system usually has a
current pulsing scheme to enable tracking down the fault with
a clamp-on ammeter. However, due to the benefits of providing
for continuous operation, minimum ground-fault damage, and
avoiding voltage escalation during ground faults, the HRG
system, used in conjunction with a ground-fault indicator (GFI)
meter, has become the standard practice in the petrochemical
process industry.
B. GFI Alarm Philosophy
Voltage-sensing and current-sensing GFI schemes are shown
-to-ground
in Fig. 2. In voltage-sensing GFIs, voltage across
is sensed with a high-impedance voltmeter. The high input
impedance of the GFI meter insures the desired ground-fault
current trip point is not desensitized by a parallel current path
through the meter.
(a)
(b)
(c)
Fig. 2. Ground-fault detection schemes with HRG/GFI systems. (a) Voltage
sensing. (b) Current sensing. (c) GFI sensing ungrounded system.
In the Fig. 2 current-sensing GFI scheme, current through the
grounding resistor passes through a current transformer (CT)
and is sensed by a low-impedance ammeter on the CT secondary. The low input impedance of the GFI ammeter insures
-to-ground voltage is across the resistor and
essentially all of
not the meter, so that the desired ground fault current trip point
is not desensitized.
Various GFI alarm philosophies exist for both GFI schemes.
) is typically sized for 1–5 A
Maximum rms fault current (
).
is the neutral grounding reand is defined by (
is rated rms line-to-neutral voltage. Maximum
sistor and
to ground is
, which is 277 V on a 480-V
voltage across
system. In general, it is desirable to set an alarm trip point as
low as possible without nuisance tripping.
In a balanced three-phase system without ASDs, neutral
voltage is theoretically near zero potential, so a GFI voltmeter
would normally read 0 V and a GFI ammeter scheme would
read 0 A. A relatively low GFI alarm level ( 10% of the maximum voltage or current value) is desirable to provide sensitive ground-fault detection, especially for wye-connected loads
where a failure near the neutral point results in relatively little
fault current. In practice, line-to-ground 60-Hz capacitive cable
charging current, 60-Hz nonlinear loads such as transformer
magnetizing current or fluorescent lighting ballasts wired line
to ground, and unbalanced loads may exist. These parasitics are
discernible on most voltmeters. As a result, a minimum practical voltage trip point is usually in the 10-V range. GFI voltmeters usually contain high-frequency filters or time delays to
prevent false alarms during normal system transients that occur,
e.g., across the line motor starting. A GFI ammeter approach
is somewhat less sensitive in that neutral current for a typical
and 10-V
-to-ground voltage is barely seen
on the 0–5-A scale of the GFI. Historically, a minimum pickup
setting of 10% of maximum voltage or current has proven satisfactory for most applications.
In a balanced three-phase system with ASDs, neutral
voltage is not close to zero potential, so nuisance alarms and
masking of legitimate faults on existing standard HRG and GFI
system is possible. This paper discusses how ASDs generate
transient zero-sequence current, which, in turn, creates a
voltage at the GFI. The effect the ASD transient
nonzero
noise current has on GFI and methods to mitigate the effect of
this noise source on existing HRG systems are described.
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
Fig. 3. Line-to-ground I
system.
1425
cable charging path in a solidly grounded ASD
II. GENERATION OF ASD ZERO-SEQUENCE CURRENT
A. Zero-Sequence Current of Present-Generation Insulated
Gate Bipolar Transistor (IGBT) Drives
Fig. 3 shows how an ASD generates a repetitive transient
zero-sequence current in the neutral circuit of the drive feeder
) is connected to
transformer. Transformer secondary (
) ac input. An ASD input diode bridge conthe ASD (
). The ASD
verts the ac input voltage to a dc-bus voltage (
three-phase inverter uses six IGBTs semiconductor switches to
to the line-to-line (
) ac output. The
connect
motor frame is grounded to a plant ground grid at Potential
#3 and a motor Power Equipment (PE) ground wire is also
connected back to the ASD PE frame ground and plant grid
ground at Potential #1. The typical ASD system of Fig. 3 is
earth grounded at a steel girder or ground rod arrangement at
Potential #4, where the transformer neutral is solidly grounded.
) of
The ASD line-to-line output voltage waveform (
Fig. 4 consists of a series of pulses controlled by a pulsewidth
.
modulator (PWM) with peak pulse voltage equal to
Pulsewidth ( ) and dwell time at zero are PWM controlled so
a fundamental sine-wave output voltage at a desired variable
output frequency ( ) is obtained. Pulse spacing is controlled
to maintain a constant V/Hz ratio (460V at 60 Hz or 7.6 V/Hz)
to maximize torque production over the range. Output pulse
rise/fall times of Fig. 4 are dictated by the ASD semiconductor
switching times. Present-generation ASDs use IGBTs with
ns. The pulse repetition rate is called
rise/fall times
carrier frequency ( ) and for IGBT ASDs is user selectable
1–2 kHz on high-horsepower drives and up to 12
with
kHz possible on low-horsepower drives.
A transient line-to-ground capacitive cable charging current
outputs of Fig. 3
( ) is sourced from the ASD , , or
pulse edge of Fig. 4. The
magnitude
during every
and is between 5000–15 000 V/ s, depending
paths exist in
on system voltage and IGBT rise time. Two
Fig. 3, one through line-to-ground cable capacitance ( - )
and another through line-to-ground capacitance ( - ) of the
magnitude is simply defined as
motor stator winding. Peak
,
and
observed as high as 25 Apk
current refor long cable runs and high-horsepower motors.
turns to ASD ground Potential #1 either by the PE ground wire
or through the motor-ASD ground grid, depending on which is
Fig. 4. PWM line-to-line voltage V , line-to-ground I cable charging
current, voltage V at earth ground Potential #4, and common-mode (CM)
voltage V due to I flowing across Potential #1 to Potential #2 in the ground
grid.
Fig. 5. Laboratory-measured V
and I of 480-V drive at 30 Hz, 300-ft
motor cable (“0” at offset 4 div; 2 ms/div; 2 A/div; 100 V/div).
the lower impedance at the
oscillation frequency. No physical connection exists between ASD PE ground Potential #1 and
the ASD dc bus. Thus, the only remaining path to complete the
route back to the ASD is through the feeder transformer neu).
tral and drive input phase wires (
goes through the feed transThe key point is that transient
former neutral circuit and its magnitude is determined by the
sum total of system impedance in the path. Also, the transformer
currents from multiple ASDs on the
neutral circuit will sum
same feeder. The resulting instantaneous neutral current waveform will change, based on the additive or subtractive nature of
polarity magnitude and set point of the respective ASDs.
A convenient way of investigating all three ASD
line-to-ground outputs simultaneously is by establishing a
voltage in Fig. 3
virtual neutral reference node to ground.
current sourced from the
is the voltage generator behind the
ASD and is measured from a wye-connected node of three 1
waveform
M resistors to ground. As seen in Fig. 5, the
is comprised of a low-frequency 180-Hz-amplitude modulated
ripple component that is further modulated by the PWM
high-frequency switching. The ( ) 180-Hz ripple waveform
is due to the three-pulse-ripple voltage measured at the input
diode bridge ( ) terminal to true earth ground [3]. Likewise,
( ) 180 Hz ripple is due to three-pulse-ripple voltage measured
at the input diode bridge ( ) terminal to ground.
showing the
Fig. 6 is a time-expanded scale of Fig. 5
higher frequency step-like PWM voltage component. Fig. 6
1426
Fig. 6. Expanded V
voltage and I charging current at f
at offset 3 div; 50 s/div; 2 A/div; 100 V/div).
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
= 4 kHz (“0”
current waveform is the combined zero-sequence or CM line-tocurrent
ground current from all ASD phases. The 8-Apk
spike of Fig. 6 is typical of systems using solidly grounded
magnitude is discussed
transformers. Exact calculation of
kHz in Fig. 6, the current spike repein [4]. Although
tition rate into the ground circuit is closer to 20 kHz, because
every phase switching contributes line-to-ground current tranrepetition rate is estimated as
– .
sients. Thus, the
B. Zero-Sequence Current of Past-Generation BJT Drives
Previous-generation ASDs used BJT semiconductors with
rise/fall times of 1–2 s. For the same system capacitance,
magnitude is
lower than in IGBT drives because of the
. Also, BJT drives had a maximum of 750
lower output
current and lower
explains
Hz–1.5 kHz. Thus, reduced
why there were fewer problems of nuisance GFI alarms with
past-generation drives.
Fig. 7. Voltage across R with ASD operating at f = 6 kHz and with ASD
off. Top trace: with ASD operating (“0” level at offset 4, 50 V/div, 1 ms/div).
Bottom trace: with ASD off (“0” level at offset 1, 50 V/div, 1 ms/div).
across the high resistance in
to ground. The GFI meter usually has insufficient filter action to prevent false alarms during
transients. As a
the voltage spike intervals induced by ASD
result, the initial desired GFI alarm/trip point must be substantially increased to prevent false or nuisance trips. Consider the
single ASD operating waveform of the top trace of Fig. 7 which
contains a high-frequency rms voltage of 45 Vrms and voltage
. It is unclear where the GFI
spikes up to 175 Vpk across
alarm must be set. The problem is that the GFI set point may
have to be so high that it cannot reliably detect a ground fault
when the ASDs are operating. Also, unless precautions are taken
in rating the neutral grounding resistor, the power dissipation of
the ASD high-frequency component in the resistor may exceed
the power rating, especially when combined with the added fundamental frequency component during a ground fault.
IV. METHODS TO REDUCE
III. EFFECT OF ASD ZERO-SEQUENCE CURRENT ON GFIs
tranSystem grounding philosophy affects the ASD
sient zero-sequence current magnitude and circuit path. The
return path back
ungrounded system of Fig. 1 breaks the
and CM noise voltage
to the ASD in Fig. 3. Thus,
) developed across the ground grid of Fig. 3 does
(e.g.,
not exist. The grounded wye system of Fig. 3 detrimentally
completes a transient zero-sequence noise current return path
current is highest with grounded
as previously discussed.
current is contained totally within
systems. However, the
the transformer’s solidly grounded neutral ( ) and secondary
zero-sequence current cannot flow into the transcircuit.
former primary circuit of Fig. 3. Thus, the benefit of a solidly
return
grounded transformer of Fig. 3 is that it controls the
path to the ASD and insures that other sensitive equipment or
plant HRG unit connected on the primary side are not affected.
An HRG system has 55–277 (1–5 A) in series with the
zero-sequence noise current return path which significantly recurrent. Thus, CM voltage differences across
duces peak
the ground grid are smaller than with solidly grounded systems.
The bottom trace of Fig. 7 shows that in a balanced three-phase
is
system with the ASDs off, the neutral point voltage at
close to zero potential. The top trace of Fig. 7 shows that in a balanced three-phase system with ASDs operating, the neutral
point voltage is not close to zero potential, which is due to the
zero-sequence current developing a transient voltage
ASD
CURRENT IN THE
NEUTRAL
There are a few methods that minimize, but may not eliminate, ASD zero-sequence current and its effect on the “ASDgenerated voltage” developed across the neutral grounding resistor.
• Lower the carrier frequency .
• Reduce the number of ASDs on a system transformer.
• Operate the ASD closer to 60 Hz.
• Add CM chokes to drive output.
• Add output line reactor.
• Add an electromagnetic interference (EMI) filter to ASD
input.
• Add CM capacitors to each ASD dc bus.
• Investigate lighting surge capacitors in plant.
• Use shielded or armor cable.
• Use a drive isolation transformer.
Lowering the carrier frequency to the minimum allowable
value (typically 2 kHz) will increase the time between trancurrent spikes in Figs. 6 and the top trace of 7, so
sient
that the GFI meter responds to a lower effective RMS noise
voltage. Section VII-D test results of a single ASD drive show
the “ASD-generated voltage” read on the GFI meter is 1/4–1/6
kHz as compared to 6 kHz.
lower at
Decreasing the number of ASDs on a transformer is imprac-to-ground voltage to accepttical and may not reduce the
able GFI levels. The GFI meter readout obtained under single
drive operation may not necessarily increase as the number of
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
1427
Fig. 9. Addition of drive EMI filter to reroute I
from the HRG unit in the Xo neutral circuit.
back to ASD input and away
Fig. 8. Zero-sequence current generated during switching interval with and
without CM cores added.
ASDs on the transformer is increased. This result is verified and
discussed in Section VIII-A using actual field test data. In any
event, unacceptable GFI readouts still occur as increased numbers of ASDs are added to the system.
-to-ground
Section VII-D test data show ASD-induced
is increased from 0 to 60 Hz. At low
voltage decreases as
speeds, the PWM controller typically gates two or three phases
on simultaneously, which tends to double and triple the total
transient magnitude into ground. As increases, the higher
motor counter EMF (CEMF) voltage per phase tends to reduce
magnitude in the
- branch circuit to ground in Fig. 3.
value (and, therefore,
-to-ground voltage)
Thus, the total
tends to decrease as speed increases. As increases past 45 Hz,
the PWM controller enters the “pulse-dropping mode” which
continuously reduces the actual number of output carrier frequency pulses as speed is increased toward 60 Hz. Thus, as
transpeed is further increased toward 60 Hz, the quantity of
rms
sient current spikes into ground is reduced as well as the
-to-ground rms voltage. However, placing restricvalue and
-to-ground
tions on the ASD operating to reduce the GFI
voltage is impractical.
Adding CM inductors to each drive output may reduce peak
transient current (6 Apk reduced to 2 Apk in Fig. 8), but is
usually not enough to get acceptable GFI meter readings.
Adding low-impedance output line reactors (3%–5%) to each
transient current to lower levels, but at
drive will reduce peak
substantial size, cost, weight, and output voltage drop penalties.
An acceptable 10-V GFI reading with an ASD operating with
output reactors and a 1-A neutral grounding resistor results in
neutral circuit. Thus, there
only 35-mA rms allowed in the
is no guarantee that GFI meter readings will be acceptable with
output reactors.
The drive EMI filter of Fig. 9 may reduce transient
magnitude to very low values in the
neutral circuit. The
filter’s high-frequency bypass capacitors ( ) from line to
away from the grounding resistor
ground reroute transient
). Series filter inductors
and back to the ASD inputs (
( ) present high impedance to the filter input and further
flowing in the
neutral circuit.
reduce the possibility of
to
CM capacitors added on each drive from
to ground act as high-frequency bypass
ground and
current back to the ASD dc
capacitors which reroute transient
bus in Fig. 3 and away from the grounding resistor in the trans-
Fig. 10. Use of a solidly grounded drive isolation transformer to reroute I
back to ASD input and eliminate any possibility of I flowing in the HRG unit.
former neutral. The difficulty lies in choosing a capacitor that is
an effective solution for every possible system configuration.
) to ground in Fig. 9
Surge capacitors connected line (
may be part of a plant’s lightning protection scheme. These ca-to-ground voltages,
pacitors may inadvertently reduce GFI
since they act as high-frequency bypass capacitors to transient
current.
Shielded tray cable or continuous aluminum sheath armor
cable with a PVC jacket will help prevent interference with other
sensitive equipment when applied to the drive input and output,
and when the shields are grounded on both ends, [5, Fig. 20].
thus tends to flow on the cable shield and not in the
Transient
ground grid. Line-to-shield capacitance of the input cable may
back to the ASD and reduce
-to-ground
return some of
voltage at the GFI. However, shielded cable will not completely
solve the GFI problem, since the grounded shield at the HRG
current to reenter the neutral
circuit and
unit will allow
transformer secondary.
Use of a solidly grounded drive isolation transformer, in
addition to the existing main transformer and HRG unit,
will totally eliminate the GFI problem as shown in Fig. 10.
flows only in the isolation transformer neutral
Transient
circuit and not in the plant HRG neutral circuit. Section VIII-E
field test data show the validity of this approach. The isolation
transformer also beneficially prevents high-frequency
current from flowing in the remainder of the plant electrical
system ground. Although the isolation transformer solves the
problem of the high-frequency current on the neutral ground
resistor, the application must permit immediate tripping of the
motor for a ground fault due to the solidly grounded neutral,
thus negating one of the benefits of the HRG system.
1428
IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
(a)
Fig. 11.
Input impedance and phase angle versus frequency of GFI voltmeter.
The above methods attempted to reduce the ASD
zero-se-to-ground circuit, so that
ground
quence current in the
resistor voltage is reduced and lower GFI alarm set points are
allowed. Each method has a performance versus cost expense
associated with it that must be evaluated. The following seccurrent to flow, but
tions investigate allowing system ASD
solving the GFI readout problem with filtering at the GFI meter.
V. CHARACTERIZATION OF VOLTAGE AND CURRENT GFIS
(b)
Prior to designing a high-frequency zero-sequence noise filter
between the HRG and GFI, it is necessary to characterize the
frequency response of the respective GFIs.
Fig. 12. Neutral grounding resistor voltage and CT secondary current of
current-sensing GFI system measured at field site with multiple PWM IGBT
2:5 kHz. (a) Voltage across
drives operating between 30–60 Hz with f
neutral grounding resistor (“0” at offset 5, 25 V/div). Time scale: 200 s/div.
(b) CT secondary current (“0” at offset 2, 0.2 A/div). Time scale: 2 ms/div.
=
A. Characterization of Voltage-Sensing GFI
Fig. 11 plots input impedance versus. frequency of a typical GFI voltmeter shown in Fig. 2(a). The meter looks like a
simple lag filter with 60-Hz impedance of 620 k and corner
cutoff frequency of 20 kHz. Meter input impedance at 60 Hz is
, and avoids desensitizing the 60-Hz
high, as compared to
ground-fault alarm set point. The 20-kHz corner frequency implies an 8- s filter time constant. Thus, some filtering of the
rise-time component exists, but the ASD zero-sequence
voltage at
–
will overload the meter, causing GFI readout
problems.
B. Characterization of Current-Sensing GFI
Frequency response of a typical current-sensing GFI system
of Fig. 2(b) was evaluated at a field site. System voltage was
was rated at 130 so that 60-Hz ground-fault
480 Vac.
current was limited to 2 Arms. The CT had a unity primary-tosecondary ratio (5 A : 5 A) and fed a digital GFI with a 1-A alarm
setting and 3-s delay. The GFI digital meter read “0” A, even
with 15 PWM IGBT ac drives powered on the feed transformer.
Fig. 12 shows an ASD-induced 75-Vpk voltage developed
across the neutral grounding resistor (0.5 Apk in CT primary),
while the CT secondary response read zero current. The CT iron
lamination thickness, chosen for 60-Hz frequency, could not retransient current.
spond to the high-frequency ASD
Thus, some GFI current-sensing schemes may “inadvertently” solve the GFI readout problem under ASD operation by
their inherent high-frequency rejection characteristics.
Fig. 13.
GFI meter plus high-frequency bypass capacitor.
VI. METHODS TO REDUCE ZERO-SEQUENCE VOLTAGE AT GFI
The advantages and disadvantages of the following solutions
used at the GFI meter location to reduce the effect of zero-sequence voltage are discussed:
• high-frequency bypass capacitor across grounding resistor
and GFI meter;
• low-pass – filter between grounding resistor and GFI
voltmeter;
• impedance-buffered – filter between grounding resistor and GFI meter.
Fig. 13 shows a high-frequency bypass capacitor placed in
parallel with the grounding resistor and GFI meter. There is a
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
Fig. 14.
GFI voltmeter with L–C low-pass filter.
Fig. 15.
95 Hz.
low-frequency and high-frequency circuit design requirement.
The two capacitors in series allow continued operation in the
event of a shorted capacitor. The capacitors are fused to handle
rms value and a 60–Hz fault current component.
the
Requirement 1 :
(1)
Requirement
1429
2:
GFI meter plus impedance-buffered low-pass filter with cutoff at
and greater. Low-pass input to output filter attenuation is ( )
40 dB per decade for every decade past the cutoff frequency.
Requirement #1 implies there are infinite combinations of
– values that satisfy the cutoff frequency. A cutoff frequency
of 200 Hz is required to obtain ( ) 40-dB attenuation at ASD
kHz.
Requirement
(2)
Hz
Requirement #1 implies an effective lowering of parallel
and higher frequencies, so that the
to-ground impedance at
ASD-induced zero-sequence voltage across the parallel combination is substantially reduced. An 8- F capacitor is required to
obtain tolerable GFI voltage readings for an ASD operating at
kHz and grounding resistor values between 150–277 .
through the
The filter does not reduce , but just bypasses
capacitor.
Requirement #2 implies 60-Hz bypass filter impedance be
10 greater than the neutral grounding resistor value, so that
a 60–Hz ground-fault current is not totally bypassed around the
grounding resistor. The10 factor still desensitizes the original
60-Hz fault setting by 10%. Requirement #2 implies
F, while Requirement #1 implies
F to reduce
the ASD-induced high-frequency voltage. Thus, design tradeoffs are required.
The main circuit disadvantage is that a GFI nuisance alarm,
occurring as a result of ASD-induced GFI voltage at frequen, can only be lowered at the expense of desensitizing
cies
the desired 60-Hz set-point value. Another disadvantage is that
high rms ripple current goes through the capacitor. Field measurement may be required for multiple ac drive systems to determine the effective capacitor current rating required.
Advantage of the bypass filter circuit is that it is a quick
low-cost solution, where a desensitized 60-Hz trip value is tolerable. It may be used for applications with a low number of
ASDs in the system, which have low rms capacitor current and
short output cables. Circuit test results are discussed in Sections VII-D and VIII-B.
Fig. 14 shows a low-pass – filter inserted between the
grounding resistor and the GFI voltmeter. The filter is used to
attenuate the ASD-induced zero-sequence voltage occurring at
Requirement
1:
cuto frequency
(3)
2:
(4)
and
impedances are seRequirement #2 implies filter
ries connected and frequency sensitive. Ground-fault current
flowing has parallel paths through the series filter branch and
grounding resistor. Thus, the series filter branch must have a
60-Hz impedance 10 greater than the parallel grounding resistor, so that the original GFI alarm current set point is not desensitized.
It can be shown that selecting – values to meet both remH and
quirements is not possible. For example, an
F with cutoff frequency
of 200 Hz has a 60-Hz
. This circuit is not recommended, since
impedance of
it desensitizes the actual ground-fault 60-Hz component going
grounding resistor by 70%, with most
through a
of the 60-Hz ground-fault current component bypassed through
the lower impedance – filter. This circuit would also allow a
higher ground-fault current, which may not be acceptable on a
system where a ground fault causes an alarm, but is not immediately tripped.
Fig. 15 shows an impedance-buffered low-pass – filter between the grounding resistor and the GFI voltmeter to attenuate
the zero-sequence ASD-induced voltage at and greater. The
is 95 Hz with input–output attenuation of ( ) 40 dB
filter
. The main difference is that the –
per decade beyond
filter impedance is impedance buffered by an input autotransformer. With the autotransformer, the HI–LO filter input terminals have a referred – impedance at 60 Hz which meets the
requirement. This GFI filter is andesired
alyzed and tested in Sections VII and VIII.
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
(a)
Fig. 17.
Measured input–output attenuation of GFI filter versus frequency.
(b)
Fig. 16. (a) Top trace: applied 60–Hz voltage across HI–LO filter input
– ; bottom trace:
terminals; middle trace: voltage across T 1 terminals
voltage across 1 terminals
– . (b) Applied 60-Hz voltage across filter
input terminals HI–LO and filter output terminals HI–LO.
T
X X
H H
VII. IMPEDANCE-BUFFERED LOW-PASS FILTER DESIGN
A. Conceptual Design of the Filter
The input section of Fig. 15 consists of a voltage step-down
. A standard 480-V/120-V control transautotransformer
former is reconnected as an autotransformer with a 5 : 1 ratio.
Fig. 16(a) shows 60-Hz voltage applied to the filter input
–
of
terminals, with 80% of the input voltage across
and 20% across
–
of .
–
is applied to a low-pass filter section
Voltage across
consisting of , , and . Low-voltage capacitor and inductor
components rated for 600 Vac may handle much higher tran1:5
sient voltages across the grounding resistor because of
provides impedance buffering
step-down ratio. Transformer
and is critical in determining the low-pass . Low-pass section
components along with the parameters set a cutoff frequency
Hz. Filter input–output attenuation follows the stan.
dard 40 dB per decade past
voltage is applied to voltage step-up autoCapacitor
, which is a standard 120-V/480-V control
transformer
transformer reconnected as an autotransformer with a 1 : 5
also acts as an impedance buffer, in case
ratio. Transformer
the output terminals are shorted together.
Fig. 16(b) shows there is no input-to-output attenuation of
the 60-Hz component through the filter. Voltage applied to the
HI–LO filter input terminals is stepped downed, passed through
the low-pass section, and stepped up back to the original input
voltage magnitude at the filter output terminals HI–LO. The
sealed filter box physical dimension is 9 in 6 in 4 in and
may be placed at the HRG location.
B. Filter Input–Output Attenuation Versus Frequency
Fig. 17 is a plot of filter input-to-output attenuation versus frequency. Fundamental frequencies between 0–100 Hz are not attenuated. Attenuation of ( ) 40 dB per decade exists beyond the
Fig. 18. Measured input impedance magnitude and phase angle versus
frequency of the impedance-buffered low-pass GFI filter.
95-Hz cutoff frequency up to 1000 Hz. Third harmonic (180 Hz)
voltage in the transformer neutral is attenuated by ( ) 14 dB, if
it exists.
Attenuation greater than ( ) 40 dB/decade occurs for
frequencies 1000 Hz due to high-frequency skin effects,
substantially increasing the ac resistance of the inductor and
transformer coils. The ambient noise floor [( ) 85 dB] is
reached at 6–20 kHz. At 20 kHz and greater frequencies,
the inductance of the iron components is reduced to air core
values. A high-frequency magnetic skin-effect phenomenon
reduces the effective permeability of the iron lamination to the
permeability of air. Beyond 20 kHz, the gradual decrease in
attenuation versus frequency is due to parasitic winding capacitance of the magnetic components becoming predominant
along with the coil skin-effect ac resistance. The following
discussion identifies the critical ASD noise frequencies, which
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
1431
TABLE I
GFI METER READINGS WHEN USED ALONE, WITH A BYPASS CAPACITOR FILTER CIRCUIT, AND WITH AN IMPEDANCE-BUFFERED LOW-PASS FILTER CIRCUIT
(a)
Fig. 20. Single ASD operating in laboratory at f = 6 kHz, f = 3 Hz
with 0.05 F added line to ground on drive outputs. Top trace: Xo-to-ground
voltage (“0”level at offset 4, 250 V/div, 200 s/div); bottom trace: Filter output
(“0” level at offset 1, 50 V/div, 200 s/div).
TABLE II
GFI Vrms OUTPUT IN FIELD VERSUS LABORATORY WITH NO SOLUTION
APPLIED
(b)
=
Fig. 19. (a) Single 2-hp ASD operating at f
6 kHz, f = 30 Hz, 30-ft
cable. Top trace: Xo-to-ground voltage (“0” level at offset 4, 50 V/div, 1
ms/div); bottom trace: filter output (“0” level at offset 1, 50 V/div, 1 ms/div).
(b) Time scale expansion of (a) but at 50 s/div.
need to be attenuated by the filter before entering the GFI
meter. Corresponding filter attenuation at these frequencies is
obtained from Fig. 17. Conversion of decibel-to-attenuation
.
ratio may be done using
The predominant frequency of ASD-induced voltage across
the grounding resistor that corrupts the GFI readout is the carrier
–
frequency components of Fig. 6. Refrequency and
sults of Fig. 17 show a 80-dB to 85-dB reduction (1 : 17 800)
and total removal of these components prior to the GFI meter
connection.
A second ASD frequency component developed across the
grounding resistor is that due to the IGBT 50–200-ns rise times
shown in Fig. 4. Equivalent frequency ( ) relative to rise time
is obtained from Fourier analysis of the PWM pulse and defined
. The
6.4–1.2 MHz frequencies are
by
also greatly attenuated by the filter at ( ) 40 dB to ( ) 50 dB
from Fig. 17 and verified Section VII-D.
A third ASD frequency component developed across the
oscillation
grounding resistor is that due to a 100–400-kHz
following the IGBT rise time of Fig. 8. These components are
also greatly attenuated by the filter at ( ) 60 dB to ( ) 70 dB
from Fig. 17 and verified in Section VII-D.
Thus, the impedance buffered low pass filter is effective in
removing ASD frequency components that might corrupt and
falsely trigger a GFI alarm. It also protects the GFI meter from
transient voltages that might occur in the system.
C. Input Impedance of the GFI Filter
The filter 0–100-Hz input impedance value is critical when
accurate ground fault detection is required. Fig. 18 is a plot of
measured input impedance and phase angle with the filter output
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
(a)
(c)
(b)
(d)
Fig. 21. Waveforms at Field Refinery Site #1a with No Filter Solution and a High-Frequency Bypass Capacitor Solution across the HRG for an ASD operating
at fc
2 kHz and fo = 60 Hz. (a) Xo-to-ground voltage with No Filter Solution. (b) Xo-to-ground voltage using 13-F Bypass Filter Cap Solution across
HRG. (c) Spectrum of No Filter Solution (a) showing 52 Vrms at a fundamental frequency content of 2 kHz and 2-kHz harmonics. (d) Spectrum of Bypass Filter
Cap Solution (b) showing 5 Vrms at a fundamental frequency content of 2 kHz and 2-kHz harmonics.
=
terminals open circuited and short circuited. The open-circuit
case simulates operation with the GFI meter’s 620-k input
impedance. The filter 60-Hz input impedance with its output
open circuited is 4296 . The 60-Hz ground-fault current
component, when it exists, will divide between the grounding
resistor and filter input impedance. The GFI trip level is only
desensitized by 6% and is an improvement as compared to other
filters discussed. The 20-kHz resonant point in Fig. 18 is where
the filter-input impedance is maximum. Beyond 20 kHz, input
impedance drops and input phase angle becomes negative, due to
capacitive effects in the windings of the magnetic components.
The filter input impedance does not change much with the
output terminals short circuited, so that there are no large current
inrushes into the filter when there is an accidental short circuit
on the GFI meter terminals.
D. Laboratory Test Results of Impedance-Buffered
Low-Pass Filter
A single 480-V 2-hp ASD was connected to an input HRG
and to a motor with 30 ft of
transformer with
unshielded cable. The GFI voltmeter reading was recorded for
5, 30, and 60 Hz and with two carrier
ASD operation at
kHz and
kHz. Tests were
frequencies settings of
repeated with an 8- F Bypass Capacitor Filter of Fig. 13 and
Impedance-Buffered Low-Pass Filter of Fig. 15 connected to the
GFI meter.
Table I results show that with no filter solution, the false GFI
readings increase with higher carrier frequency and tend to decrease with increasing ASD output frequency. GFI readings are
generally 20 V for this single drive application with a low-capacitance cable of short length. Both filter circuits of Fig. 13
and Fig. 15 functioned properly when connected, in that they
allowed the GFI meter to accurately read the desired 60-Hz
zero-voltage value.
Fig. 7 shows the input waveform to Fig. 15 filter for the corresponding test conditions described and the ASD operating at
kHz, and
Hz. Filter output voltage recorded
was “0 V.” Fig. 19(a) shows the Fig. 15 filter input and filter
output voltage waveforms for the ASD now operating at
kHz, and
Hz. Fig. 19 shows that the grounding resistor
voltage peaks are lower at the higher fundamental output frequencies. Filter output voltage at the desired zero-voltage value
is still achieved under this operating condition. Fig. 19(b) is a
time-scale expansion of Fig. 19(a) showing the step-like modulation voltage across the neutral grounding resistor.
The next test condition attempted to simulate the appreciable
cable line-to-ground capacitance of a long shielded or armor
cable or a motor with a large line-to-ground capacitance. This
was accomplished by adding 0.05- F CM capacitors on each
ASD output between line and ground. The results of Table I
show large erroneous readings when the GFI meter is used alone
with no filter solution. Table I shows that ASD operation at low
fundamental output frequency is the worst case, as well as ASD
operation at higher carrier frequency.
The 8- F Bypass Capacitor Filter circuit reduces the worst
case 220-V GFI reading down to 20 V in Table I, but did not
reduce it to zero. This suggests that this circuit is probably best
suited for single drive operation with short cables and is subject
to system analysis for every drive application condition.
The Impedance-Buffered Low-Pass Filter circuit reduces the
worst case 220-V GFI reading down to the desired 0-V value
for any ASD operating condition and provides both a technical
and cost-effective filter solution. Fig. 20 shows input and output
filter waveforms with the CM output capacitors added. The
-to-ground
higher ASD output capacitance changes the
voltage waveform as compared to Fig. 19(b). It is seen that the
-to-ground voltage in Fig. 20 does not fully decay to zero before the next switching instant. Peak-to-peak voltage excursions
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
1433
(a)
(b)
(c)
(d)
(e)
Fig. 22. Waveforms at Field Refinery Site #2 with no solution, a GFI filter solution, and a solidly grounded drive isolation transformer solution applied. (a)
Line-to-line output voltage of 200-hp ASD (500 V/div, 2 ms/div). Zero-sequence I current at ASD output (10 A/div, 2 ms/div). (b) Xo-to-ground voltage with
No Solution showing 150-Vpk spikes and step-like modulation (50 V/div, 200 s/div). (c) GFI voltage with Impedance-Buffered Low-Pass Filter Solution (10
V/div, 10 ms/div). (d) Xo-to-ground I current in the ASD isolation transformer when a Drive Isolation Transformer Solution is added to the plant (10 A/div, 50
s/div). (e) GFI voltage across the HRG unit of the main plant transformer when a Drive Isolation Transformer Solution is applied (50 V/div, 200 s/div).
on the transformer neutral
-to-ground voltage can swing from
( ) 750 Vpk to ( ) 750 Vpk depending on system conditions.
The GFI filter of Fig. 15 attenuates these repetitive high-frequency high-magnitude spike voltages and correctly outputs a
zero-voltage value to the GFI meter. The filter output waveform
in Fig. 20, with the added CM capacitors, show a small 5-V peak
spike that decayed to zero in 10 s. The 8- s filter time constant
in the GFI meter can effectively handle this 5-V spike and allowed
the GFI meter to record the desired zero-voltage value.
VIII. FIELD TEST OF GFI FILTER SOLUTIONS
The GFI voltmeter results taken before and after filter solutions were installed are presented. Results were taken at separate
refinery sites during running of the standard process line.
Field Site #1a): This is a 480-V HRG unit of 300 installed
on a 750-kVA transformer which supplied a single 200-hp ASD
and other fixed-speed loads. The ASD was connected to the
motor with 600 feet of 600 V, nonshielded continuous welded
kHz and
armor cable. ASD operation was at
50–60 Hz.
Field Site #1b): This is a 480-V HRG unit of 304 installed
on a 1-MVA transformer with a total bus loading of 400 A. Two
25-hp ASDs with output line reactors were installed and connected to a motor with 700 ft of continuous welded armor cable.
kHz and
Hz.
ASD operation was at
Field Site #2): This is a 480-V HRG unit of 250 installed
on a 1-MVA main transformer which supplied a single 200-hp
ASD and other plant fixed-speed motor loads. The ASD was
connected to a motor with 600 ft of three single conductors plus
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
a ground wire which are unshielded and installed in rigid steel
kHz and
49–50 Hz.
conduit. ASD operation was at
Field Site #3): This is a 480-V HRG unit of 130 installed
on a 1.5-MVA transformer with a total bus loading of 40% contribution from 15 installed ASDs and 60% from other plant
fixed-speed motor loads. The 15 ASDs each had 3% input line
reactors and ranged in horsepower from 7.5 to 150 hp. Continuous welded armor cable was installed on the drive input and
output with lengths ranging from 100 to 700 ft. ASD operation
kHz and
30–60 Hz.
was at
(a)
A. Test Results With No Filter Solution
Field Site #4) had a single 4-kHz IGBT ASD, connected to
a motor with armor cable, and operating on a transformer with
an HRG unit using No Filter Solution at the GFI meter. Table II
field results verify the previous laboratory data taken from Section VII-D and show erroneous GFI voltmeter readings decrease
when ASD output frequency is increased. Field Site #1) data
also show a similar trend for and frequency .
-to-ground voltage across the grounding
Field Site #1a)
resistor with No Filter Solution applied is shown in Fig. 21(a).
output
The waveform is similar in appearance to the drive
voltage waveform of Fig. 6. The ring up voltage on every drive
output switching step is due to reflected wave interaction of
the long output cable and motor line-to-ground capacitance.
The HRG waveform has a peak voltage of 150 Vpk and a cycle
that repeats at the expected 2-kHz (500 s) carrier frequency.
Fig. 21(c) shows the voltage harmonic spectrum taken across
the HRG/GFI unit that corresponds to the time window of
Fig. 21(a). The 2-kHz 150-Vpk waveform has a fundamental
component of 52 Vrms. This voltage gives false GFI readings.
Due to the step-like modulated waveform, the appearance of
(8 kHz)–
(12 kHz) harmonics in Fig. 21(c) are also seen
as previously discussed in Section II. The long armor cable
adds significant line-to-ground capacitance, so that the GFI
52-Vrms readout is close to the Table I 60-Hz laborarory data
entry labeled “GFI meter alone with CM output capacitors.”
-to-ground voltage across the grounding reField Site #2)
sistor with No Filter Solution applied is shown in Fig. 22(b).
The waveform is also similar in appearance to the drive
output voltage waveform of Fig. 6. The HRG waveform also
has a peak voltage of 150 Vpk and a cycle that repeats at the expected 2-kHz (500 s) carrier frequency. In this case, the large
line-to-ground capacitance of the 200-hp motor is predominant
over the smaller cable-to-conduit capacitance in determining the
neutral voltage waveform.
-to-ground voltage across the grounding reField Site #3)
sistor with No Filter Solution applied is shown in Fig. 12. The
waveform is not at all similar in appearance to previous Field Site
cases. The HRG waveform, which had voltage peaks to 150 Vpk,
now has an occasional peak voltage of only 75 Vpk with 25 Vpk
being more common. This is due to the instantaneous ( ) and ( )
waveform adding and subtracting in the
peaks of each ASD
neutral circuit. The instantaneous sum of all 15 ASD
currents tends to reduce the high peak values and create more of a
“white noise average” waveform at a lower rms value. This is interesting effect, since many of the 15 drives investigated had
currents of 25 Apk to ground on the ASD output.
(b)
Fig. 23. Waveform at Field Refinery Site #1b with ASD Output Line Reactor
Solution and Impedance-Buffered Low-Pass Filter Solution for a single ASD
operating at
2 kHz and
= 10 Hz. (a)
-to-ground voltage with
ASD Output Line Reactor Solution installed. (b) Voltage across GFI (output of
Impedance-Buffered Low-Pass Filter) with ASD Output Line Reactor Solution
and Impedance-Buffered Low-Pass Filter Solution installed.
fc =
fo
Xo
B. Solution at GFI Using a High-Frequency Bypass Capacitor
-to-ground voltage across the grounding
Field Site #1a)
resistor with a parallel 13- F High-Frequency Bypass Capacitor
Solution applied is shown in Fig. 21(b). The HRG peak voltage
is now reduced by a factor of ten from 150 Vpk with No Solution
to less than 10 Vpk, with a waveform cycle that repeats at the
expected 2-kHz carrier frequency. The capacitor addition causes
increased high-frequency noise ringing on every step edge
due to interaction with transformer leakage inductance, cable
inductance, or possibly some amount of motor stator leakage
inductance. Fig. 21(d) shows the voltage harmonic spectrum
taken across the HRG/GFI unit that corresponds to Fig. 21(b).
The 2-kHz 10-Vpk waveform has a fundamental component
of a 5 Vrms, which is an acceptable GFI reading The step-like
waveform with the increased high-frequency ringing leads to the
to
harmonics being higher in magnitude.
C. Solution at the ASD Using Output Line Reactor
-to-ground voltage across the grounding
Field Site #1b)
resistor is shown in Fig. 23(a) when output line reactors were
installed and one of the 25-hp ASDs shut off. The waveform
is not similar in appearance to the previous No Filter Solution
voltage waveforms of Fig. 21(a). The line reactor interacts with
line-to-ground capacitance to smooth out the waveform. The
HRG waveform has a peak voltage of 20 Vpk. In this case, it is
seen that the line reactor can reduce, but not eliminate the ASD
-to-ground voltage across the grounding resistor.
induced
D. Solution at GFI Using Impedance-Buffered Low-Pass Filter
Field Site #1b) conditions of Fig. 23(a) were modified to insert an Impedance Buffered Low Pass Filter Solution between
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
1435
TABLE III
PRO AND CON COMPARISON OF NO SOLUTION WITH VARIOUS SOLUTIONS LOCATED AT THE DRIVE OR AT THE GFI LOCATION
the
-to-ground voltage across the grounding resistor and the
GFI voltmeter input. These results are shown in Fig. 23(b). The
20-Vpk carrier-frequency-based ripple voltage across the HRG
of Fig. 23(a) is effectively eliminated on the filter output side
that feeds the GFI meter. The 180-Hz 1.0-Vpk waveform of
Fig. 23(b) that remains on the filter output side, is due to the
ASD ( ) and ( ) dc-bus rectifier ripple voltage-to-ground component that can charge output cable capacitance. This phenomenon was discussed in Section II. Filter attenuation at 180 Hz is
only 14 dB, so that the GFI side voltage at this frequency is
not totally eliminated.
Field Site #2) conditions of Fig. 22(b) were modified to insert an Impedance-Buffered Low-Pass Filter Solution between
-to-ground voltage across the grounding resistor and the
the
GFI Voltmeter input. The Impedance-Buffered Low-Pass Filter
output voltage waveform is shown in Fig. 22(c) for the input
filter waveform of Fig. 22(b). The filter reduced peak GFI meter
voltage from 150 Vpk to less than 20 Vpk. Filter output voltage
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IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS, VOL. 37, NO. 5, SEPTEMBER/OCTOBER 2001
has a 180-Hz component voltage that is further amplitude modulated by a 10-Vpk–15-Vpk ( 10 Vrms) component of unknown
25-Hz–30-Hz origin. The corresponding GFI meter reading was
an acceptable 8 V. Due to the high value of neutral grounding
resistance, a relatively small amount (50 ma) of 180- and 25-Hz
harmonic current develops a significant voltage (8 Vrms) across
the resistor and indicating voltmeter with GFI voltage sensing
schemes.
E. Solution at the ASD Using Drive Isolation Transformers
Field Site #2) was modified per Fig. 10 to insert a solidly
grounded drive isolation transformer between the 200-hp ASD
and main plant transformer where the HRG unit is located.
Fig. 22(a) shows the line-to-line output voltage of the 200-hp
current at every ASD
ASD and the resulting zero-sequence
output switching instant after the drive isolation transformer
currents up to 30 Apk were recorded.
was installed. Peak
zero-sequence current is
Fig. 22(d) shows the transient
-to-ground neutral circuit of
totally contained within the
the ASD isolation transformer. Fig. 22(e) shows the resulting
GFI voltage across the HRG unit of the main plant transformer
is completely clean. The advantage is it effectively solves
the problem and keeps ASD zero-sequence high-frequency
current from the rest of the plant. Also, several drives can share
one isolation transformer. A disadvantage is cost. Another
disadvantage is that for the solidly grounded neutral the benefit
of a high-resistance grounded system is eliminated on the
transformer output, that is, you have to trip immediately for a
ground fault.
results of these investigations are supported by laboratory and
field measurements.
It is always desirable to know when to apply corrective measures to avoid a GFI problem up front before the equipment is
installed, so that the system can be properly designed from the
beginning rather than finding the problem in the ASD startup
phase. There is a wide variety of voltage-sensing GFI vendors,
current-sensing GFI vendors, and ASD vendors. It is difficult
to predict when a GFI problem will occur since vendor variation must also be considered along with system application
variations in line-to-ground capacitance with horsepower, cable
length, cable type, HRG value, pulse rise-time variation, carrier
frequency, output frequency, line reactor usage, and use of single
versus multidrive systems. However, from observations made, it
appears that lower hp drives ( 10 hp) with short output cables
may not create enough ASD-induced zero-sequence voltage to
cause GFI problems.
Having determined when we need to apply a corrective measure, we can conclude that a filter can be applied at the GFI
and the Impedance-Buffered Low-Pass Filter is the most effective filter for voltage-sensing-type GFIs. See Table III for limitations. We can also conclude that some types of current-sensing
GFIs may have CTs that act as inherent filter mechanisms to the
high-frequency components of the zero-sequence current.
However, if it is desired to avoid ASD-generated zero-sequence currents and voltages from entering other portions of
the plant electrical system, an isolation transformer connected
delta-wye grounded can be used and is recommended. See
Table III for limitations.
ACKNOWLEDGMENT
IX. GFI APPLICATION GUIDELINES WHEN USING ASDS
The GFI is designed to provide safe and reliable protection
by sensing and detecting the 60-Hz low-level ground-fault or
leakage current present in a system. This paper has shown a
methodology whereby the erroneous GFI readings and nuisance
false trips/alarms caused by PWM ASDs can be eliminated.
This was done either by: 1) reducing the high-frequency ASD
zero-sequence current that occurs at the drive switching frequency rate; 2) rerouting the high-frequency current away from
the GFI; 3) desensitizing the GFI to the effect of ASD zero-sequence current with local filters; or 4) changing the GFI to current sensing scheme, which on some models, may have some
inherent high-frequency filter mechanism. Table III provides a
summary guideline of the pros and cons of using no filter and
applications that require a GFI solution, be it at the drive or at
the GFI meter location.
X. CONCLUSION
The method and conditions by which an ASD produces a
high-frequency zero-sequence current which interferes with
HRG-GFI systems has been described.
Several corrective measures including different filter arrangements and an isolation transformer have been investigated. The
The authors would like to thank J. Sands, J. Ulloa, and P.
O’Brien of Chevron, J. McQuacker and F. Shewchuk of Candor
Engineering, and J. Mistry, D. Dahl, and H. Jelinek of RA for
their valuable assistance in arranging for and conducting the
field measurements, which provided essential data for the development of this paper. Additionally, the authors would like to
thank L. Berg for his insightful comments and J. Propst for his
support of the paper.
REFERENCES
[1] J. Nelson and P. Sen, “High resistance grounding of low voltage systems:
A standard for the petroleum and chemical industry,” presented at the
IEEE PCIC, Philadelphia, PA, Sept. 26, 1996.
[2] D. Beeman, Industrial Power Systems Handbook. New York: McGraw-Hill, 1955, pp. 286–289.
[3] J. Schaefer, Rectifier Circuits: Theory and Design. New York: Wiley,
1965, pp. 27–30.
[4] G. Skibinski, D. Dahl, K. Pierce, R. Freed, and D. Gilbert, “Installation
considerations for multi motor IGBT AC drives & filters used in metals
industry applications,” presented at the IEEE-IAS Annu. Meeting, St.
Louis, MO, Oct. 1998.
[5] E. Bulington and G. Skibinski, “Cable alternatives for PWM AC drive
applications,” presented at the IEEE PCIC, San Diego, CA, Sept. 26,
1999.
[6] D. Anderson, R. Kerkman, L. Saunders, D. Schlegel, and G. Skibinski,
“Modern drives application issues and solutions tutorial,” presented at
the IEEE PCIC, Philadelphia, PA, Sept. 26, 1996.
SKIBINSKI et al.: EFFECT OF ASDs ON THE OPERATION OF LOW-VOLTAGE GFIs
1437
Gary L. Skibinski received the B.S.E.E. and
M.S.E.E. degrees from the University of Wisconsin,
Milwaukee, and the Ph.D. degree from the University
of Wisconsin, Madison, in 1976, 1980, and 1992,
respectively.
From 1976 to 1980, he was an Electrical Engineer
working on naval nuclear power at Eaton Corporation. From 1981 to 1985, his work as a Senior Project
Engineer at Allen-Bradley Company concerned
servo controllers. During the Ph.D. program, he was
a Consultant for UPS and switch-mode power supply
products at R.T.E. Corporation. He is currently a Principal Research Engineer
with Rockwell Automation–Allen-Bradley Company, Inc., Mequon, WI. His
current interests include power semiconductors, power electronic applications,
and high-frequency high-power converter circuits for ac drives.
John J. Nichols (S’85–M’88) received the B.S.E.E.
degree from the University of Missouri, Columbia,
and the M.S.E.E. degree from the University of Missouri, Rolla, in 1988 and 1994, respectively.
He is currently with the Wood River Refinery,
Tosco Refining Company, Roxana, IL, as the
Electrical Supervisor—Technical.
Mr. Nichols is a member of the IEEE Industry Applications Society and a Licensed Professional Engineer in the State of Illinois.
Barry M. Wood (M’73–SM’87) received the
B.S.E.E. degree from Virginia Polytechnic Institute
and State University, Blacksburg, and the M.S.E.E.
degree from the University of Pittsburgh, Pittsburgh,
PA, in 1972 and 1978, respectively.
From 1972 through 1977, he was with Westinghouse Electric Corporation, Pittsburgh, PA,
as a Power Systems Engineer for the Industry
Services Division. In 1978, he joined McGraw
Edison Company, Canonsburg, PA, as a Senior
Power Systems Engineer and, in 1981, he joined
Electro-Test, Inc., San Ramon, CA, where he held positions of Senior Electrical
Engineer and Supervisory Electrical Engineer. Since 1987, he has been with
Chevron Corporation, where he is currently a Staff Electrical Engineer with
Chevron Research and Technology Company, Richmond, CA. His primary
responsibilities include consulting in the areas of electrical power systems,
adjustable-speed drives, motors, and generators.
Mr. Wood is a Registered Electrical Engineer in the States of California and
Pennsylvania.
Louis A. Barrios (S’84–M’86–SM’00) received the
B.S.E.E. and M.S.E.E. degrees from Louisiana Tech
University, Ruston, in 1987 and 1989, respectively.
From 1989 to 1998, he was with Shell Oil
Company, where he was an Electrical Engineer
in petrochemical plants in Louisiana and Illinois.
In 1998, he joined Equilon Enterprises, a joint
venture between affiliates of Shell Oil Company
and Texaco Inc. Since 1999, he has been with
Equilon Technology, Houston, TX, where he is
currently a Staff Electrical Engineer providing
electrical consulting services to the petrochemical industry. He is a member
of the American Petroleum Institute Subcommittee on Electrical Equipment,
an alternate member on Code Making Panel #1 of the National Electrical
Code, and an alternate member on NFPA 70E—Standard for Electrical Safety
Requirements for Employee Workplaces.
Mr. Barrios is a member of the Executive Subcommittee of the Petroleum and
Chemical Industry Committee of the IEEE Industry Applications Society and a
Registered Electrical Engineer in the State of Louisiana.