88
IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 1, JANUARY 2008
Coupled Input-Series and Output-Parallel
Dual Interleaved Flyback Converter for
High Input Voltage Application
Ting Qian and Brad Lehman, Member, IEEE
Abstract—This paper proposes an integrated magnetic dc–dc
converter suitable for high input voltage application. The converter is based on a coupled input-series and output-parallel dual
interleaved Flyback converter concept. All the center and outer
legs are gapped, and the transformers are integrated into one
magnetic core with not so tight coupling. The gap is beneficial for
suppressing current spike caused by the voltage mismatch between
the windings. The two transformers are inversely coupled, and
current ripple reduction can be achieved with suitable coupling
design. A prototype with 350–450-V input and 24-V/4-A output
is built. Experimental results verify the performance of the new
topology.
Index Terms—Flyback converter, integrated magnetic, interleaved.
I. INTRODUCTION
HERE is an increasing demand in modern power electronics for high density power converters. In most cases,
the size of the magnetic components, including transformers and
inductors, significantly influences the overall profiles of the converters. Integrated magnetic techniques [1]–[17] seem to be suitable solutions for high density application. The attraction is that
transformers and inductors are combined in a single core, and
therefore, cost and size of the converters may be reduced. Generally, there are two dominant types of isolated topologies using
integrated magnetics: buck mode topologies, such as forward
[3], [4], push–pull [6], [7], half-bridge [8], [9] and full-bridge
[8], [9], and buck-boost mode topologies, such as dual Flyback
[15], [16]. More recently, there are newly proposed integrated
boost converters [10], [17] that are beginning to see applications
in automotive power electronics.
For large step-down power conversions, such as 48 to 1 V,
buck mode isolated topologies have become popular solutions.
As a result, integrated magnetic techniques [1]–[9] have been
widely developed for buck mode topologies to minimize the
number of magnetic components and improve the ripple cancellation. Historically, magnetic integration is utilized for forward
converters [3], [4]. However, recent research pays more attention to full wave integrated magnetic dc–dc converters [6]–[9],
T
Manuscript received March 29, 2007; revised June 15, 2007. Recommended
for publication by Associate Editor J. Pomilio.
The authors are with Northeastern University, Department of Electrical and
Computer Engineering, Boston, MA, 02115 USA (e-mail: tqian@ece.neu.edu;
lehman@ece.neu.edu).
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/TPEL.2007.911867
which includes push–pull, half-bridge and full-bridge, etc. For
example, magnetic integration is utilized for push–pull forward
circuit with coupled-inductor current doubler circuit [6], [7]. A
single EI or EE core is used for all the magnetic components,
including the input inductor, the step-down transformer and the
output filtering inductor. The transformer’s primary and secondary windings, as well as the inductor windings, are wound
on the two outer legs. Only the center leg has an air gap. The
flux ripple is cancelled in the center leg, resulting in a lower core
loss in the center leg. As well as reducing the overall size of the
converter, coupled output inductor greatly reduces the current
ripple, and thus improves the power efficiency. Incorporating
an independent inductor winding into the transformer is another
example for full wave integrated magnetic topologies [8], [9].
The original full wave buck mode circuit operation is retained
for this approach. A flexible output inductor winding is added in
the center leg to optimize the design of output inductor. The inductor can be designed specifically to satisfy the current ripple
restrictions and the magnetic constraints.
This research mainly focuses on the discussion of integrated
magnetic techniques related with buck-boost mode isolated
topologies. In this definition, an integrated magnetic buck-boost
mode topology means double or multiple Flyback circuits using
a single magnetic core, since a single Flyback utilizes only one
gapped magnetic core to store the energy in the power transformer. Flyback converter is a popular choice for low power
application due to its simplicity and low cost [18], [19]. For
example, it is suitable for ac–dc converters (a dc–dc converter
stepping down a 400-V input to a relatively low voltage output,
such as 24 V, is generally connected in cascade with a power
factor correction circuit) at a power level of 150 W or less.
Fig. 1 shows a traditional full wave buck-boost (Flyback)
power converter [15], [16] that utilizes one magnetic core to
integrate two Flyback transformers. In this case, the two transformers are coupled with a low reluctance in the outer legs of the
magnetic core. The relatively higher reluctance magnetic property of the inductor is integrated into the magnetic structure
center leg. In other words, only the center leg has the gap. The
or
is turned ON, and
transformers store the energy when
and
are turned
release the energy to the load when both
OFF. Ideally,
and
conduct simultaneously to deliver the
energy to the load so that the two inductors are coupled to reduce
the current ripple. Since the converter has full wave operation,
the duty ratio of each Flyback is less than 50%.
This paper proposes a new integrated magnetic dual Flyback
converter that is more suitable for high input voltage applica-
0885-8993/$25.00 © 2007 IEEE
QIAN AND LEHMAN: COUPLED INPUT-SERIES AND OUTPUT-PARALLEL DUAL INTERLEAVED FLYBACK CONVERTER
Fig. 1. Traditional full wave buck-boost power converter [15], [16].
tions. Specifically, two interleaved Flyback converters are connected in series on the primary side and in parallel on the secondary side. Due to interleaved operation of the Flyback circuits, the current ripple is, thus, reduced as in typical interleaved
Flyback circuits [20]–[22]. A benefit of sharing the output current is that each Flyback handles half of the load power. All the
center and outer legs are gapped in the new converter, and the
transformers are integrated into one magnetic core with not so
tight coupling. This is beneficial for suppressing current spike
caused by the voltage mismatch between the windings. The two
transformers are inversely coupled so that significant current
ripple reduction can still be fulfilled without tight coupling. It
should be remarked that leakage inductance between the primary winding and the related secondary winding cannot be minimized. This problem also exists in other Flyback converters,
since Flyback converters utilize gapped magnetic core for transformers to store and release energy. The details of current ripple
reduction are explained in Section III. The following features
are presented.
• Connecting the primary side in series reduces the voltage
stresses of the primary components. The primary switches
are rated at half of the corresponding voltage values in a
single Flyback or a traditional full wave buck-boost power
converter [15], [16]. The on-resistance and cost of the primary switches are, thus, significantly reduced. Alternatively, this implies that for the same rated switches, the proposed converter can handle twice the input voltage compared with the converter in Fig. 1 [15], [16].
• Interleaved operation with paralleled connection on the
secondary side keeps the benefits of ripple reduction as a
traditional full wave buck-boost power converter [15], [16]
does.
• To satisfy a certain current ripple requirement, magnetizing inductance can be reduced by means of suitable
inductor coupling design. Due to the mutual influence of
the two coupled inductors that is combined with the transformers, further current ripple reduction can be achieved
when compared with interleaved Flyback converters
without coupling. Also, the amount of the magnetic components is reduced by integrating the two transformers
89
into one magnetic core, although the efficiency of the
new approach may not be improved when compared with
conventional discrete interleaved Flybacks.
• Current spike caused by the voltage mismatch between the
windings can be suppressed by gapping all the center and
outer legs. Compared with tight coupling, such as the condition in a traditional full wave buck-boost power converter
shown in Fig. 1, gapping all the legs weakens the coupling
between the two Flybacks’ windings, and thus, reduces the
effect of the voltage mismatch.
Section II of the paper presents basic operation principles of
the proposed converter, highlighting the differences between the
proposed converter and known topologies [15], [16]. Then, Section III presents more technical detail of the coupled inductor
interaction. Section IV gives further magnetic flux analysis. Experimental results are presented in Section V, followed by the
conclusion.
II. OPERATION PRINCIPLE
Fig. 2 shows the proposed coupled input-series and outputparallel dual interleaved Flyback converter, and Fig. 3 represents the related waveforms. The primary switches are rated at
half of the corresponding voltage values in a single Flyback or a
traditional full wave buck-boost power converter [15], [16] due
to series connection of the two Flybacks. Also, gap filling is
used for all the three legs. As shown in Fig. 2, the upper Fly, ,
,
and two windback (Flyback 1) consists of
ings in one outer leg. The lower Flyback (Flyback 2) consists of
, ,
,
and two windings in the other outer leg. Since
gapping all the three legs weakens the coupling, the two Flybacks can actually operate independently (if desired). The current ringing caused by the voltage mismatch between the two
Flybacks’ windings can be significantly suppressed due to the
weakened coupling. Therefore, the duty ratio can exceed 50%.
It should be emphasized that weakened coupling does not necessarily sacrifice ripple cancellation in this case. Detailed ripple
calculation is available in Section III. However, synchronized
operation with suitable coupling design is beneficial for ripple
reduction. This paper mainly analyzes the interleaved condition
and
when duty ratio is less than 50% as shown in Fig. 3.
are the main primary switches.
and
are the voltages
across the primary windings as shown in Fig. 2.
and
are
and
are the
the current through the primary windings.
current through the secondary windings. An E magnetic core is
used in this paper. Each transformer utilizes one outer leg for
windings. The following description explains the detailed principle of the proposed topology. – is defined as one duty cycle
. The length of – and – are defined as
, which repand
are the turns of the
resents the duty ratio. Assume
primary and secondary windings. The input to output transfer
ratio is
(1)
Interval 1 – : The condition of this period is shown in
is turned on. The voltage
Fig. 4(a). During this interval,
across the primary winding of the upper Flyback is equal to
. The upper Flyback circuit delivers energy from
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 1, JANUARY 2008
Fig. 2. Proposed dual coupled Flyback converter.
circuits switch the condition. is turned on. The voltage across
the primary winding of the upper Flyback is equal to
. The lower Flyback circuit delivers energy from
to
,
the transformer on the primary side. The primary current
which is shown in Fig. 2, increases linearly. At the same time,
remains off. The secondary side current of the upper Flyback,
, flows through
and charges
. The current
which is
is applied to
value begins to decrease. The output voltage
the secondary winding of the upper Flyback.
Interval 4 – : Fig. 4(d) shows the condition of the last
interval. This case is the same as interval 2. Both Flyback circuits release energy on the secondary sides. The output voltage
is applied to the two secondary windings.
III. RIPPLE CALCULATION FOR COUPLED CONDITION
For two coupled inductors, the following typical relations can
be obtained [6], [11], [14]:
(2)
(3)
Fig. 3. Timing and waveforms of the proposed circuit.
to the transformer on the primary side. The primary current
, which is shown in Fig. 2, increases linearly. The secondary
is reverse-biased. At the same time,
is off. The
diode
secondary side current of the lower Flyback, which is , flows
and charges
. The current value begins to dethrough
is applied to the seccrease. In this case, the output voltage
ondary winding of the lower Flyback.
– : As shown in Fig. 4(b),
is turned off
Interval 2
during this interval. Both Flyback circuits start to release the
stored energy from the transformer to the output capacitor and
goes through diode
and
goes
load. Therefore, current
through diode
. The voltage across each secondary winding
is equal to .
– : The condition of this interval, which is
Interval 3
shown in Fig. 4(c), is similar as interval 1. The two Flyback
In the above case, let
, and and are
is
the voltages applied on the two corresponding windings.
the coupling inductance.
For this proposed coupled Flyback, the following equations
about the current ripple can be achieved based on typical equations and specific conditions. Using the derived equations, optimized design can be achieved for converters according to their
specific requirement. When all the legs have gaps with approximately same distances, is approximately equal to 1/3. Assume
is the coupling inductance between the two primary windings.
and
are the magnetizing inductance of the primary
and are the input and output voltand secondary winding.
is applied.
ages. For each Flyback,
: During this interval, currents circuInterval 1
late through the primary winding of Flyback 1 and the secondary winding of Flyback 2. These two windings are considis applied to the priered as two coupled inductors. Since
is applied to the secondary
mary winding of Flyback 1, and
QIAN AND LEHMAN: COUPLED INPUT-SERIES AND OUTPUT-PARALLEL DUAL INTERLEAVED FLYBACK CONVERTER
91
Fig. 4. Detailed description of the operation principle of the proposed circuit. (a) Interval 1 (t –t ); (b) Interval 2 (t –t ); (c) Interval 3 (t –t ); (d) Interval 4
( t –t ) .
winding of Flyback 2, the following formulas related with current ripple are obtained according to (2) and (3):
(4)
Since
, we also have
.
The capacitor is in a discharge state.
is increasing approxiis positive.
mately linearly because
– : During this interval, currents circulate
Interval 2
through the secondary winding of both Flyback circuits. These
two windings are considered as two coupled inductors. Since
is applied to the secondary winding of each Flyback, the following formulas related with current ripple can be achieved according to (2) and (3):
, we also have
.
Since
Similar to Interval 1, the capacitor is in a discharge state and
is increasing approximately linearly.
– : During this interval, currents circulate
Interval 4
through the secondary winding of both Flyback circuits. These
two windings are considered as two coupled inductors. Since
is applied to the secondary winding of each Flyback, the following formulas related with current ripple are obtained:
(7)
Similar to Interval II,
.
The capacitor is in a charge state. The two coupled secondary
inductors are both releasing their energy into the load.
Further Discussions on the Proposed Circuit: As shown in
Fig. 3, the current ripple is determined by
(5)
(8)
. The capacitor is
Also,
. The two coupled
in a charge state since
secondary inductors are both releasing their energy into the load.
– : During this interval, currents circulate
Interval 3
through the second winding of Flyback 1 and the primary
winding of Flyback 2. These two windings are considered as
is applied to the secondary
two coupled inductors. Since
winding of Flyback 1, and
is applied to the primary
winding of Flyback 2, the following formulas related with
current ripple are obtained:
(6)
When compared with no coupling condition (suppose using
same inductance value , and the current ripple is determined
.),
by
should be satisfied in order to achieve a reduced current ripple. In other words, smaller inductor can
be used to satisfy identical current ripple requirement when
.
In the prototype for experimental verification, is approximately equal to 1/3 (all the legs have gaps with approximately
same distances).
(
,
,
, and
) is achieved.
Therefore, smaller inductance is required when compared with
two discrete interleaved Flybacks.
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 1, JANUARY 2008
Fig. 5. Magnetic reluctance circuit of the proposed topology.
Specifically, the following differences are remarked when
compared with a traditional converter [15], [16] in Fig. 1.
1) The new approach connects the input of the two Flybacks
in series, while the traditional method connects the input
in parallel.
2) The proposed converter has three gaps in all legs. The traditional converter only has one gap in the center leg.
3) The secondary windings are inversely coupled in the new
converter, and directly coupled in the traditional converter.
From the first difference, it can be concluded that the proposed method has half of the voltage stress when compared
with the traditional method. According to the second and third
differences, the secondary windings of the traditional converter
are directly coupled with tight coupling. However, for practical
circuits, especially for this high voltage application, switches
cannot operate exactly simultaneously. There is a voltage mismatch between the two secondary windings. The voltage difference is applied to the leakage inductance between the two
windings. Because the two outer legs have low reluctances, and
the center leg has a high reluctance, the two secondary windings have good coupling and small leakage inductance. Minor
mismatch can lead to high current spikes and resonance. On
the other hand, the secondary windings of the new converter
are inversely coupled with not so tight coupling since all three
legs are gapped. By gapping all the three legs, a big leakage
inductance between the windings of the two transformers is obtained, since the flux generated by each winding in the outer legs
can go through all three legs. ( is approximately equal to 1/3
when all the legs have gaps with approximately same distances.)
The large inductance value can avoid high current spike when a
voltage mismatch is applied. Also, current ripple reduction can
still be achieved with suitable coupling design. Detailed calculation for current ripple is given in Section III. We also remark that
the weakened coupling between the two transformers allows the
duty ratio to be greater than 50% in the propose circuit.
Fig. 6. Flux path in the core of transformer.
Fig. 7. Equivalent magnetic reluctance circuit during each interval. (a) Interval
1 (t –t ); (b) Interval 2 (t –t ); (c) Interval 3 (t –t ); (d) Interval 4 (t –t ).
metrical. Also, Fig. 6 shows the flux path of in the core of transformer. Because of the air gap, the center leg is no longer a
low magnetic reluctance path for the fluxes. The flux generated by each winding in the outer legs can go through all three
legs. The flux interaction between the windings affects the current ripple and peak values of the flux densities. According to
Fig. 5, the following fundamental equations is, thus, obtained
[6], [11], [14]:
(9)
IV. FLUX DISTRIBUTIONS ANALYSIS
For the convenience of magnetic design, it is necessary to provide the formulas for flux distributions. To prevent the magnetic
core from saturating, the peak values of the flux densities in all
three legs should be kept below a specified maximum value.
Therefore, this section will discuss the detailed calculation for
both the ac fluxes and dc fluxes in all three legs. As a result, the
peak values of the fluxes are the sum of the dc fluxes and half
of the ac fluxes.
Fig. 5 shows the magnetic reluctance circuit of the proposed
topology, assuming the conditions of the two outer legs are sym-
(10)
Furthermore, Fig. 7 shows the detailed equivalent magnetic
reluctance circuit during each interval. The effect of the corresponding winding is removed when the current in the winding
is zero.
A. AC Fluxes
According to Farad’s law, the ac fluxes in the two outer legs,
and
, are determined by the time integral of voltages
QIAN AND LEHMAN: COUPLED INPUT-SERIES AND OUTPUT-PARALLEL DUAL INTERLEAVED FLYBACK CONVERTER
across the windings. Therefore, the magnitude of the peak-topeak ac fluxes in the two outer legs can be derived as follows:
93
and
Since the averaged value of
is
, according to (9), (10), and Fig. 7, the
dc fluxes for all the outer and center legs are derived as
(11)
(21)
On the other hand, the flux rate of the center leg is derived by
summing the flux rates of both the outer legs
(22)
(12)
As a result, the peak values of the fluxes are the sum of the
dc fluxes and half of the ac fluxes.
During Interval 1
– ,
and
are applied to the
primary winding of Flyback 1 and the secondary winding of
Flyback 2. The flux rates of the two outer legs are
and
can be derived as
(23)
. Similarly, the flux rate during all intervals
–
(13)
–
(14)
–
(15)
–
(16)
(24)
To avoid the saturation of the magnetic core, the peak value of
the flux densities of all three legs should be computed. Assume
is the crossing area of the center leg, the peak values are
derived as follows:
(25)
As a result, the peak-to-peak ac flux in the center leg is
achieved according to (13)–(16)
(17)
(26)
By using the calculation of the current ripple (8) and the peak
flux densities (25), (26), suitable magnetic core size and the turn
numbers of the windings can be chosen to satisfy the current
ripple restrictions and the magnetic constraints.
B. DC Fluxes
V. EXPERIMENTAL RESULTS
Another important factor in integrated magnetic design is the
dc flux bias in the core. The detailed flux distribution is shown
in Fig. 6. The fluxes , , and in the three core legs can be
determined from Fig. 5 and Fig. 7 as
(18)
(19)
(20)
To verify the principle of the proposed scheme, a prototype with 350–450-V input and 24-V/4-A output is built
by using the proposed topology switched at 200-kHz hard
switched. FDD6N50TF (500-V, 6-A, DPAK) is used for the
primary switches. (It should be remarked that MOSFETs rated
at 800–1000 V are needed in this specification for a single
Flyback or a full wave Flyback [15], [16] shown in Fig. 1.)
and
) use 12CWQ10FN (100-V,
Secondary diodes (
12-A, DPAK). The two coupled transformers are integrated
into one E22 magnetic core. The cross-sectional area of the
magnetic core is 78.5 mm . The gaps are the same in each
leg. Therefore, is approximately equal to 1/3. Each power
transformer has 60 turns for the primary winding and 13 turns
for the secondary winding. Using (25) and (26), the designed
peak values of the flux densities of all three legs are below 2600
Gauss during the whole input voltage variation range at full
load. In this prototype, the drain-source of each primary switch
is paralleled with a 330-pF capacitor. On the secondary side, a
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IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 1, JANUARY 2008
Fig. 8. V
(Channel 1, 250 V/div), V
(Channel 2, 250 V/div), primary current (Channel 3, 1 A/div) and voltage across the secondary diode (Channel 4,
50 V/div). (a) Primary current of the upper Flyback is shown; (b) primary current of the lower Flyback is shown.
Fig. 9. V
(Channel 1, 250 V/div), V
(Channel 2, 250 V/div), secondary current (Channel 3, 5 A/div) and voltage across the secondary diode (Channel 4,
50 V/div). (a) Secondary current of the upper Flyback is shown; (b) secondary current of the lower Flyback is shown.
75- resistor and a 1000-pF capacitor (connected in series) are
connected in parallel with each secondary winding.
(Channel 1, 250 V/div),
Fig. 8 shows the waveforms of
(Channel 2, 250 V/div), primary current (Channel 3, 5
A/div) and voltage across the secondary diode (Channel 4,
50 V/div) when the input voltage is 400 V. Fig. 9 shows the
(Channel 1, 250 V/div),
(Channel
waveforms of
2, 250 V/div), secondary current (Channel 3, 5 A/div) and
voltage across the secondary diode (Channel 4, 50 V/div)
when the input voltage is 400 V. The two Flyback circuits are
interleaved. The waveforms of the secondary side are in phase
with the primary side. The change rate of the secondary current
of one Flyback differs as expected when the other Flyback
operates at different modes. This phenomenon occurs due to
the mutual effect between the coupled windings. By comparing
the waveforms from the upper and lower Flyback circuits, it
can be clearly seen that the two parts are self-balanced. Each
part shares half of the input voltage. As seen in Fig. 8, the
voltage across the primary winding of the transformer is equal
to half of the input voltage when the main primary switch is
drops to zero. Since
is approximately
turned on and
equal to 1/3 when all the legs have gaps with approximately
same distances, the expected current ripple can be derived
according to (8). The magnetizing inductance of the primary
. The calculated primary
windings in the prototype is 460
peak-to-peak current ripple is, thus, approximately 0.71 A,
which is close to the peak-to-peak values of about 0.75 A in
channel 3 of Fig. 8(a) and (b). Also, there are some oscillations
on the primary currents. The ringing is caused due to the effect
of the leakage inductance between the primary winding and the
related secondary winding.
To keep more safety margin, clamp circuit is recommended
to reduce the voltage ringing across the switches. An alternate
method to achieve more safety margin is to use higher voltage
rated MOSFETs. It is possible that the power conversion efficiency of the converter might decrease 1%–2%. Also, improving
the coupling between the primary winding and the related secondary winding is beneficial for ringing suppression.
By applying the proposed coupled input-series and
output-parallel dual interleaved Flyback converter, an efficiency of 89.1% at 400-V input and 24-V/4-A output is
obtained. Fig. 10 shows the efficiency curve during the whole
input voltage variation range. The experimental results verify
the principle and performance of the scheme.
QIAN AND LEHMAN: COUPLED INPUT-SERIES AND OUTPUT-PARALLEL DUAL INTERLEAVED FLYBACK CONVERTER
Fig. 10. Efficiency of the built prototype.
VI. CONCLUSION
A coupled input-series and output-parallel dual interleaved
Flyback converter for high input voltage application is proposed. Connecting the primary side in series reduces the voltage
stresses of the primary components. All the center and outer
legs are gapped, and the transformers are integrated into one
magnetic core with not so tight coupling. The current ringing
caused by the voltage mismatch between the two Flybacks’
windings can be suppressed due to the weakened coupling.
The two transformers are inversely coupled so that significant
current ripple reduction can still be fulfilled with not so tight
coupling. Experimental results verify the performance of the
new topology.
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Ting Qian received the B.S and M.S. degrees in
electrical engineering from Zhejiang University,
Hangzhou, Zhejiang, China, in 1999 and 2002,
respectively, and the Ph.D. degree in electrical
engineering from Northeastern University, Boston,
MA, in 2007.
He is currently with Texas Instruments, Warwick,
RI. His research interests include high-density dc–dc
power converters, active power filters, and digital
control on power electronics.
Brad Lehman (M’92) received the B.E.E. degree
from the Georgia Institute of Technology, Atlanta,
the M.S.E.E. from University of Illinois at Champaign-Urbana, and the Ph.D. E.E. from the Georgia
Institute of Technology in 1987, 1988, and 1992,
respectively.
Previously, he was a Hearin Hess Distinguished
Assistant Professor at Mississippi State University,
Mississippi State, MS, an NSF Presidential Faculty
Fellow, a recipient of an Alcoa Science Foundation
Fellowship, a Visiting Scientist at the Massachusetts
Institute of Technology (MIT), and the head swimming and diving coach at the
Georgia Institute of Technology. He is presently a Professor in the Department
of Electrical and Computer Engineering at Northeastern University, Boston,
MA. He conducts esearch in the areas of power electronics, with a primary
focus of his research is in the modeling, design and control of dc–dc converters
with applications to high brightness LEDs, battery chargers, telecommunication
power supplies, and VRMs.
Prof. Lehman serves as an Associate Editor of the IEEE TRANSACTIONS ON
POWER ELECTRONICS, and from 1993–1997, he served as an Associate Editor
for the IEEE TRANSACTIONS ON AUTOMATIC CONTROL. In 1999, he served as a
science advisor to the to Commonwealth of Massachusetts, Science and Technology Committee (State Senate) for the Y2K issue in the Power Industry.