Application Note 21
July 1986
Composite Amplifiers
Jim Williams
Amplifier design, regardless of the technology utilized, is
a study in compromise. Device limitations make it difficult
for a particular amplifier to achieve optimal speed, drift,
bias current, noise and power output specifications. As
such, various amplifier families emphasizing one or more
of these areas have evolved. Some amplifiers are very good
attempts at doing everything well, but the best achievable
performance figures are limited to dedicated designs.
designed with little attention to DC biasing considerations
if a separate stabilizing stage is employed.
Figure 1 shows a composite made up of an LT®1012 low drift
device and an LT1022 high speed amplifier. The overall circuit is a unity-gain inverter, with the summing node located
at the junction of three 10k resistors. The LT1012 monitors
this summing node, compares it to ground and drives the
LT1022’s positive input, completing a DC stabilizing loop
around the LT1022. The 10k-300pF time constant at the
LT1012 limits its response to low frequency signals. The
LT1022 handles high frequency inputs while the LT1012
stabilizes the DC operating point. The 4.7k-220Ω divider
at the LT1022 prevents excessive input overdrive during
start-up. This circuit combines the LT1012’s 35μV offset
and 1.5V/°C drift with the LT1022’s 23V/μs slew rate and
300kHz full-power bandwidth. Bias current, dominated by
the LT1012, is about 100pA.
Practical applications often require an amplifier that
has extremely high performance in several areas. For
example, high speed and DC precision are often needed.
If a single device cannot simultaneously achieve the desired characteristics, a composite amplifier made up of
two (or more) devices can be configured to do the job.
Composite designs combine the best features of two or
more amplifiers to achieve performance unobtainable in
a single device. More subtly, composite designs permit
circuit approaches which are normally impractical. This
is particularly true of high speed stages which may be
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10pF
10k
10k
EIN
10k
300pF
–
–
4.7k
+
LT1012
+
LT1022
OUTPUT
AN21 F01
200Ω
Figure 1. Basic DC-Stabilized Fast Amplifier
an21f
AN21-1
Application Note 21
Figure 2 is similar, but uses discrete FETs to more than
triple the speed. Here A1’s input stage is turned off by
connecting its inputs to the negative rail. The differentially
connected FETs bias the second state via A1’s offset pins.
This connection replaces A1’s input stage, reducing bias
current and increasing speed. FET mismatch would normally result in excessive offset and drift. A2 corrects this by
monitoring the summing point (the junction of the two 4.7k
resistors) and forcing Q2’s gate to eliminate overall offset.
The 10k-1000pF pair limits A2’s response to low frequency,
and the 1k divider chain prevents overdrive to Q2 on startup. The 1k-10pF damper at the summing node aids high
frequency stability. Figure 3 shows pulse response. Trace
A is the input and Trace B the output. Slew rate exceeds
100V/μs, with clean damping. Full-power bandwidth is
about 1MHz, and the input bias current is in the 100pA
range. DC offset and drift are similar to Figure 1.
Figure 4 shows a highly stable unity-gain buffer with good
speed and high input impedance. Q1 and Q2 constitute
a simple, high speed FET input buffer. Q1 functions as a
source follower, with the Q2 current source load setting
the drain-source channel current. The LT1010 buffer
provides output drive capability for cables or whatever
load is required. Normally, this open-loop configuration
would be quite drifty because there is no DC feedback. The
LTC®1052 contributes this function to stabilize the circuit.
It does this by comparing the filtered circuit output to a
similarly filtered version of the input signal. The amplified
difference between these signals is used to set Q2’s bias,
and hence Q1’s channel current. This forces Q1’s VGS to
whatever voltage is required to match the circuit’s input
and output potentials. The 2000pF capacitor at A1 provides
stable loop compensation. The RC network in A1’s output
prevents it from seeing high speed edges coupled through
8pF
4.7k
Q1
4.7k
INPUT
Q2
–
2N5638
1
5
A1
LT318A
1k
OUTPUT
+
10k
10pF
4
10k
1000pF
–
A2
LT1012
–15V
1k
1k
+
AN21 F02
Figure 2. Fast DC-Stabilized FET Amplifier
A = 5V/DIV
B = 5V/DIV
HORIZONTAL = 100ns/DIV
AN21 F03
Figure 3. Figure 2’s Waveforms
an21f
AN21-2
Application Note 21
Q2’s collector-base junction. A2’s output is also fed back to
the shield around Q1’s gate lead, bootstrapping the circuit’s
effective input capacitance down to less than 1pF.
employed provide exceptionally wide bandwidth, fast
slewing and very little delay. Figure 5 shows the LTC1052
stabilized buffer circuit’s response using the discrete stage.
Response is clean and quick, with delay inside 4ns. Slew
exceeds 2000V/μs with full-power bandwidth approaching
50MHz. Note that rise time is limited by the pulse generator and not the circuit. For either stage, offset is set by the
LTC1052 at 5μV, with gain about 0.95.
The LT1010’s 15MHz bandwidth and 100V/μs slew rate,
combined with its 150mA output, are fast enough for most
circuits. For very fast requirements, the alternate discrete
component buffer shown will be useful. Although its output is current limited at 75mA, the GHz range transistors
5V
5V
TO POINT A
INPUT
1.5k
100Ω
Q1
2N5486
A
B
A2
LT1010
OUTPUT
FERRITE BEAD
FERRONICS
#21-110J
–5V
10M
–5V
8Ω
5V
8Ω
TO POINT B
OUTPUT
10M
0.1
0.1
1
4
10k
Q2
2N2222
+
8
6
A1
LTC1052
0.01
–
100Ω
NPN = 2N3866
PNP = 2N5160
0.1
3
AN21 F04
–5V
2
ALTERNATE BUFFER
7
5V
2000pF
–5V
1.5k
(4b)
1k
0.1
(4a)
Figure 4. Wideband FET Input Stabilized Buffer
A = 0.5V/DIV
B = 0.5V/DIV
HORIZONTAL = 10ns/DIV
AN21 F05
Figure 5. Figure 4’s Waveforms
an21f
AN21-3
Application Note 21
A potential difficulty with Figure 4’s circuit is that the gain
is not quite unity. Figure 6 maintains high speed and low
bias while achieving a true unity-gain transfer function.
A = 10 (e.g., 1k adjustment set at 50Ω) full-power bandwidth
stays at 10MHz while the –3dB point falls to 22MHz.
With the optional discrete stage, slew exceeds 1000V/μs
and full-power bandwidth (1VP-P) is 18MHz. –3dB bandwidth is 58MHz. At A = 10, full power is available to 10MHz,
with the –3dB point at 36MHz.
This circuit is somewhat similar to Figure 4, except that the
Q2-Q3 stage takes gain. A2 DC stabilizes the input-output
path, and A1 provides drive capability. Feedback is to Q2’s
emitter from A1’s output. The 1k adjustment allows the gain
to be precisely set to unity. With the LT1010 output stage
slew and full-power bandwidth (1VP-P) are 100V/μs and
10MHz, respectively. –3dB bandwidth exceeds 35MHz. At
Figures 7a and 7b show response with both output
stages. The LT1010 is used in Figure 7a (Trace A = input,
Trace B = output). Figure 7b uses the discrete stage and is
slightly faster. Either stage provides more than adequate
15V
1k
470
Q1
2N5486
INPUT
10pF
Q3
2N3906
0.01
15V
A
Q2
2N3904
B
A1
LT1010
OUTPUT
3k
2N3904
1k
1k
GAIN
ADJ
10M
10k
2k
300Ω
3Ω
A
50Ω
5.6k
3k
B
3Ω
10M
2N3906
0.1
3k
+
–15V
A2
LT1012
–
1k
0.1
0.002
–15V
(6b)
(6a)
Figure 6. Gain Trimmable Wideband FET Amplifier
A = 0.2V/DIV
B = 0.2V/DIV
A = 0.2V/DIV
B = 0.2V/DIV
HORIZONTAL = 10ns/DIV
AN21 F07a
Figure 7a. Figure 6’s Waveforms Using the LT1010
HORIZONTAL = 10ns/DIV
AN21 F07a
Figure 7b. Figure 6’s Waveforms Using Discrete Stage
an21f
AN21-4
Application Note 21
performance for driving video cable or data converters, and
the LT1012 maintains DC stability under all conditions.
Figure 8 is another DC-stabilized fast amplifier which functions over a wide range of gains (typically 1-10). It combines
the LT1010 and a fast discrete stage with an LT1008 based
DC stabilizing loop. Q1 and Q2 form a differential stage
which single-ends into the LT1010. The circuit delivers
1VP-P into a typical 75Ω video load. At A = 2, the gain is
within 0.5dB to 10MHz with the –3dB point occurring at
16MHz. At A = 10, the gain is flat (±0.5dB to 4MHz) with
a –3dB point at 8MHz. The peaking adjustment should be
optimized under loaded output conditions.
Normally, the Q1-Q2 pair would be quite drifty, but the
LT1008 corrects for this. This correction stage is similar
to the one in Figures 4 and 6, except that the feedback is
taken from a divided down sample of the fast amplifier.
The ratio of this divider should be set to the same value
as the circuit’s closed-loop gain. Frequency roll-off of this
stage is set by the 1M-0.022μF filters in the LT1008’s input
lines. The 0.22μF capacitor at the amplifier eliminates
oscillations. The DC loop servo controls drift by biasing
the DC operating point of Q2’s collector to force zero error
between the LT1008’s inputs.
This is a simple stage for fast applications where relatively
low output swing is required. Its 1VP-P output works nicely
for video circuits. A possible problem is the relatively high
bias current, typically 10μA. Additional swing is possible,
but more circuitry is needed.
0.22
0.22
–
LT1008
+
8
2k
0.22
15V
8.2k
1M
TYPICAL SPECIFICATIONS
1VP-P INTO 75Ω AT
A=2
1/2dB TO 10MHz
3dB DOWN AT 16MHz
AT A = 10
1/2dB TO 4MHz
–3dB = 8MHz
25Ω
1M
+
22μF
BIAS
+
22μF
OUTPUT
75Ω
LT1010
900Ω
–15V
PEAKING
5pF TO 25pF
INPUT
Q1 Q2
2N3866
s2
900Ω
R–GAIN DEPENDENT
SEE TEXT
AN21 F08
1k
GAIN SET
5.1k
–15V
Figure 8. Fast, Stabilized Noninverting Amplifier
an21f
AN21-5
Application Note 21
Figure 9’s circuit addresses these issues. It trades speed
for output swing and reduced bias current. As before, a
separate loop maintains DC stability. This circuit is a good
example of an approach made practical by composite
techniques. Without the separate stabilizing loop, the DC
imbalances in the signal path would preclude any level
of operation.
Figure 8’s bias current errors are eliminated by Q3, a FET
source follower. This device buffers the summing point from
the relatively high bias current required by Q2. Normally,
this configuration would cause volts of offset, due to Q3’s
gate-source voltage. Here, A1 closes a DC restoration
loop, forcing Q1’s base to whatever point is required to
compensate for the offset. Thus, A1’s operation not only
provides low DC error but permits a simplistic approach
to minimizing summing point bias current. Figure 10
shows operating waveforms for a 10V output. Trace A is
the input, while Trace B is the output. Slew rate is about
100V/μs, with a full-power bandwidth of 1MHz. The LT1010
allows 100mA outputs and makes cable driving practical
at these speeds.
In this arrangement a PNP level-shifting stage (Q4) has
been added to Figure 8’s circuit to increase available swing
at the LT1010 output. This is obtained at the expense
of available bandwidth and amplifier stability. The 33pF
capacitor from Q4’s collector to the circuits summing node
(Q3’s gate) affords stable loop compensation.
15V
7.5k
25Ω
100Ω
25
Q4
2N3906
+
BIAS
33pF
1k
A2
LT1010
INPUT
OUTPUT
15V
1k
Q3
2N4391
Q1
8pF
Q2
2N3866
s2
10k
5.6k
5.6k
1k
0.001
–15V
10k
AN21 F09
–
A1
LT1012
+
Figure 9. Fast, Stabilized Inverting Amplifier with Low Summing Point Bias Current
A = 5V/DIV
B = 5V/DIV
HORIZONTAL = 100ns/DIV
AN21 F10
Figure 10. Figure 9’s Pulse Response
an21f
AN21-6
Application Note 21
Figure 11 shows another fast stage with wide output swing.
The circuit is noninverting, and has higher input impedance than Figure 9. Additionally, it’s operation is based on
an arrangement commonly referred to as “current mode”
feedback. This technique, well established in RF design
and also employed in some monolithic instrumentation
amplifiers, permits fixed bandwidth over a wide range of
closed-loop gains. This contrasts with normal feedback
schemes, where bandwidth degrades as closed-loop gain
increases.
Q2. Correction is implemented by controlling the current
through Q3, which shunts Q2’s base bias resistor. Adequate
loop capture range is assured by deliberate skewing of
Q1’s operating point via the 330Ω unit. The 9k-1k feedback
divider feeding A3 is selected to equal the gain ratio of the
circuit, in this case 10.
The feedback scheme makes A1’s output look like the
negative input of the amplifier, with closed-loop gain set
by the ratio of the 470Ω and 51Ω resistors. The outstanding feature of this connection is that bandwidth becomes
relatively independent of closed-loop gain over a reasonable
range. For this circuit, full-power bandwidth remains at
1MHz over gains of 1 to about 20. The loop is quite stable,
and the 15pF value at A2’s input provides good damping over a wide range of gains. The LT1010 buffers limit
bandwidth in this circuit. Dramatic speed improvement is
possible if they are replaced by discrete stages.
The overall amplifier is composed of two LT1010 buffers
and a gain stage, Q1 and Q2. A3 acts as a DC restoration
loop. The 33Ω resistors sense A1’s operating current,
biasing Q1 and Q2. These devices furnish complementary
voltage gain to A2, which provides the circuit’s output.
Feedback is from A2’s output to A1’s output, which is a
low impedance point.
A3’s stabilizing loop compensates large offsets in the
signal path, which are dominated by mismatch in Q1 and
15V
33Ω
20Ω
330
Q1
2N2907
470Ω
A1
LT1010
INPUT
1M
51Ω
A2
LT1010
0.1
OUTPUT
15pF
BIAS
25μF
Q2
2N2222A
33Ω
9k
25Ω
20Ω
15V
–15V
0.002
1M
0.1
10k
1k
–
A3
LT1001
+
470Ω
+
Q2
2N2222A
AN21 F11
Figure 11. “Current Mode Feedback” Amplifier
an21f
AN21-7
Application Note 21
Figure 12 substitutes discrete elements for Figure 11’s
LT1010s. Although this arrangement is substantially more
complex, it provides an extraordinarily wideband amplifier.
This composite design is composed of three amplifiers;
the discrete wideband stage, a quiescent current control
amp and an offset servo. Q1-Q4 replace Figure 11’s
A1, although complementary voltage gain is taken at
the collectors Q3 and Q4. Q5 and Q6 provide additional
330pF
EOS
CONTROL
0.1
–
A2
1/2 LT1013
+
5.1k
1M
0.1
15V
2700pF
20Ω
20Ω
1.2k
1M
3k
Q5
Q9
100Ω
Q7
Q3
100Ω
3Ω
Q1
INPUT
–15V
470Ω
OUTPUT
–15V
51Ω
3Ω
*
15V
100Ω
9k
1k
15V
Q2
100Ω
Q4
Q8
Q10
Q6
3k
20Ω
–15V
10k
2700pF
0.05
2k
PNP = 2N3906
NPN = 2N3904
–
= 1N4148
*10pF TRIMMER (SEE TEXT)
A1
1/2 LT1013
+
IQ CONTROL
20Ω
1.2k
IQ ADJUST
SET AT
80mA
10k
AN21 F12
13.8k
15V
1.2k
Figure 12. Stabilized, Ultra-Wideband “Current Mode Feedback” Amplifier (Son of Godzilla Amplifier)
an21f
AN21-8
Application Note 21
gain, similar to Q1 and Q2 in Figure 11. Q7-Q10 form the
output buffer stage. The feedback scheme is identical to
Figure 11’s, with summing action at the Q3-Q4 emitter
connection. To obtain maximum bandwidth, quiescent
current is quite high. Without closed-loop control, the
circuit will quickly go into thermal runaway and destroy
itself. A1 provides the required servo control of quiescent
current. It downs this by sampling a resistively divided
version of the voltage across Q5’s emitter resistor and
comparing it to a power supply derived reference. A1’s
output biases Q4, completing a loop which forces fixed
current through Q5. This action effectively controls overall
quiescent current in the discrete stage. Simultaneously,
A2 corrects for offset by forcing Q3’s base to equalize the
DC input and output values at the discrete stage. Because
the closed-loop gain is set at 10 (470Ω and 51Ω ratio), A2
samples the output via the 10:1 divider. Both A1 and A2
have local roll-off, limiting their response to low frequency.
Casual consideration of A1 and A2’s operation might raise
concern about interaction, but detailed analysis shows this
is not so. The offset and quiescent current loops do not
influence each others operation.
When this circuit is constructed using high frequency
layout techniques and a ground plane, performance is
quite impressive. For gains of 1 to 20, full-power bandwidth remains at 25MHz, with the –3dB point beyond
110MHz. Slew rate exceeds 3000V/μs. These figures can
be improved upon by using RF transistors, although the
types shown are inexpensive and readily available. Figure
13 shows pulse response for a ±12V output (Trace B) at
a gain of 10 (input is Trace A). Delay is about 6ns, with
rise time limited by the input pulse generator. Damping
is optimized with the 10pF trimmer at the Q5-Q6 collector line. To use this circuit, adjust the IQ level to 80mA
IMMEDIATELY after turn on. Next, set A2’s input resistor
divider to a ratio appropriate to the closed-loop circuit
gain. Finally, adjust the 10pF trimmer for best response.
Note that, in the interests of speed, this circuit has no
output protection.
A = 0.4V/DIV
B = 4V/DIV
HORIZONTAL = 10ns/DIV
AN21 F13
Figure 13. Figure 12’s Pulse Response (Measurement Limited by Pulse Generator)
an21f
AN21-9
Application Note 21
Although speed and offset combinations are the most
common area for composite techniques, other circuits
are possible. Figure 14 shows a way to combine a low
drift chopper-stabilized amplifier with an ultralow noise
bipolar amplifier. The LTC1052 measures the DC error at
the LT1028’s input terminals and biases its offset pins to
force offset to a few microvolts. The 1N758 Zeners allow
the LTC1052 to function from ±15V rails. The offset pin
biasing at the LT1028 is arranged so the LTC1052 will always
be able to find the servo point. The 0.01μF capacitor rolls
off the LTC1052 at low frequency, and the LT1028 handles
high frequency signals. The combined characteristics of
these amplifiers yield the following performance:
Offset Voltage
5μV Max
Offset Drift
50nV/°C Max
Noise
1.1nV √Hz Max
15V
1N758
10V
+
INPUT
7
A1
LTC1052
–
8
4
1
0.1
0.1
0.01
15V
1N758
10V
30k
–15V
100k
130Ω
68Ω
1
+
A2
LT1028
–
15V
7
4
8
OUTPUT
10k
–15V
(A = 1000)
10Ω
AN21 F14
Figure 14. DC-Stabilized, Low Noise Amplifier
an21f
AN21-10
Application Note 21
Figure 15 plots noise amplitude over time in a 0.1Hz to
10Hz bandwidth.
Figure 16 uses multiple LT1028 low noise amplifiers in a
statistical noise reduction technique. It is based on the fact
that noise decreases by the √N of the number of devices in
parallel. For example, for nine paralleled amplifiers, noise
would decrease by a factor of three, to about 0.33nV√Hz
at 1kHz. A potential penalty of this connection is that the
input current noise increases by √N devices.
20nV
AN21 F15
2 SEC
0.1Hz TO 10Hz VOLTAGE NOISE
Figure 15. Figure 14’s Noise vs Time
+
A1
LT1028
1.5k
–
7.5Ω
GAIN = N s 200
OUTPUT NOISE = √N s 200 s 1.1nV/√Hz
INPUT NOISE = OUTPUT NOISE = 1.1 nV/√Hz
N x 200
√N
470Ω
4.7k
+
A2
LT1028
1.5k
–
7.5Ω
+
470Ω
+
INPUT
AN
LT1028
–
1.5k
LT1028
OUTPUT
WHERE N = AS MANY AMPLIFIERS
AS DESIRED
–
7.5Ω
470Ω
AN21 F16
Figure 16. Low Noise Amplifier Using Paralleled Amplifiers
an21f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN21-11
Application Note 21
A final circuit, Figure 17, uses a composite of paralleled
LT1010 buffers to create a simple, high current stage.
Parallel operation provides reduced output impedance,
more drive capability and increased frequency response
under load. Any number of LT1010s can be directly paralleled as long as the increased dissipation in individual
units caused by mismatches of output resistance of offset
voltage is taken into account.
When the inputs and outputs of two buffers are connected
together, a current, ΔIOUT , flows between the outputs:
ΔIOUT =
VOS1 – VOS2
ROUT1 + ROUT2
where VOS and ROUT are the offset voltage and output
resistance of the respective buffers.
Normally, the negative supply current of one unit will
increase and the other decrease, with the positive supply
current staying the same. The worst-case (VIN → V+)
increase in standby dissipation can be assumed to be
ΔIOUT VT , where VT is the total supply voltage.
Offset voltage is specified worst-case over a range of supply voltages, input voltage and temperature. It would be
unrealistic to use these worst-case numbers above because
paralleled units are operating under identical conditions.
The offset voltage specified for VS = ±15V, VIN = 0 and
TA = 25°C will suffice for a worst-case condition.
Output load current will be divided based on the output
resistance of the individual buffers. Therefore, the available output current will not quite be doubled unless output
resistances are matched. As for offset voltage above, the
25°C limits should be used for worst-case calculations.
Parallel operation is not thermally unstable. Should one
unit get hotter than its mates, its share of the output and
its standby dissipation will decrease.
As a practical matter, parallel connection needs only some
increased attention to heat sinking. In some applications
a few ohms equalization resistance in each output may be
wise. Only the most demanding applications should require
matching, and then just of output resistance at 25°C.
V+
IS
VIN
IS
A1
LT1010
VOUT
ΔIOUT
A2
LT1010
IS – ΔIOUT
IS + ΔIOUT
AN21 F17
V–
Figure 17. Paralleling Scheme for High Current Output
an21f
AN21-12
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